JP2002209386A - Power conversion device and drive control method for polyphase load - Google Patents

Abstract

PROBLEM TO BE SOLVED: To inhibit the generation of a switching loss in a switching element, as much as possible for constituting an inverter main circuit of a power conversion device. SOLUTION: A command value converter performs conversion so that a biphase command value level can be fixed to the maximum (steps A5, A9, and A13) or the minimum (steps A4, A8, and A12), in a period (steps A2, A6, A10: Yes) in which either biphase command value level among three-phase voltage command values U*, V*, and W* outputted from a command value generator is less than a prescribed value x.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power converter for driving a polyphase load by an inverter main circuit, and a drive control method for the polyphase load.

[0002]

2. Description of the Related Art FIG. 25 shows an electrical configuration of an inverter device for driving a polyphase AC motor used for traveling of an electric vehicle, for example. In FIG. 25, six IGBTs 1 to 6 are connected in a three-phase bridge to form an inverter main circuit 7. DC buses 7a and 7b of the inverter main circuit 7 are connected to a positive terminal and a negative terminal of a driving battery 8, respectively. It is connected. The output terminals 7u, 7v, 7w of the inverter main circuit 7 are respectively connected to respective phase windings (not shown) of a three-phase polyphase AC motor (for example, a synchronous motor, an induction motor, a brushless motor, etc.) 9. ing. Freewheeling diodes D1 to D6 are connected in reverse z parallel between the collector and emitter of each of the IGBTs 1 to 6.

The command value generator 10 has voltage command values U *, V
*, W * are stored around the ROM. For example, the output terminals 7u to 7w of the inverter main circuit 7 and each phase winding of a polyphase AC motor (hereinafter simply referred to as a motor) 9 are provided. A phase θ of a rotor constituting the AC motor 9 is detected based on a zero-cross point in an output signal of a current sensor (not shown) disposed between the AC motor 9 and the output signal of a rotary encoder or a resolver. Voltage command values U *, U *,
V * and W * are read and output to the PWM waveform generator 11. The voltage command values U *, V *, W *
Is a command value based on the amplitude of a sine wave, for example.

FIG. 26 is a functional block diagram showing a detailed electrical configuration of the PWM waveform generator 11. As shown in FIG. The voltage command values U *, V *, W * output from the command value generator 10 are given to the non-inverting input terminals of the comparators 12a, 12c, 12e and the inverting input terminals of the comparators 12b, 12d, 12f. Comparators 12a, 12c, 1
2e inverting input terminal and comparators 12b and 12
The non-inverting input terminals of d and 12f are provided with a PWM-modulated carrier (triangular wave) output from the carrier generator 13.

The comparators 12a to 12f are magnitude comparators when the voltage command values U *, V *, W * and the carrier are all output as digital data, and both of them are output as analog data. The case is an analog comparator.

Then, the comparators 12a, 12c, 1
From 2e, the voltage command values U *, V
The signal C which becomes a high level when the levels of * and W * are high
1, C3 and C5 are output and the comparators 12b and 12
d, 12f, the inverted signal C of the signals C1, C3, C5
2, C4 and C6 are output and supplied to the dead time generator 14. The dead time generator 14 provides a signal C1 / to provide a dead time in which both the positive and negative IGBTs of one arm are turned off in order to prevent the IGBTs on both sides from being turned on at the same time. C3 / C
5, on / off timing between C2 / C4 / C6.

The dead time generator 14 outputs gate signals G1 'to G6'.
1 'to G6' are drivers 1 composed of a photocoupler or the like.
5, a gate signal G1 is applied to the gates of the IGBTs 1-6.
To G6.

In this control method, for example, considering the U phase, if the command value U * is larger than the triangular wave as the carrier, IGBT1 is turned on, IGBT2 is turned off, and DC
The positive potential of the power supply is output. Conversely, when U * is smaller than the triangular wave, IGBT1 is turned off, IGBT2 is turned on, and the negative potential of the DC power supply is output. By this operation, the positive voltage of the DC power supply is output for a time proportional to the command value in each carrier cycle.

As shown in FIG. 27, if the voltage command values U *, V *, W * are sine waves, the pulse width becomes a sine wave.
The WM-modulated voltage is output, and the output current can be made substantially sinusoidal. Further, as the frequency of the carrier wave increases, a current closer to a more ideal sinusoidal waveform can be output. If the frequency is set to 15 kHz or more, the magnetic noise of the motor 9 can be significantly reduced. Therefore, the inverter main circuit 7 uses IGBTs 1 to 6 that can perform such high-speed switching.

[0010]

However, when the inverter main circuit 7 is driven by a large amount of power, since heat generated by power conversion loss is large, cooling by water cooling or the like is necessary, and the system can be reduced in size and cost can be reduced. It is an obstacle to the development. Switching losses in the IGBTs 1 to 6 occupy a large proportion of the power conversion loss. Since the switching loss increases as the switching frequency increases, there is a problem that it cannot be used at a sufficiently high frequency.

The present invention has been made in view of the above circumstances, and an object of the present invention is to provide a power conversion device capable of minimizing the occurrence of switching loss in a switching element constituting an inverter main circuit, and a polyphase load. Is to provide a drive control method.

[0012]

According to a first aspect of the present invention, when the command value output means outputs a voltage command value for each phase to a polyphase load, the command value conversion means: During a period in which at least any two-phase voltage command values of these voltage command values are substantially equal, conversion is performed so that these two-phase voltage command values are equal to the maximum or the minimum and output.
The control signal output means outputs a switching control signal to the inverter main circuit driving the multi-phase load by pulse width modulating the carrier with the voltage command value output to the command value conversion means.

That is, at least during the period when the two-phase voltage command values are substantially equal, the converted voltage command values in the control signal output means are converted so that the voltage command values are equal to maximum or minimum. Is 100% or 0%. That is, the switching of the at least two-phase switching element is stopped, and the total number of switching operations is reduced. Therefore, the switching loss can be suppressed without lowering the switching frequency, and the cooling mechanism can be reduced in size and simplified, so that the entire apparatus can be made compact.

Since the control signal output means performs pulse width modulation, the present invention can be effectively applied to a control method in which switching is performed a large number of times at an extremely high frequency as compared with the driving cycle of the polyphase load. .

According to the second aspect of the present invention, the command value converting means calculates a voltage command value for a phase other than a conversion target by averaging a line voltage of each of the voltage command values of the phases to be converted. Output as a value. Therefore, even if the conversion is performed so that the voltage command value is equal to maximum or minimum,
The distortion of the phase current waveform can be suppressed, and the total drive power can be maintained substantially equal to that before the conversion of the command value.

According to the power converter of the third aspect, the present invention can be applied when the polyphase AC motor is a load. According to the power converter of the fourth aspect, the present invention can be applied to a configuration for driving a three-phase AC motor that is generally widely used.

According to a fifth aspect of the present invention, when the command value output means outputs a voltage command value for each phase to the multi-phase load, the command value conversion means outputs the voltage command value. Among them, in a period in which at least one of the two-phase voltage command values is substantially equal, a first conversion is performed so that the at least one of the two-phase voltage command values is equal to the maximum or the minimum.
The conversion period and the second conversion period for converting only the one-phase voltage command value to be the maximum or the minimum are output so as to be alternately repeated. And the control signal output means,
By modulating the carrier with the voltage command value output to the command value conversion means, a switching control signal is output to the inverter main circuit that drives the polyphase load.

That is, a first conversion period in which the modulation degree of at least one of the two-phase voltage command values is 100% or 0%;
The second conversion period in which the modulation degree of the one-phase voltage command value is 100% or 0% is alternately executed, and the number of times of switching while reducing the distortion of the phase current of the polyphase load is further reduced. Can be reduced. Therefore, the multi-phase load can be driven with lower vibration and lower noise.

According to the power converter of the present invention, since the command value converter sets the conversion period to an integral multiple of the carrier wave period, the conversion process can be easily executed at a timing synchronized with the carrier wave. . Further, the first conversion period and the second conversion period
Switching can be prevented from occurring at the time of switching to the conversion period.

According to the power converter of the present invention, the command value converting means sets the phase to be converted in the second conversion period so that the line voltage with the voltage command value of the phase other than the conversion target is the maximum. Is set to With such a setting, it is possible to further reduce the influence of distortion when the voltage command value is converted to be the maximum or the minimum.

According to the power conversion device of the eighth aspect, the command value conversion means sets the second conversion period to n times the first conversion period, and in the first conversion period, sets the voltage command for the phase other than the conversion target. Level is shifted in the second conversion period so as to maintain the line voltage with the voltage command value of the phase to be converted,
In the conversion period, the line command voltage of the phase to be converted only in the first conversion period and the voltage command value of the phase to be converted in the second conversion period are (1 + 1 / n) times. Level shift as follows.

Therefore, the distortion caused by the conversion of the voltage command value to the maximum or the minimum in each of the first and second conversion periods can be corrected by level-shifting the command values other than those to be converted. Further, for example, if the second conversion period is set to n (> 1) times the first conversion period, the level shift of the command value in the second conversion period is less than twice the line voltage. Thus, it is possible to prevent the phase current waveform from being distorted.

According to the ninth aspect of the present invention, since the command value conversion means sets the value of n to "1", the first conversion period and the second conversion period are equal in time. Control becomes easy.

According to the power converter of the tenth aspect,
Since the command value conversion means changes the ratio between the first conversion period and the second conversion period, it becomes possible to change the correction amount of the line voltage in response to the change in the voltage command value. . Therefore, the adjustment can be performed so as to prevent the overmodulation state.

According to the eleventh aspect of the present invention,
The command value conversion means sets the second conversion period relatively long as the difference between the voltage command values before conversion to be converted in the first conversion period output by the command value output means increases. That is, it is desirable to increase the amount of correction (level shift) of the line voltage as the difference between the voltage command values increases. However, if the level shift amount is increased, there is a possibility that an over-modulation state may occur. Therefore, if the second conversion period is set longer as the difference between the voltage command values increases, the value of n increases, and the level shift amount decreases accordingly. Therefore, it is possible to effectively prevent an overmodulation state.

According to the power converter of the twelfth aspect,
The invention according to any one of claims 5 to 10 can be applied to a configuration for driving a three-phase AC load that is generally widely used.

According to the power converter of claim 13,
When the command value output means outputs a voltage command value corresponding to each phase of the multi-phase load, the command value conversion means determines that at least one of the two-phase voltage command values is equal based on the magnitude relation between the voltage command values. During a specific period including a certain point in time, the conversion is performed such that the at least two-phase voltage command values are equal to the maximum or the minimum and output. And the control signal output means,
The switching control signal is output to the inverter main circuit that drives the multi-phase load by pulse width modulating the carrier with the voltage command value output to the command value conversion means.

That is, the command value conversion means can easily determine the specific period in which at least the two-phase voltage command values are fixed so as to be substantially equal based only on the magnitude relation of the voltage command values. Therefore, in the region where the driving frequency of the multi-phase load is high, the specific period can be determined and followed in a short time.

According to the power converter of claim 14,
Since the command value output means outputs a voltage command value based on a change in the amplitude of the sine wave, the phase current waveform of the polyphase load can be made substantially a sine wave, and the motor is driven with extremely low vibration and low noise. It becomes possible.

[0030]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS First Embodiment A first embodiment of the present invention will be described below with reference to FIGS.
The same parts as those in FIGS. 25 and 26 are denoted by the same reference numerals, and description thereof will be omitted. Only different parts will be described below.
In FIG. 1 showing the functional blocks of the electrical configuration, a command value converter (command value conversion means) is provided between a command value generator (command value output means) 10 and a PWM waveform generator (control signal output means) 11. ) 21 is inserted.

The command value converter 21 includes, for example, a DSP (Digital S) together with the command value generator 10 and the PWM waveform generator 11.
The voltage command values U *, V *, W * output from the command value generator 10 are newer than the voltage command values U **, V **, and W * according to the calculation program of the flowchart shown in FIG. V ** and W ** are converted and generated, and PWM
It is configured to output to the waveform generator 11. Other configurations are the same as those in FIGS. 25 and 26.

Next, the operation of this embodiment will be described with reference to FIGS. In FIG. 2 showing a flowchart of the calculation program in the command value converter 21, the command value converter 21 first reads the voltage command values U *, V *, W * output from the command value generator 10 (step S1). A1) It is determined whether or not the absolute value of the difference between the amplitude levels of the two-phase voltage command values is smaller than a predetermined value x.
The determination is made in steps A2, A6 and A10, respectively.

Here, FIG. 4A shows the voltage command values U *,
Although the waveform data of V * and W * and the waveform of the carrier that is PWM-modulated in the PWM waveform generator 11 are superimposed on each other, the period of the carrier is longer than the actual period for convenience of illustration. Although the amplitude on the vertical axis is ± 1, the actual data is output by shifting the amplitude level from −1 to +1 to “0 to 255” in the case of 8 bits, for example.

FIG. 4B is an enlarged view of the portion indicated by the arrow in FIG. 4A, which is the vicinity where the values of the voltage command values V * and W * intersect. Is a region where the values are relatively similar. In the flowchart of FIG. 2, as shown in FIG. 4B, the command value level is set to “0” during a period in which the absolute value of the difference between the voltage command values V * and W * is smaller than the predetermined value x. The command values V ** and W ** converted into are output.

That is, for example, during the period shown in FIG. 4B, the command value converter 21 determines "YES" in step A6.
Then, it is determined whether the polarity of the voltage command value U * is positive (the 8-bit data value is greater than "127") (step A7). If the polarity of command value U * is positive (“YES”), command value converter 21 outputs voltage command value V
** and W ** are converted and generated as follows (step A
8). V ** = W ** = − 1 (data value “0”) (1) By doing so, the modulation degree of the V phase, the W phase, and the PWM signal becomes 0%, that is, the duty becomes 0%,
The phases are no longer switched. At this time, the remaining U-phase is converted into a voltage command value U ** which is level-shifted as follows so that the deviation of the line voltage between the V-phase and the W-phase of the line voltage is minimized. U ** = U *-(V * + W *) / 2-1 (2) That is, the level shift of the command value U ** in the equation (2) is as follows.
The average of the line command voltages U * and V * and W * is taken. Then, Step A15
Then, the converted voltage command values U **, V **, W ** are output to the PWM waveform generator 11.

The above two equations are conversion equations when the U-phase is positive and the V-phase and the W-phase are negative and cross, but conversely, the U-phase is negative and the V- and W-phases are positive. The conversion formula in the case of crossing is (step A7, NO), after conversion using the following formulas (3) and (4) instead of formulas (1) and (2) (step A7).
9) The process proceeds to step A15.

V ** = W ** = + 1 (data value “255”) (3) U ** = U * − (V * + W *) / 2 + 1 (4) Thus, the V phase and W The modulation degree of the phase and the PWM signal is 10
0%, that is, the duty becomes 100%, and the V phase and the W phase are not switched.

During a period in which the absolute value of the difference between voltage command values U * and V * is smaller than predetermined value x, command value converter 21 determines "YES" in step A2 and returns to steps A3 to A3. In A5, the processes corresponding to steps A7 to A9 are performed by exchanging phases. That is, the command value W
If the polarity of * is positive (step A3, "YES"),
The command value converter 21 converts and generates the voltage command values U **, V **, W ** as follows (step A4). U ** = V ** =-1 (5) W ** = W *-(U * + V *) / 2-1 (6) If the polarity of the command value W * is negative (step) A3
"NO"), the command value converter 21 outputs the voltage command values U **, V
** and W ** are converted as shown in equations (7) and (8) (step A5). U ** = V ** = 1 (7) W ** = W *-(U * + V *) / 2 + 1 (8)

Similarly, during a period in which the absolute value of the difference between voltage command values W * and U * is smaller than predetermined value x, command value converter 2
No. 1 determines "YES" in step A10, and performs the processes corresponding to steps A7 to A9 in steps A11 to A13 with the phases switched. That is, if the polarity of the command value V * is positive (step A11, “YE
S "), the command value converter 21 outputs the voltage command values U **, V **,
W ** is converted and generated as follows (step A12). U ** = W ** =-1 (9) V ** = V *-(W * + U *) / 2-1 (10) If the polarity of the command value V * is negative (step A1
1, "NO"), the command value converter 21 outputs the voltage command value U *
*, V **, and W ** are converted as in equations (11) and (12) (step A13). U ** = W ** = 1 (11) V ** = V *-(W * + U *) / 2 + 1 (12)

Then, the command value converter 21 executes step A
If it is determined “NO” in any of A2, A6, and A12, the process proceeds to step A14, and the voltage command values U *, V
*, W * are directly used as voltage command values U **, V **, W **, and the process proceeds to step A15. The voltage command values U **, V **, W ** converted by the command value converter 21 as described above.
3 (d), (e), and (f).

As described above, according to the present embodiment, the command value converter 21 outputs the three-phase voltage command values U *, V *, W *
During a period in which any two-phase command value level is smaller than the predetermined value x, the two-phase command value level is set to the maximum (+1).
Alternatively, since the conversion is performed so as to be fixed to the minimum (−1), switching is not performed for the two phases during that period.

Therefore, the IGBT of the inverter main circuit 7 is controlled by the PWM control without lowering the frequency of the carrier.
(Switching elements) The number of switching operations performed on the switching elements 1 to 6 can be reduced to reduce switching loss. Therefore, the cooling mechanism can be downsized and simplified, and the apparatus can be downsized and the cost can be reduced. In addition, the switching frequency can be set higher.

Further, according to this embodiment, the voltage command values for the remaining one phase other than the conversion target are output as the average values of the line voltages with the two-phase voltage command values to be converted. In addition, the distortion of the phase current waveform of the motor (polyphase AC motor, polyphase load) 9 can be suppressed, and the driving power can be made substantially equal to the case where the driving power is driven by the voltage command value before conversion. Further, according to the present embodiment, the command value generator 10
Are voltage command values U *, V *,
Since W * is output, the phase current waveform of the motor 9 can be made substantially sinusoidal, and the motor 9 can be driven with extremely low vibration and low noise.

Although the output current is slightly distorted by fixing the two-phase voltage command value for a certain period as in this embodiment, a relatively large load such as an electric vehicle or a hybrid electric vehicle is generated. In a system for driving a small amount, slight distortion is not particularly a problem.

(Second Embodiment) FIGS. 5 to 11 show a second embodiment of the present invention. The same parts as those of the first embodiment are denoted by the same reference numerals, and the description thereof will be omitted. Only the portions will be described. PWM instead of PWM waveform generator 11
The M waveform generator (control signal output means) 22 outputs the clock signals ck1 and ck2 to a command value converter (command value conversion means) 23 instead of the command value converter 21.

Here, referring to FIG. 6, clock signal c
k1 is a clock signal whose cycle is twice as long as the carrier cycle of the PWM signal and rises in synchronization with the point when the level of the carrier is "0". Also, the clock signal ck2
Is a clock signal whose phase is delayed by 90 ° from the clock signal ck1. And the command value converter 23
Performs the conversion process of the voltage command values U *, V *, W * with reference to the levels of the clock signals ck1, ck2.

Next, the operation of the second embodiment will be described with reference to FIGS. 7 to 9 show a part of a flowchart of a conversion process in the command value converter 23. FIG. 7 shows step A3 in the first embodiment.
FIG. 8 shows step A7 and subsequent steps, and FIG. 9 shows step A
The processing after 11 is partially modified.

First, the portion shown in FIG. 8 will be described.
Command value converter 23 determines "YES" in step A7.
Is determined, the result of the difference between the command values V * and W * is negative (V
* <W *) (step A23).
If it is negative ("YES"), the process shifts to step A24 to determine whether or not the level of the clock signal ck2 is "1 (high level)".

If the level of the clock signal ck2 is "1"("YES"), the flow shifts to step A25 to convert and generate the voltage command values U **, V **, W ** as follows. U ** = U * -V * -1 (13) V ** = W ** =-1 (14) It should be noted that the voltage command value U ** is the same as the U-phase and V-phase in the original command value. Are shifted so as to maintain the line voltage.

On the other hand, if the level of the clock signal ck2 is "0 (low level)" in step A24 ("NO"), the flow shifts to step A26 where the voltage command value U
**, V **, W ** are converted and generated as follows. U ** = U * -V * -1 (15) V ** =-1 (16) W ** = 2 (W * -V *)-1 (17) In this case, the voltage command value W ** is level-shifted so as to output a voltage twice the line voltage between the W phase and the V phase in the original command value. In other words, this is to compensate for the command value W ** fixed to “−1” in the immediately preceding carrier cycle.

In step A23, the command value V
If the result of the difference between * and W * is positive (V *> W *) ("NO"), the processing shifts to step A27 and step A2
As in the case of No. 4, it is determined whether or not the level of the clock signal ck2 is "1". If the level of the clock signal ck2 is "1"("YES"), the flow shifts to step A28, and the equation (13) is replaced with the equation (18) and executed. U ** = U * -W * -1 (18)

On the other hand, if the level of the clock signal ck2 is "0" in step A27 ("NO"), the flow shifts to step A29 to change the voltage command values U **, V **, W ** as follows. Is generated. U ** = U * -W * -1 (19) V ** = 2 (V * -W *)-1 (20) W ** =-1 (21) That is, in steps A28 and A29. The processing is performed in accordance with the command values V * and W * with respect to the processing of steps A25 and A26.
This is based on the fact that the magnitude relationship with

FIG. 10 shows the voltage command values U **, V **, W ** thus converted and generated. That is, in the period d, the voltage command values V * and W * intersect and their magnitudes are switched, so that V * <W * in the first half periods a to d and V * in the second half periods d to g. > W *. In the periods b, d, and f (when the level of the clock signal ck2 is “1”), the voltage command values V ** and W ** are fixed to the minimum level, and one phase is performed only with the voltage command value U **. Modulation is performed (first conversion period).

On the other hand, in the first half periods a and c during which the level of the clock signal ck2 becomes "0", only the voltage command value V ** is fixed to the minimum level, and the voltage command values U ** and W ** are used. Phase modulation is performed, and in the latter periods e and g, only the voltage command value W ** is fixed to the minimum level, and one-phase modulation is performed only with the voltage command values U ** and V ** (second conversion period). . Therefore, the one-phase modulation and the two-phase modulation are performed alternately every carrier cycle (that is, n = 1).

The case where the voltage command value U * is negative and "NO" is determined in step A7 is not specifically shown, but is the same as in the first embodiment (13). (21), the expression described as “A ** = − 1” is replaced with “A ** = 1”, and “A ** = B * −C * −1” or “A ** = ** =
2 (B * -C *)-1 "as" A ** =
B * -C * + 1 "or" A ** = 2 (B * -C *)
Only some of the signs need to be changed like +1 "
A, B and C are U, V and W phases). However, in this case, the clock signal ck1 is referred to instead of the clock signal ck2. Steps corresponding to step A23 are "A * -B * <0" and "A * -B *>0".

Similarly, in FIG. 7, the command value converter 2
3 is "YES" in step A3,
It is determined whether or not the result of the difference between the command values U * and V * is negative (U * <V *) (step A16). If the result is negative ("YES"), the process proceeds to step A17. If the level of the clock signal ck2 is "1"("YE
S ") The process proceeds to Step A18, where the voltage command values U **, V
** and W ** are converted and generated as follows. U ** = V ** =-1 (22) W ** = W * -U * -1 (23)

On the other hand, if the level of the clock signal ck2 is "0" in step A17 ("NO"), the flow shifts to step A19 to change the voltage command values U **, V **, W ** as follows. Is generated. U ** =-1 (24) V ** = 2 (V * -U *)-1 (25) W ** = W * -U * -1 (26) Also, U *> V * In step A21 in the case of
The conversion is performed by replacing the equation (23) in step A18 with the following equation (27), and W ** = W * -V * -1 (27) In step A22, the equations (24) to (26) are used. Instead, conversion is performed according to the following equations (28) to (30). U ** = 2 (U * -V *)-1 (28) V ** =-1 (29) W ** = W * -V * -1 (30)

In FIG. 9, the command value converter 23
Determines “YES” in step A11,
It is determined whether or not the result of the difference between the command values W * and U * is negative (W * <U *) (step A30). If the result is negative ("YES"), the process proceeds to step A31. If the level of the clock signal ck2 is "1"("YE
S ") The process proceeds to step A32, where the voltage command values U **, V
** and W ** are converted and generated as follows. U ** = W ** =-1 (31) V ** = V * -W * -1 (32)

On the other hand, if the level of the clock signal ck2 is "0" in step A31 ("NO"), the flow shifts to step A33 to change the voltage command values U **, V **, W ** as follows. Is generated. U ** = 2 (U * -W *)-1 (33) V ** = V * -W * -1 (34) W ** =-1 (35) Also, W *> U * In step A35 in the case of
The conversion is performed by replacing the equation (32) in the step A32 with the following equation (36). V ** = V * -U * -1 (36) In the step A36, the equations (33) to (35) are used. Instead, conversion is performed by the following equations (37) to (39). U ** =-1 (37) V ** = V * -U * -1 (38) W ** = 2 (W * -U *)-1 (39)

The voltage command value U converted and generated as described above
FIG. 11 shows voltage waveforms for one cycle of the command value of **, V **, and W **. FIGS. 12A and 12B show simulation results of phase current waveforms output by the voltage command values U **, V **, and W ** of the first embodiment and the second embodiment, respectively. Things. Comparing the two, it can be seen that the current waveform according to the first embodiment is slightly distorted, but the current waveform according to the second embodiment has little distortion and is improved.

As described above, according to the second embodiment, the command value converter 23 determines that any two-phase command value level among the three-phase voltage command values U *, V *, W * is a predetermined value. In a period that is smaller than x, the first conversion period in which the two-phase command value levels are fixed to the maximum or the minimum and one-phase modulation is performed, and the command value level of any one of the two phases is set to the maximum or the minimum. The second conversion period in which the two-phase modulation is fixed at the minimum is alternately executed at the same time ratio.

Then, in the first conversion period, the voltage command value of the phase other than the conversion target (for example, the U phase in the example of the equations (13) to (17)) is converted into the phase (the conversion target) in the second conversion period. Same V
Level is shifted so as to maintain the line voltage with the voltage command value of the phase (phase), and in the second conversion period, the phase (V) to be converted is set.
Is set to the phase in which the line voltage with the voltage command value of the phase (U, W) other than the conversion target is the maximum, and the voltage command value of the phase (W) to be converted only in the first conversion period To
In the second conversion period, the level was shifted so that the line voltage with the voltage command value of the phase (V) to be converted was doubled.

Accordingly, the number of times of switching can be reduced as compared with the related art, and the distortion (insufficient line voltage) caused by the voltage command value being converted to the maximum or minimum in each of the first and second conversion periods. ) Can be corrected by level-shifting command values other than those to be converted, so that the phase current waveform output to the motor 9 can be made less distorted. This can be done with noise. In addition, the command value converter 23
Since the conversion period is set to twice the carrier cycle, the conversion process can be easily executed at a timing synchronized with the carrier.

(Third Embodiment) FIG. 13 shows a third embodiment of the present invention. In the second embodiment, a first conversion period in which the two-phase voltage command value is fixed to the maximum or the minimum to perform one-phase modulation, and a second conversion period in which the one-phase voltage command value is fixed to perform the two-phase modulation Are alternately executed for each carrier cycle. However, in the third embodiment, the ratio is set to 1: 2 between the first and second conversion periods.

In this case, during the second conversion period, the second
For example, as a process corresponding to step A26 in the embodiment,
The equation (17) is replaced with the equation (40) and executed. W ** = 1.5 (W * −V *) − 1 (40) Further, as processing corresponding to step A29, (20)
The equation is replaced with the equation (41) and executed. V ** = 1.5 (V * -W *)-1 (41)

Here, the coefficient “1.5” indicates that the second conversion period for performing the two-phase modulation is twice as long as the first conversion period. This is because the correction amount of the phase line voltage is reduced. That is, the first and second
If the conversion period ratio is 1: n, for example, (4
Equation (0) is expressed as follows. W ** = (1 + 1 / n) (W * -V *)-1 (40a) Conversely, the ratio of the first and second conversion periods is m: 1 (or n is 1: n) Is less than 1).
It is necessary to increase the correction amount of the line voltage, and (40)
The formula is expressed as follows. W ** = (1 + m) (W * −V *) − 1 (40b)

The timing of the first and second conversion periods is determined, for example, by multiplying the clock signal ck2 to generate a clock signal having the same cycle as the carrier wave, and inputting the rising edge of the clock signal. A state counter that repeatedly counts 1 to 3 ″ is used. And
If the count value is “1”, the first conversion period is executed, and if the count value is “2, 3”, the second conversion period is executed.

As described above, according to the third embodiment, the ratio between the first conversion period for performing one-phase modulation and the second conversion period for performing two-phase modulation is set to, for example, 1: 2. The same effect as the example can be obtained. And the second which performs two-phase modulation
Since the amount of correction (level shift) of the line voltage is reduced as the conversion period is made relatively longer than the first conversion period, an overmodulation state occurs and distortion occurs in the phase current waveform. Can be prevented.

(Fourth Embodiment) FIG. 14 shows a fourth embodiment of the present invention. In the fourth embodiment, the time ratio between the first and second conversion periods is dynamically changed. In the periods a and c shown in FIG.
2, and the ratio is 1: 1 in the period b. That is, as the difference between the voltage command values V * and W * increases,
The second conversion period is dynamically changed so as to be set relatively long.

Here, for example, as shown in FIG.
The difference between the voltage command values V * and W * gradually increases with reference to the point in time at which the command value curves intersect during a period in which the voltage command values are substantially equal to each other within the predetermined value x. Accordingly, it is desirable to increase the correction (level shift) amount of the line voltage as the difference increases. However, if the level shift amount is increased, an over-modulation state may occur.

Therefore, in the fourth embodiment, the voltage command value V
By setting the second conversion period longer as the difference between * and W * becomes larger, the value of n in the equation (40a) becomes larger, so that the level shift amount can be reduced accordingly, and the excess The modulation state can be effectively prevented. In the second and third embodiments, the ratio between the two periods is fixed to 1: 1 or 1: 2, so that the change period of the voltage signal (1/2 of the carrier wave period, 1/3). Depending on,
It is expected that the frequency spectrum of the unwanted radiation generated from the device has a relatively large peak at a specific point.

On the other hand, according to the fourth embodiment, since the change period of the voltage signal is dynamically changed, the frequency spectrum can be prevented from being dispersed and concentrated on a specific point, and unnecessary radiation can be avoided. Measures can be taken more easily.

(Fifth Embodiment) FIG. 15 shows a fifth embodiment of the present invention. In the fifth embodiment, contrary to the third embodiment, the execution is performed with the ratio of the first and second conversion periods being 2: 1. In this case, in the second conversion period, the equation (40) is replaced with the equation (42) and executed. W ** = 3 (W * -V *)-1 (42) Further, the equation (41) is replaced with the equation (43) and executed. V ** = 3 (V * -W *)-1 (43)

As described above, according to the fifth embodiment, the ratio between the first conversion period for performing one-phase modulation and the second conversion period for performing two-phase modulation is set to 2: 1. The same effect can be obtained.

(Sixth Embodiment) FIGS. 16 to 18 show a sixth embodiment of the present invention. Only parts different from the first embodiment will be described. The configuration of the sixth embodiment shown in FIG. 16 is obtained by replacing the command value converter 21 in the first embodiment with a command value converter (command value conversion means) 24.
The command value converter 24 is provided with the phase command value θ * together with the command value generator 10. Other configurations are the same as in the first embodiment. Here, the phase command value θ * is generated by a phase command value generator (not shown) based on the phase θ of the rotor of the AC motor 9.
A leading component and a lag component are added to the phase θ in accordance with the driving state of. Note that the phase θ may be used as it is in the same manner as in the first embodiment, instead of the phase command value θ *.

Next, the operation of the sixth embodiment will be described with reference to FIGS. The command value generator 10
The phase of the U-phase voltage command value U * is output in synchronization with the change in the phase command value θ *, and the voltage command values V * and W * are delayed by 2π / 3 from the value of the voltage command value U *, respectively. , 2π / 3. Then, the command value converter 24
In step B1 shown in FIG. 17, each phase voltage command value U
The phase command value θ * is read together with *, V *, and W *. In the following steps B2 to B7, it is determined in which section (electrical angle) the phase command value θ * is.

That is, the six sections in steps B2 to B7 are sections in which any two-phase command values of the three-phase command values are substantially equal, and the electrical angles π / 6, π / 2, 5
It is a range centered on π / 6, 7π / 6, 3π / 2, 11π / 6 (see FIG. 18). In addition, each of steps B2 to B
“± φ” in 7 is an electrical angle corresponding to the predetermined value x in the first embodiment.

Then, the command value converter 24 proceeds to step B
If "YES" is determined in step 2, the process proceeds to step B8 (corresponding to step A13), and a new voltage command value U **,
V ** and W ** are converted and generated. Then, step A15
Then, the process proceeds to step B15 in which the same processing as described above is performed, and the voltage command values U **, V **, W ** are output to the PWM waveform generator 11.

Similarly, the command value converter 24 determines in step B
If "YES" is determined in step 3, the process proceeds to step B9 corresponding to step A8, and the voltage command values U **, V **, W **
When the conversion is generated, the process proceeds to step B15. If "YES" is determined in step B4, step A
The process proceeds to Step B10 corresponding to Step 5, and if “YES” is determined in Step B5, the process proceeds to Step B11 corresponding to Step A12, and to “YES” in Step B6.
Is determined, Step B12 corresponding to Step A9 is performed.
Then, if "YES" is determined in the step B7, the process shifts to a step B13 corresponding to the step A4.

In other cases ("NO" in all of steps B2 to B7), after step B14 similar to step A14, the voltage command values U *, V *, W * are directly used as voltage command values U *. Step B1 as *, V **, W **
Move to 5.

As described above, according to the sixth embodiment, the command value converter 24 converts and generates new voltage command values U **, V **, W ** based on the phase command value θ *. Therefore, the same effect as in the first embodiment can be obtained.

(Seventh Embodiment) FIGS. 19 and 20 show a seventh embodiment of the present invention. Only parts different from the second and fifth embodiments will be described. In the configuration of the seventh embodiment shown in FIG. 19, the command value converter 21 in the second embodiment is replaced by a command value converter (command value conversion means) 25. A phase command value θ * is provided similarly to the command value converter 24 of the fifth embodiment.
Other configurations are the same as in the second embodiment.

Next, the operation of the seventh embodiment will be described with reference to FIG. In the seventh embodiment, the processing of the second embodiment is performed based on the phase command value θ * as in the fifth embodiment. That is, as shown in FIG.
5 executes steps A16 to A22 in the same manner as in the case where "YES" is determined in step B7 and "YES" is determined in step A2 of the second embodiment.

Although not specifically shown in the other cases, if "YES" is determined in step B3, steps A25 to A29 are executed similarly to the case where "YES" is determined in step A6. B
If "YES" is determined in step 5, the control proceeds to step A1
Steps A32 to A36 are executed in the same manner as when “YES” is determined in 0. The other cases are the same as in the fifth embodiment.

As described above, according to the seventh embodiment, the command value converter 25 selects one of the three-phase voltage command values U *, V *, and W * based on the phase command value θ *. In a period in which the two-phase voltage command value levels are relatively close, a first conversion period in which the two-phase command value levels are fixed to the maximum or minimum to perform one-phase modulation, and a first conversion period in any one of the two phases Since the command value level is fixed to the maximum or minimum and the second conversion period for performing the two-phase modulation is alternately executed at the same time ratio, the same effect as in the second embodiment can be obtained.

(Eighth Embodiment) FIG. 21 shows an eighth embodiment of the present invention. For example, a well-known device for controlling a brushless motor (polyphase AC motor) used for driving an electric vehicle is known. FIG. 3 is a functional block diagram when the present invention is applied to a control system (current control). That is, the current command value iq
* Is a torque command value given by an ECU (Electronic Control Unit) according to a throttle opening signal, a motor rotation speed, etc., and a current command value id * is a command value i.
A magnetic flux command value read from the table and given according to q *.

The current command values id * and iq * are given to subtracters 26a and 26b as subtracted values, respectively. The current detection values iu, iv, iw are obtained by detecting the output current of the inverter main circuit by a current sensor.
It is provided to the vw / dq converter 27. uvw / d
The −q conversion unit 27 converts (three-phase / two-phase conversion) the current detection values iu, iv, and iw into vector components of dq axes (rotary coordinate axes) based on the phase detection value θ, and obtains a two-phase current value. id, i
q is output to the subtracters 26a and 26b as a subtraction value.

The output signals of the subtracters 26a and 26b are applied to current control units 28a and 28b for performing, for example, PI control, and are converted into voltage command values vd '* and vq' *. The subtractors 29a and the adders 29b Are given as the subtracted value and the added value, respectively. The decoupling unit 30 is uvw / dq
When the back electromotive force generated in the brushless motor is detected based on the two-phase current values id and iq output from the converter 27, the back electromotive force is output to the subtractor 29a and the adder 29b as a subtraction value and an addition value. Then, the subtractor 29a and the adder 2
9b is d as voltage command values vd * and vq *.
-Q / uvw conversion unit 31.

The dq / uvw converter 31 converts the voltage command values vd *, vq * into three-phase voltage command values U *, V *, W * based on the phase detection value θ (two-phase / Three-phase conversion)
The voltage command values U *, V *, W * are output to a command value converter (command value conversion means) 32. Command value converter 3 here
Reference numeral 2 may be a command value conversion means used in any of the first to seventh embodiments. Then, the command value converter 32 converts the voltage command values U *, V *, W * into the voltage command values U **, V **, W
** and outputs it to the PWM pattern generator 11. The subsequent configuration is the same as that of the first embodiment. The configuration of the eighth embodiment can be realized by any of hardware and software.

As described above, according to the eighth embodiment, the present invention can be applied to the case where the current control of a brushless motor is performed.

(Ninth Embodiment) FIGS. 22 and 23 show a ninth embodiment of the present invention.
Only parts different from the embodiment will be described. In the ninth embodiment, as in the first embodiment, the period during which one-phase modulation is performed is determined with reference to the values of the voltage command values U *, V *, and W *. That is, in the flowchart shown in FIG. 22, the command value converter 21 determines the voltage command value U as in step A1.
When *, V *, and W * are read (step C1), the voltage command values U *, V are determined in the subsequent determination steps C2 to C7.
It is determined whether or not the respective relationships are established between * and W *. Step C2: (W *> U *) · (U * ≧ 0) Step C3: (W *> V *) · (W * ≦ 0) Step C4: (U *> V *) · (V * ≧ 0) ) Step C5: (U *> W *) · (U * ≦ 0) Step C6: (V *> W *) · (W * ≧ 0) Step C7: (V *> U *) · (V * ≦ 0)

Here, when the respective conditions are satisfied in each of the determination steps C2 to C7 ("YES"), FIG.
Correspond to the phase periods に shown in the pattern of the voltage command values U *, V *, and W *, and each phase period と is a period shown below based on the phase θ of the voltage command value U *. I have.

Phase period: 0 <θ <π / 6 Phase period: π / 3 <θ <π / 2 Phase period: 2π / 3 <θ <5π / 6 Phase period: π <θ <7π / 6 Phase period: 4π / 3 <θ <3π / 2 Phase period: 5π / 3 <θ <11π / 6

If the respective conditions are satisfied in each of the determination steps C2 to C7 ("YES"), steps C8 to B13 having the same contents as steps B8 to B13 of the sixth embodiment are performed.
The process proceeds to C13. Thereafter, if the respective conditions are not satisfied in any of the determination steps C2 to C7 ("NO"), step C14 having the same contents as step B14 is executed. It should be noted that “Va” shown in the equations showing the phase voltage command values U *, V *, W * shown in FIG.
This is a parameter for changing the amplitude level of the voltage command value.

As a result of the above-described processing, the period in which two phases of the three-phase voltage command values U *, V *, and W * are fixed and only one phase is subjected to PWM modulation is equal to that of the first embodiment. Π for the period
/ 12 (15 °). In this case, the distortion of the output current waveform is slightly increased.
It can be applied to an application in which there is no problem even if the vibration or the like is somewhat large.

Further, assuming that the present invention is applied to an electric vehicle, for example, when the electric vehicle runs at a relatively low speed, if the distortion of the output current waveform increases, the vibration level of the AC motor 9 increases and the ride comfort of the occupant increases. Has a significant effect. However, when the electric vehicle runs at a relatively high speed, even if the distortion of the output current waveform is slightly larger,
The vibration of the AC motor 9 based on the distortion does not significantly affect the riding comfort of the occupant. In the ninth embodiment, the period during which the one-phase modulation is performed can be determined in a short time based only on the magnitude relationship between the voltage command values U *, V *, and W *. It is extremely effective to perform in high areas.

(Tenth Embodiment) FIG. 24 shows a tenth embodiment of the present invention.
It shows an embodiment. The tenth embodiment is an example in which the seventh embodiment and the ninth embodiment are combined. That is, for the portion shown in FIG. 20 in the seventh embodiment, steps C7 and C7 'are arranged instead of steps B7 and A16. Step C7 'is a step for determining whether or not the following condition is satisfied. (V * ≦ U *) · (U * <0) If “YES” is determined in the step C7, the process shifts to the step A17, and “N” is determined in the step C7.
If "O" and "YES" in step C7 ', the process proceeds to step A20. In step C7 ', "N
If it is determined to be "O", the process proceeds to step B14 (C14).

Further, in the seventh embodiment, the other parts which replace the second embodiment as shown in FIG. 20 can be implemented by replacing the phase voltage command values in the same manner as shown in FIG. good.

The present invention is not limited to the embodiment described above and shown in the drawings, and the following modifications or extensions are possible. In the sixth embodiment, the command value output means and the command value conversion means may be configured integrally. In the eighth embodiment, the decoupling unit 30 may be provided as needed. Further, the present invention is not limited to the current control of the eighth embodiment, but may be applied to a speed control, a position control, or a control method completely different from that for a brushless motor as long as it finally outputs a voltage command value. In the ninth embodiment, the phase periods may be shifted in the π / 6 delay direction. Further, similarly to the sixth embodiment, the phase command value θ
* Or the phase θ may be read, and a similar phase period may be determined according to the value.

The voltage command value does not necessarily have to be based on the change in the amplitude of the sine wave, but may be another voltage waveform. In each of the above embodiments, the description has been made of the triangular wave comparison PWM. However, the modulation method is not limited to the method based on such a carrier wave, and may be any method that converts power by changing a pulse width by switching. Any other modulation method may be used. In the case of digital control, shortage of line voltage due to fixing two phases is stored in a memory, and
It may be superimposed in real time during the period in which only the phase is fixed. The switching element is not limited to the IGBT, but may be, for example, a power transistor or a power MOSFET. The position information of the rotor of the motor 9 is not limited to the one that obtains the position information of the rotor of the motor 9 using a current sensor, a Hall IC, a rotary encoder, a resolver, or the like. A so-called sensorless driving method may be adopted.

It is not limited to n = 1 and 2, and n may be appropriately set to a positive real number. The same applies to m, and m and n may be appropriately set according to the situation such as loss and distortion. The configuration centering on the command value conversion means is a DSP
However, the present invention is not limited to the configuration using a CPU, and may be configured using a CPU. For example, for a three-phase AC motor in which only one phase is fixed and a two-phase modulation method in which modulation is performed only in two phases is always performed, one phase is further fixed from that state, thereby implementing each of the above-described embodiments. As in the example, two phases may be fixed for a predetermined period. In this case, the number of times of switching can be further reduced. The present invention may be applied to a four-phase or more polyphase AC motor. The present invention is not limited to a polyphase AC motor used for an electric vehicle or a hybrid electric vehicle, but may be applied to any device that outputs a polyphase line voltage such as a UPS (Uninterruptible Power Supply).

[Brief description of the drawings]

FIG. 1 is a functional block diagram showing an electric configuration of a power converter according to a first embodiment of the present invention.

FIG. 2 is a flowchart showing control contents of a command value converter.

3 (a) to 3 (c) are voltage command values U *, V *, W *
(D) to (f) indicate voltage command values U **, V **, W **
Show

4A is a diagram showing voltage command values U *, V * and W * superimposed, FIG. 4B is a diagram showing a part of FIG. Command values U **, V **,
Diagram showing W **

FIG. 5 is a second embodiment of the present invention, and is an electrical configuration diagram of a main part.

6 (a) shows a carrier wave of PWM modulation, FIG. 6 (b) shows a clock signal ck1 synchronized with the carrier wave, and FIG. 6 (c) shows a clock signal ck2 delayed by 90 ° from the clock signal ck1.

FIG. 7 is a diagram corresponding to FIG. 2 of a main part (part 1);

FIG. 8 is a diagram corresponding to FIG. 2 of a main part (part 2);

FIG. 9 is a view corresponding to FIG. 2 of a main part (part 3).

FIG. 10 is a diagram corresponding to FIG.

FIG. 11 is a diagram corresponding to FIG. 3;

12A shows a simulation result of a phase current waveform according to the first embodiment, and FIG. 12B shows a simulation result of a phase current waveform according to the second embodiment.

FIG. 13 is a view corresponding to FIG. 4C showing a third embodiment of the present invention.

FIG. 14 is a view corresponding to FIG. 4C showing a fourth embodiment of the present invention.

FIG. 15 is a view corresponding to FIG. 4C showing a fifth embodiment of the present invention.

FIG. 16 is a view corresponding to FIG. 1, showing a sixth embodiment of the present invention;

FIG. 17 is a diagram corresponding to FIG. 2;

FIG. 18 is a diagram corresponding to FIG. 4;

FIG. 19 is a view corresponding to FIG. 1, showing a seventh embodiment of the present invention.

FIG. 20 is a view corresponding to FIG. 17 showing a main part.

FIG. 21 is an eighth embodiment of the present invention, and is a functional block diagram in a case where the present invention is applied to a control system for driving and controlling a brushless motor used for driving an electric vehicle.

FIG. 22 is a view corresponding to FIG. 1, showing a ninth embodiment of the present invention;

FIG. 23 is a diagram corresponding to FIG. 17;

FIG. 24 is a view corresponding to FIG. 20, showing a tenth embodiment of the present invention;

FIG. 25 is a diagram corresponding to FIG. 1 showing a conventional technique.

FIG. 26 is a diagram corresponding to FIG. 5;

FIG. 27 is a view corresponding to FIGS.

[Explanation of symbols]

1 to 6 are IGBTs (switching elements), 7 is an inverter main circuit, 9 is a polyphase AC motor, 10 is a command value generator (command value output means), 11 is a PWM waveform generator (control signal output means), 21 Is a command value converter (command value conversion means), 22 is a PWM waveform generator (control signal output means),
Reference numerals 23, 24, 25 and 32 denote command value converters (command value converters).

 ──────────────────────────────────────────────────続 き Continued on the front page F term (reference) LL41 LL58

Claims (28)

[Claims] 1. An inverter main circuit for applying a phase voltage to a polyphase load via a switching element, command value output means for outputting a voltage command value for each phase, and a voltage output by the command value output means. Command value conversion means for converting and outputting such that at least one of the two-phase voltage command values becomes equal to the maximum or the minimum during a period in which at least one of the two-phase voltage command values is substantially equal; And a control signal output means for outputting a switching control signal to the inverter main circuit by pulse width modulating the carrier with the voltage command value output by the command value conversion means. apparatus. 2. The method according to claim 1, wherein the command value conversion means outputs a voltage command value for a phase not to be converted as an average value of line voltages of the voltage command values of the phases to be converted. The power converter according to claim 1. 3. The power converter according to claim 1, wherein the polyphase load is a polyphase AC motor. 4. The power converter according to claim 3, wherein the number of phases of the polyphase AC motor is three. 5. An inverter main circuit for applying a phase voltage to a polyphase load via a switching element, command value output means for outputting a voltage command value for each phase, and a voltage output by the command value output means. A first conversion period in which at least one of the two-phase voltage command values is substantially equal to each other during the period in which the at least one of the two-phase voltage command values is substantially the same; Command value conversion means for outputting a second conversion period for converting only the one-phase voltage command value to a maximum or a minimum, and a voltage command value output by the command value conversion means. A power converter, comprising: control signal output means for outputting a switching control signal to the inverter main circuit by pulse width modulating a carrier. 6. The power conversion device according to claim 5, wherein said command value conversion means sets said conversion period to an integral multiple of said carrier wave period. 7. The command value converting means sets a phase to be converted in the second conversion period to a phase in which a line voltage with a voltage command value for a phase other than the conversion target is the maximum. The power converter according to claim 5 or 6, wherein 8. The command value conversion means sets the second conversion period to n times (n is a positive real number) times the first conversion period, and in the first conversion period, a phase other than a phase to be converted is set. Is shifted in the second conversion period so as to maintain the line voltage with the voltage command value of the phase to be converted in the second conversion period. In the second conversion period, only during the first conversion period The voltage command value for the phase to be converted is
The power converter according to any one of claims 5 to 7, wherein a level shift is performed so that a line voltage with a voltage command value of a phase to be converted becomes (1 + 1 / n) times in a conversion period. . 9. The power converter according to claim 8, wherein said command value converter sets n = 1. 10. The power converter according to claim 5, wherein the command value converter changes a ratio between the first conversion period and the second conversion period. 11. The method according to claim 1, wherein the command value conversion unit outputs the second conversion period as the difference between the voltage command values before conversion to be converted in the first conversion period, which is output by the command value output unit, increases. The power converter according to claim 10, wherein is set relatively long. 12. The power converter according to claim 5, wherein the number of phases of the polyphase load is three. 13. An inverter main circuit for applying a phase voltage to a polyphase load via a switching element, command value output means for outputting a voltage command value for each phase, and a voltage output by the command value output means. Based on the magnitude relationship between the command values, conversion is performed such that at least one of the two-phase voltage command values is equal to the maximum or the minimum during a specific period including a time point when at least one of the two-phase voltage command values is equal. Command value converting means for outputting a switching control signal to the inverter main circuit by pulse width modulating the carrier with the voltage command value output from the command value converting means. A power converter, comprising: 14. The power converter according to claim 1, wherein the command value output unit outputs a voltage command value based on a change in amplitude of a sine wave. 15. A voltage command value is output for each phase with respect to a polyphase load, and at least one of the voltage command values is output during a period in which at least any two-phase voltage command values are substantially equal. Inverter main circuit that converts a two-phase voltage command value so that it becomes equal to maximum or minimum, and that applies a phase voltage to the polyphase load via a switching element by performing pulse width modulation on a carrier with the voltage command value. And a switching control signal. 16. The multi-phase device according to claim 15, wherein a voltage command value for a phase other than the conversion target is output as an average value of line voltages with the voltage command values of the phase to be converted. Load drive control method. 17. The drive control method for a polyphase load according to claim 15, wherein the polyphase load is a polyphase AC motor. 18. The drive control method for a polyphase load according to claim 17, wherein the number of phases of the polyphase AC motor is three. 19. A voltage command value is output for each phase with respect to a multi-phase load, and at least one of the two or more phase command values is substantially equal during the period when the voltage command values are substantially equal. Output is performed so that a first conversion period for converting the voltage command values of the phases to be equal and constant and a second conversion period for converting only the voltage command value of one phase thereof to be constant are alternately repeated. Modulating a carrier with the voltage command value to output a switching control signal to an inverter main circuit that applies a phase voltage to the polyphase load via a switching element; Method. 20. The drive control method for a polyphase load according to claim 19, wherein the conversion period is set to an integral multiple of the carrier wave period. 21. The method according to claim 19, wherein a phase to be converted in the second conversion period is set to a phase in which a line voltage with a voltage command value of a phase other than the conversion target is maximized. 21. The drive control method for a polyphase load according to claim 20. 22. The second conversion period is set to be n times (n is a positive real number) times the first conversion period. In the first conversion period, a voltage command value for a phase other than a conversion target is In the second conversion period, the level is shifted so as to maintain the line voltage with the voltage command value of the phase to be converted. The voltage command value is
22. The multi-phase load according to claim 19, wherein a level shift is performed so that a line voltage with a voltage command value of a phase to be converted becomes (1 + 1 / n) times in a conversion period. Drive control method. 23. The method according to claim 22, wherein n = 1 is set. 24. The drive control of a multi-phase load according to claim 19, wherein a ratio between the first conversion period and the second conversion period is changed for two phases to be converted. Method. 25. The multi-phase load according to claim 24, wherein the second conversion period is set relatively long as the difference between the voltage command values before conversion for the two phases to be converted becomes large. Drive control method. 26. The driving control method for a polyphase load according to claim 19, wherein the number of phases of the polyphase load is three. A voltage command value corresponding to each phase of the polyphase load is output, and a specific value including a time point at which at least one of the two phase voltage command values becomes equal is determined based on a magnitude relationship between the voltage command values. During the period, the voltage command value of at least one of the two phases is converted so as to be equal to the maximum or the minimum, and the carrier is pulse-width-modulated by the voltage command value, so that the multi-phase load is phase-changeable via a switching element. A driving control method for a polyphase load, comprising: outputting a switching control signal to an inverter main circuit that applies a voltage. 28. The drive control method for a multi-phase load according to claim 15, wherein said command value output means outputs a voltage command value based on a change in amplitude of a sine wave.