JP2001037282A - Electric controller for multiphase ac motor - Google Patents

Abstract

PROBLEM TO BE SOLVED: To control the operating state even of an unbalanced multiphase AC motor to desired characteristics. SOLUTION: In an electric controller for controlling a brushless motor depending on three-phase command voltage values Vu0*, Vv0*, Vw0* calculated based on control parameters, a feed forward compensating section 32 compensates for the unbalanced components of the motor by correcting the three-phase command voltage values Vu0*, Vv0*, Vw0*. In the feed forward compensating section 32, a compensation voltage waveform memory 63, an interpolating section 64 and a three-phase output section 65 calculate compensation voltage values ΔVunu, ΔVunv, ΔVunw for respective phases of the brushless motor which vary depending on the currents Iu, Iv, Iw detected by a current sensor and at an operating section. Operating sections 51-53 add the calculated compensation voltage values ΔVunu, ΔVunv, ΔVunw, respectively, to the command voltage values Vu0*, Vv0*, Vw0*.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to an electric control device for a polyphase AC motor.

[0002]

2. Description of the Related Art Conventionally, for example, Japanese Patent Application Laid-Open No. 7-10777
As shown in Japanese Patent Publication No. 2 (2003), a command voltage value of each phase is calculated according to an operation control parameter for controlling an operation state of the polyphase AC motor, and the command voltage value of each phase is calculated according to the calculated command voltage value of each phase. It is well known to control the operating state of a polyphase AC motor.

[0003]

However, in the above-mentioned conventional electric control device, it is premised that the polyphase AC motor is in a balanced state.
JP 07772 does not mention the control of unbalanced polyphase AC motors. Many multi-phase AC motors are usually designed so that they are electrically balanced, that is, the voltage and current waveforms of each phase are sinusoidal with a uniform phase shift, and their sums are always “0”. ing. The premise that this polyphase AC motor has electrically balanced characteristics is:
The simplification of the motor structure, the degree of freedom in the shape of parts, and the limited choice of materials have reduced the cost of manufacturing multi-phase AC motors, which may increase the voltage and current waveforms of each phase. Even if a conventional control method is adopted for a polyphase AC motor that is not a sine wave because it contains an unbalanced component, that is, a so-called unbalanced polyphase AC motor, the operation state of the polyphase AC motor can be changed to a desired one. Cannot be controlled by characteristics.

[0004]

SUMMARY OF THE INVENTION The present invention has been made to address the above problems, and has as its object to control the operating state of an unbalanced polyphase AC motor to desired characteristics. It is intended to provide an electric control device for a polyphase AC motor capable of performing the following.

In order to achieve the above object, a structural feature of the present invention is that a command voltage value of each phase is calculated according to operation control parameters for controlling an operation state of a polyphase AC motor, and the calculated values are calculated. In an electric control device for a polyphase AC motor for controlling an operation state of a polyphase AC motor according to a command voltage value of each phase, an operation state detecting means for detecting an operation state of the polyphase AC motor; Compensation voltage value calculation means for calculating a compensation voltage value for each phase for compensating for unbalance based on the detected operating state; and a compensation voltage value for each phase calculated as a command voltage value for each phase. Correction means for correcting the command voltage value of each phase in consideration of the above is provided.

[0006] In this case, the compensation voltage value calculating means associates, for example, data representing a compensation voltage waveform corresponding to a voltage unbalance component for one phase of the polyphase AC motor with the operating state of the same polyphase AC motor. And a compensating voltage value for each phase based on the data read from the memory means in accordance with the detected operating state.

In addition, the compensation voltage value calculating means associates data representing an unbalanced voltage waveform for one phase including a voltage unbalance component of the polyphase AC motor with the operating state of the polyphase AC motor. One phase is calculated by calculating a difference between a pre-stored storage means and an unbalanced voltage waveform represented by data read from the storage means in accordance with the detected operating state, and a fundamental component of the unbalanced voltage waveform. A compensating voltage waveform calculating means for calculating a compensating voltage waveform corresponding to the minute voltage unbalance component, wherein the compensating voltage value of each phase is calculated based on the calculated compensating voltage waveform.

According to the present invention, the command voltage value for each phase of the polyphase AC motor is changed by the operation of the operating state detecting means, the compensation voltage value calculating means and the correcting means. Is corrected to compensate for the imbalance of the motor. In this case, the command voltage value of each phase is calculated according to an operation control parameter for controlling the operation state of the polyphase AC motor, and controls the operation state of the polyphase AC motor. Even if the motor is configured to be unbalanced, the unbalanced components of each phase of the polyphase AC motor in all operating states are compensated to always obtain desired control characteristics according to the operation control parameters. This means that unbalanced polyphase AC motors can be used in various controls, so in the production of polyphase AC motors, simplification of the motor structure, freedom of component shapes, material This increases the degree of freedom in design, such as the room for selection, and in turn leads to lower manufacturing costs for the polyphase AC motor.

[0009]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS One embodiment of the present invention will be described below with reference to the drawings. FIG. 1 shows an example in which an electric control device for a polyphase AC motor according to the present invention is applied to an electric power steering device for a vehicle. Is shown in a block diagram.

This electric power steering apparatus includes a brushless motor 11 composed of a three-phase synchronous permanent magnet motor as a polyphase AC motor. The brushless motor 11 applies an assisting force to the steering of the front wheel by the turning operation of the steering handle 12, and drives the tie rod 13 connecting the front wheel at the outer end in the axial direction according to the rotation. Steering shaft 1 connected to steering wheel 12 at the upper end and connected to tie rod 13 at the lower end
4, a steering torque sensor 15 is assembled, and the sensor 15 detects a steering torque acting on the steering shaft 14 and outputs a detection signal representing the torque. The brushless motor 11 includes a rotation angle sensor 16 configured by an encoder for detecting the rotation angle of the motor 11.
Is assembled. In the case of the rotation angle sensor 16 using an encoder, a two-phase pulse train signal having a phase different by π / 2 according to the rotation of the rotor of the brushless motor 11 and a zero-phase pulse train signal representing the reference rotation position are output. In addition,
Since the rotation angle sensor 16 detects a mechanical rotation angle of the rotor of the brushless motor 11 with respect to the stator, various rotation angle sensors other than the encoder can be used.

The electric control device for controlling the rotation of the brushless motor 11 includes a basic assist force calculation unit 21, a return force calculation unit 22, and a calculation unit 23 for calculating the command torque T *. The basic assist force calculation unit 21 receives the steering torque from the steering torque sensor 15 and the vehicle speed from a vehicle speed sensor (not shown), and calculates an assist torque that increases as the steering torque increases and decreases as the vehicle speed increases. The return force calculation unit 22 inputs an electric angle θe (corresponding to a rotation angle) and an angular speed ω of the rotor described later together with the vehicle speed, and based on these input values, a return force of the steering shaft 14 to the basic position and the same. The return torque corresponding to the resistance to the rotation of the steering shaft 14 is calculated. The calculation unit 23 calculates the command torque T * by adding the assist torque and the return torque, and supplies the command torque T * to the command current value determination unit 24.

The command current value deciding section 24 has a command torque T *
The command field current value (d-axis command current value) Id * and the command torque current value (q-axis command current value) Iq * are calculated based on The two current values Id * and Iq * correspond to the d-axis in the same direction as the permanent magnet and the q-axis orthogonal to the same in the rotating coordinate system synchronized with the rotating magnetic flux generated by the permanent magnet on the rotor of the brushless motor 11. The command field current value Id * designates the magnitude of the field current of the brushless motor 11, and the command torque current value Iq * designates the magnitude of the torque generated by the motor 11. The command current value determination unit 24 also receives values detected by various sensors, corrects both current values Id * and Iq *, and outputs the corrected values. For example, when the battery voltage value is input, the two current values Id * and Iq * are corrected for field weakening control when the battery voltage value is low.

The corrected command field current value Id * and command torque current value Iq * are supplied to calculation units 25 and 26, which calculate the command field current value Id * and the command torque current value. The difference values ΔId and ΔIq are calculated by subtracting the detected field current value (d-axis detected current value) Id and the detected torque current value (q-axis detected current value) Iq from Iq *, respectively. (PI control unit) 27, 28. The proportional-integral control units 27 and 28 operate based on the difference values ΔId and ΔIq such that the detected field current value Id and the detected torque current value Iq follow the command field current value Id * and the command torque current value Iq *, respectively. Then, a command field voltage value (d-axis command voltage value) Vd * and a command torque voltage value (q-axis command voltage value) Vq * respectively corresponding to the d-axis and the q-axis are calculated. The command torque T *, the command field current value Id *, and the command torque current value Iq * correspond to operation control parameters.

The command field voltage value Vd * and the command torque voltage value Vq * are supplied to a two-phase / three-phase coordinate converter 31. The two-phase / three-phase coordinate converter 31 outputs the two voltage values Vd *, Vq *
Are converted into three-phase command voltage values Vu0 *, Vv0 *, Vw0 * and supplied to the feedforward compensation unit 32. The feedforward compensator 32 corrects the three-phase command voltage values Vu0 *, Vv0 *, and Vw0 * for various compensations described later in detail,
The corrected three-phase command voltage values Vu1 *, Vv1 *, Vw1 * are expressed as P
It is supplied to the WM voltage generator 33. PWM voltage generator 33
Outputs the PWM control voltage signals UU, VU, WU corresponding to the three-phase command voltage values Vu1 *, Vv1 *, Vw1 * to the inverter circuit 34.

As shown in FIG. 2, the inverter circuit 34 is switching-controlled by PWM control voltage signals UU, VU, WU, and switches three sets of IGBT elements, MOS-FETs and the like connected in series for each phase. Power elements) elements Tr1 to Tr6 and each switching element Tr1 to T
r6, and three-phase voltage signals Vu, Vv, Vw corresponding to the PWM control voltage signals UU, VU, WU, and generate the same voltage signals Vu, Vw.
v and Vw are respectively applied to the respective phases of the brushless motor 11 via three-phase current paths.

Two of the three-phase current paths are provided with current sensors 35 and 36, respectively.
Represents two of the three-phase currents for the brushless motor 11.
Detecting two currents (the u phase and the v phase in this embodiment),
The detected current values Iu and Iw representing the two currents are supplied to a three-phase / two-phase coordinate converter 37. This three-phase / two-phase coordinate converter 3
7 is also supplied with a detected current value Iv that represents the current of another phase (w-phase in the present embodiment) calculated based on the detected current values Iu and Iw by the arithmetic unit 38. 3 phase / 2
The phase coordinate converter 37 outputs the detected current values Iu, Iu,
Iv and Iw are two-phase detected field current values (d-axis detected current values) Id
And a detected torque current value (q-axis detected current value) Iq.

A rotation angle sensor 16 using an encoder
Is used, the two-phase pulse train signal and the zero-phase pulse train signal from the rotation angle sensor 16 are supplied to the electrical angle converter 41. The electrical angle converter 41 calculates an electrical angle θe of the rotor of the brushless motor 11 with respect to the stator based on the pulse train signals, and supplies the electrical angle θe to the angular velocity converter 42. The angular velocity converter 42 calculates the angular velocity ω of the rotor with respect to the stator by differentiating the electrical angle θe. The detected current values Iu, Iv, Iw, the detected field current value Id, the detected torque current value Iq, the electrical angle θe, and the angular velocity ω correspond to operating state parameters indicating the operating state of the brushless motor 11.

Next, the feedforward compensator 32 will be described in detail. This feedforward compensator 3
2 is an unbalanced voltage compensation value calculator 32, as shown in FIG.
A, a quantization error compensation value calculator 32B, a voltage drop compensation value calculator 32C, and a pulse width error compensation value calculator 32D;
Command voltage values Vu0 *, Vv0 from phase / 3-phase coordinate conversion unit 31
*, Vw0 *, calculation units 51 to 51 for adding the respective compensation voltage values from the respective compensation value calculation units 32A to 32D for each phase.
62, and output the corrected command voltage values Vu1 *, Vv1 *, Vw1 * from the calculation units 51 to 53.
Note that at least the operations in the operation units 51 to 53 are performed with a large number of bits (16 bits in the present embodiment), and a plurality of higher-order bits (in this embodiment, of the total number of bits of the operation outputs of the operation units 51 to 53). Only 8 bits) are supplied to the PWM voltage generator 33 as the command voltage values Vu1 *, Vv1 *, Vw1 *.

First, the unbalanced voltage compensation value calculation unit 32A will be described. The compensation value calculation unit 32A calculates compensation voltage values ΔVunu, ΔVunv, ΔVunw of each phase for compensating for unbalance of the brushless motor 11. 11, the compensation voltage waveform memory 63,
It comprises an interpolation section 64 and a three-phase output section 65. The compensation voltage waveform memory 63 stores waveform sampling data representing a compensation voltage waveform corresponding to a voltage imbalance component for one phase of the brushless motor 11 in accordance with the operation state of the motor 11.

To explain the compensation voltage waveform, if the brushless motor 11 is configured to be electrically balanced, the voltage Eu, Ev, Ew and current ( The line currents Iu, Iv, and Iw have sinusoidal waveforms as shown in FIGS. With such a balanced brushless motor 11, even if the unbalanced voltage compensation value calculation unit 32A is not provided, the motor 11 can
Command torque T * (command field current value Id *, command torque current value Iq *, command field voltage value Vd *, and command torque voltage value Vq
*) That is, the operation state is accurately controlled according to the operation control parameters.

However, if the brushless motor 11 is configured to be electrically unbalanced, the voltages Eu, Ev, Ew and the currents (line currents) Iu, Iv, Iw of each phase of the motor 11 are as shown in FIG. As shown in (C) and (D), the waveform becomes a non-sinusoidal waveform including an unbalanced component. Therefore, such an unbalanced brushless motor 11 is capable of controlling the voltages Eu, Ev, Ew and the currents Iu, Iw of each phase.
Unless the unbalanced components of v and Iw are compensated, accurate control is not performed according to the operation control parameters. Conversely, if compensation voltages for eliminating the unbalanced components are given to the command voltages Vu0 *, Vv0 *, Vw0 * for the respective phases, the currents (line currents) Iu, Iv, Iw will be as shown in FIG. As shown in B), the current becomes a sinusoidal balanced three-phase current that does not include an unbalanced component, and the operation state of the brushless motor 11 is accurately controlled according to the operation control parameters.

Focusing on such a point, the compensation voltage waveform memory 63 stores waveform sampling data representing a compensation voltage waveform obtained by making the unbalanced component waveform for one phase the opposite phase.
However, this unbalanced component waveform is
It changes according to the operating state such as the rotation speed, generated torque, field current, and the like. Therefore, in the compensation voltage waveform memory 63, the unbalanced component waveform of one phase of the brushless motor 11 in the state where the rotation speed, the generated torque, the field current, and the like are variously set, that is, in various operating states, is reversed ( Waveform sampling data representing waveforms having the same phase and the opposite phase are stored as compensation voltage waveform data.

Next, a method for creating data representing the compensation voltage waveform will be described. A motor having the same configuration as the brushless motor 11 is experimentally operated in advance under various operating conditions to measure a voltage waveform. That is, while independently changing the angular velocity ω of the motor, the field current value (d-axis current value) Id and the torque current value (q-axis current value) Iq for the motor, each voltage waveform for one phase in each change state ν
= Fm (θe, ω, Id, Iq) are sequentially measured (see the solid line in the left waveform in FIG. 5). Note that θe indicates the electric angle of the motor as in the case described above. In addition, the voltage waveform ν
, The respective voltage waveforms ν = fm (θe, ω, Id, Iq) may be sequentially obtained by simulation. In this case, the voltage waveforms ν of the respective phases are generally substantially equal except for the phase difference, and the voltage waveform ν of one phase may be obtained by measurement or simulation. In order to further improve the accuracy, a three-phase voltage waveform ν
May be measured, and the average value of the measured three-phase voltage waveforms ν may be obtained.

Next, by a frequency analysis process such as Fourier transform, each fundamental wave component ν1 = fm1 (θe, Δθe, | ν1 |) = | ν1 | · sin (θe +) of each voltage waveform ν in various operating states.
Δθe) is calculated (see the broken line in the left waveform in FIG. 5). Note that the fundamental wave component ν1 of the voltage waveform ν refers to a sine waveform having the same frequency as the voltage waveform ν, and Δθe is the phase difference of the fundamental wave component ν1 with respect to the voltage waveform ν (it can be ignored because it is a small value) And | ν1 | is the amplitude of the fundamental wave component ν1.

Next, the unbalanced component waveform Δ for one phase is calculated by the following equation 1 in which the fundamental component ν1 is subtracted from the voltage waveform ν.
νun is calculated (see the solid line in the waveform on the right side in FIG. 5).

[0026]

Equation 1 Δνun = ν−ν1 = fm (θe, ω, Id, Iq) −fm1 (θe, Δθe, | ν1 |) = fun (θe, ω, Id, Iq)

This unbalanced component waveform fun (θe, ω, Id, Iq)
Indicates an unbalanced component waveform for one phase obtained by measurement or simulation while variously changing the angular velocity ω, the field current value Id, and the torque current value Iq. The compensation voltage waveform −fun (θe, ω, Id, Iq) obtained by inverting (positive / negative inversion) the various unbalanced component waveforms fun (θe, ω, Id, Iq) is stored in the compensation voltage waveform memory 63. The angular velocity ω, the field current value Id, and the torque current value Iq are stored in association with various values. However, the stored plurality of compensation voltage waveforms −fun (θe,
ω, Id, Iq) do not correspond to all the operating states of the brushless motor 11, but include the angular velocity ω and the field current value Id.
And a typical operation state of the brushless motor 11 in which a plurality of typical values of the torque current value Iq are appropriately combined.

In the above method, the fundamental wave component ν1 is subtracted from the voltage waveform ν to calculate the unbalanced component waveform Δνun, but all harmonics other than the fundamental wave component ν1 obtained by the Fourier transform processing are calculated. Wave component waveforms are added and synthesized, and the value obtained by the addition and synthesis is used as an unbalanced component waveform Δνun = fun
(θe, ω, Id, Iq).

Returning to the description of FIG. 3 again, the compensation voltage waveform memory 63 storing the compensation voltage waveform −fun (θe, ω, Id, Iq) as described above stores The electrical angle θe is supplied, and the angular velocity ω, the detected field current value Id, and the detected torque current value Iq from the angular velocity converter 42 and the three-phase / two-phase coordinate converter 37 are respectively included in the plurality of bit signals. The higher-order multi-bit signals are supplied, and the compensation voltage waveform −fun (θe, ω, Id, Iq) is read using these supplied signals as addresses.

The read compensation voltage waveform -fun (θe,
ω, Id, Iq) are supplied to the interpolation unit 64. Interpolator 64
Are supplied with the remaining lower-order plural bit signals supplied to the compensation voltage waveform memory 63 among the plural bit signals respectively representing the angular velocity ω, the detected field current value Id, and the detected torque current value Iq. The interpolation unit 64 performs an interpolation operation on the compensation voltage waveform read from the compensation voltage waveform memory 63 using the lower-order plural bit signals. As a result, the compensation voltage waveform −fun (θe, ω, I) in the compensation voltage waveform memory 63 in a plurality of typical operating states of the brushless motor 11 is stored.
d, Iq), the compensation voltage waveform −fun (θe, ω, Id, I
q) can be obtained.

The compensation voltage waveform −fun (θe, ω, Id, Iq) interpolated by the interpolation unit 64 is supplied to the three-phase output unit 65. The three-phase output unit 65 outputs the three-phase unbalanced voltage compensation values ΔVunu, ΔVunv, ΔVun shown in Equations 2 to 4 based on the supplied compensation voltage waveform −fun (θe, ω, Id, Iq).
By calculating w, the unbalanced voltage compensation values ΔVunu, ΔVunv,
ΔVunw is calculated by the operation units 51 to 53 via the operation units 54 to 62
To supply.

[0032]

## EQU2 ## ΔVunu = −fun (θe, ω, Id, Iq)

[0033]

ΔVunv = −fun (θe−2π / 3, ω, Id, Iq)

[0034]

ΔVunw = −fun (θe + 2π / 3, ω, Id, Iq)

The operation units 51 to 53 output the command voltage values Vuo *, Vvo *,
Unbalanced voltage compensation values ΔVunu, ΔVunv, ΔVunw are added to Vwo *. Therefore, even in any operating state, the unbalance components fun (θe, ω, Id, Iq) and fun (θe−2π / 3, ω, I) of each phase due to the unbalance of the brushless motor 11 are obtained.
d, Iq) and fun (θe + 2π / 3, ω, Id, Iq) are compensated.

Next, the quantization error compensation value calculation section 32B will be described.
-Phase command voltage value Vu1 * + Vu calculated and output at steps -53
2 *, Vv1 * + Vv2 *, lower bit V of Vw1 * + Vw2 *
Upper bits Vu1 *, Vv1 obtained by truncating u2 *, Vv2 *, Vw2 *
*, Vw1 * compensates for the error caused by being supplied to the PWM voltage generator 33, and includes a delay unit 66. The delay unit 66 determines whether the supplied lower bits Vu2 *,
Vv2 * and Vw2 * are delayed by one cycle of various operations, and supplied to the operation units 51 to 53 via the operation units 54 to 56, and
The three-phase command voltage value Vu0 from the three-phase / three-phase coordinate conversion unit 31
*, Vv0 *, Vw0 *.

As a result, the lower bits Vu2 * ', Vv2 *', Vw2 * 'which have been truncated among the previously calculated three-phase command voltage values Vu1 * + Vu2 *, Vv1 * + Vv2 *, Vw1 * + Vw2 * are
The three-phase command voltage values Vu1 * + Vu2 *, Vv1 * calculated this time
+ Vv2 *, Vw1 * + Vw2 *, so that the three-phase command voltage values Vu1 *, Vv1 supplied to the PWM voltage generator 33
*, Vw1 * are compensated for errors due to the quantization operation processing (lower bit truncation processing).

Next, the voltage drop compensation value calculator 32C will be described. The compensation value calculator 32C compensates for the forward voltage drop in the switching elements Tr1 to Tr6 and the return diodes Di1 to Di6 in the inverter circuit 34. It comprises a forward voltage drop calculator 67 between C and E, a diode forward voltage drop calculator 68, and a combining calculator 69.

The C-E forward voltage drop calculator 67 calculates the switching element T that changes according to the current value (passing current value) I.
One phase of r1 to Tr6 (switching elements Tr1, Tr2, switching elements Tr3, Tr4 and switching elements Tr5, Tr6
) Are stored in advance.
The diode forward voltage drop calculator 68 calculates one phase of the return diodes Di1 to Di6 (the return diodes Di1, Di2, the return diodes Di3,
The forward voltage drop Vd (I) of each set of Di4 and freewheel diodes Di5 and Di6) is stored in advance. The CE forward voltage drop calculator 67 and the diode forward voltage drop calculator 68 have 3
The phase detection current values Iu, Iv, Iw are supplied, respectively, and the CE forward voltage drop calculator 67 calculates the three-phase forward voltage drops Vce (Iu), Vce (Iv), Vce (Iw). The diode forward voltage drop calculator 68 outputs a three-phase forward voltage drop Vd (I
u), Vd (Iv) and Vd (Iw).

In this case as well, a small number of forward voltage drops Vce (I) and Vd with respect to the current I representative of the CE forward voltage drop calculating section 67 and the diode forward voltage drop calculating section 68.
(I) is stored, and the forward voltage drop amounts Vce (Iu), Vce (Iv), Vce ( Iw),
Vd (Iu), Vd (Iv), and Vd (Iw) are read out, and the readout is performed using lower-order plural-bit signals excluding the higher-order bit among the plural-bit signals representing the same current values Iu, Iv, and Iw. Forward voltage drops Vce (Iu), Vce (Iv), Vce (Iw),
Vd (Iu), Vd (Iv) and Vd (Iw) are preferably interpolated and output.

Here, the forward voltage drops Vce (I), Vd
A brief description of (I) will be added. First, in order to measure the forward voltage drops Vce (I) and Vd (I), only a circuit portion for one phase in the inverter circuit 34 is extracted as shown in FIG.
A circuit as shown in (D) is assembled for measurement. The switching elements Tr11 and Tr12 in FIGS. 6A to 6D are the same as the switching elements Tr1 to Tr6 of the inverter circuit 34, and the return diodes Di11 to Di12 are the same as the return diodes Di1 to Di6 of the inverter circuit 34. Use the same one.

Then, as shown in FIGS. 6A and 6B, the collectors of the switching elements Tr11 and Tr12 are turned on while one of the switching elements Tr11 and Tr12 is turned on and the current I flows in the direction indicated by the arrow. The emitter-to-emitter voltage is measured, and the average value is defined as a forward voltage drop Vce. As shown in FIGS. 6C and 6D, the switching elements Tr11, Tr
With both of them turned off and the current I flowing in the direction of the arrow shown in the figure, the voltage between both ends of the return diodes Di11 and Di12 is measured, and the average value is defined as the forward voltage drop Vd. In these cases, by performing the above-mentioned measurement while changing the value of the current I, the forward voltage drops Vce (I) and Vd (I) that change according to the magnitude of the current I are obtained (the graph of FIG. 7). reference). These forward voltage drops Vce (I), Vd
(I) is stored in the C-E forward voltage drop calculator 67 and the diode forward voltage drop calculator 68 in correspondence with the magnitude of the current I.

Returning to the description of FIG. 3 again, the combining operation unit 69 calculates the forward voltage drops Vce (Iu), Vce (Iv), and Vce (Iw) from the CE forward voltage drop calculating unit 67. , The forward voltage drop Vd (Iu) from the diode forward voltage drop calculator 68,
Vd (Iv) and Vd (Iw) are added and synthesized at an appropriate synthesis ratio determined according to the electrical angle θe and the angular velocity ω for each phase, so that the forward voltage compensation values ΔVonu, Δ
Vonv and ΔVonw are calculated, and the calculated forward voltage compensation value Δ
Vonu, ΔVonv, and ΔVonw are supplied to arithmetic units 51 to 53 via arithmetic units 54 to 59.

[0044]

ΔVonu = K (Vce (Iu), Vd (Iu), θe, ω)

[0045]

ΔVonv = K (Vce (Iv), Vd (Iv), θe, ω)

[0046]

ＶVonw = K (Vce (Iw), Vd (Iw), θe, ω)

The function K (Vce (Iu, Iv, I)
w), Vd (Iu, Iv, Iw), θe, ω) are determined by the respective forward voltage drops Vce (Iu), Vce (Iv), Vce in each phase.
(Iw) and each forward voltage drop Vd (Iu), Vd (Iv), Vd (Iw)
Changes depending on the on / off state of the switching elements Tr1 to Tr6 as shown in FIG. 8, and changes according to the electrical angle θe and the angular velocity ω. Therefore, each forward voltage drop Vce (Iu), Vce (Iv), Vce in each phase of the function K
(Iw) and each forward voltage drop Vd (Iu), Vd (Iv), Vd (Iw)
It is preferable that a combination ratio is determined in advance by simulation or the like, and then a parameter representing a ratio function K that changes according to the electrical angle θe and the angular velocity ω is stored in the combination calculation unit 69.

Then, these forward voltage compensation values ΔVonu,
ΔVonv and ΔVonw are supplied to arithmetic units 51 to 53 via arithmetic units 54 to 59, and the two-phase /
Command voltage values Vuo *, Vv of each phase from the three-phase coordinate conversion unit 31
o * and Vwo *. Therefore, the currents Iu,
Switching elements Tr1 to Tr6 that change according to Iv and Iw
And the respective forward voltage drops Vce of the freewheel diodes Di1 to Di6 are compensated.

Next, the pulse width error compensation value calculation unit 32D will be described. The compensation value calculation unit 32D includes a dead time Tdt for controlling switching of the switching elements Tr1 to Tr6 in the inverter circuit 34, and a switching time between the switching elements Tr1 to Tr6. On change delay time Ton when Tr6 changes from the off state to the on state, and switching elements Tr1 to T
Compensates for the voltage decrease Vdt due to the OFF change delay time Toff when r6 changes from the ON state to the OFF state.
It comprises a pulse width error calculator 71 and a voltage decrease calculator 72.

The pulse width error calculator 71 stores in advance a pulse width error ΔT (I) that is measured in advance and changes according to the current value (passing current value) I. This pulse width error refers to the switching elements Tr1 to T1 in the inverter circuit 34.
It represents the difference between the pulse width designated by the PWM voltage generator 33 for turning on r6 and the pulse width at which the switching elements Tr1 to Tr6 are actually turned on. The pulse width error calculation unit 71 stores the detected current values Iu, Iv, I of the three phases.
w are supplied, and the calculation unit 71 outputs pulse width errors Tn (Iu), ΔT (Iv), and ΔT (Iw) for three phases. Also in this case, the pulse width error calculator 71 stores a small number of pulse width errors ΔT (I) with respect to the representative current I, and outputs the detected current values Iu, Iv, and Iw. The pulse width errors ΔT (Iu), ΔT (Iv), and ΔT (Iw) are read out using the higher-order plural bit signals, respectively, and the higher-order bits of the plural-bit signals representing the same current values Iu, Iv, Iw are read out. And the read pulse width errors ΔT (Iu), ΔT (Iv), ΔT
T (Iw) may be interpolated and output.

Here, the pulse width error ΔT (I) will be described in some detail. First, the pulse width error ΔT
To measure (I), a circuit for one phase as shown in FIGS. 9A and 9B is configured. The switching elements Tr21 and Tr22 of FIGS. 9A and 9B are the same as the switching elements Tr1 to Tr6 of the inverter circuit 34, and the freewheeling diode Di2 is used.
1 and Di22 are also freewheeling diodes Di1 to Di1 of the inverter circuit 34.
The same as Di6 is used. Although omitted in FIGS. 9A and 9B, the switching elements Tr21 and Tr22 are omitted.
There is provided a dead time circuit for delaying the rise of the pulse signal by a predetermined dead time Td when the pulse signal for switching from the off state to the on state rises.

Then, as shown in FIG. 9A, a pulse signal having a duty ratio of 50% is applied to the switching element Tr21 and a current I flows in the direction indicated by the arrow in FIG. Is measured. FIG. 10A shows the delay state of the measured pulse signal. From this, the pulse rise delay time Tup1 obtained by adding the on-time delay time Ton of the switching element Tr21 to the dead time Tdt.
And the OFF change delay time To of the switching element Tr21.
A pulse falling delay time Tdn1 equal to ff is obtained. Based on these two times Tup1 and Tdn1, the current is changed as shown in FIG.
Pulse width error ΔT1 = Tu in the case of (I> 0)
Calculate p1-Tdn1.

As shown in FIG. 9B, a pulse signal having a duty ratio of 50% is applied to the switching element Tr22 so that a current I of the same magnitude as described above flows in the direction of the arrow in the drawing. Di22 (Inverter circuit output)
Is measured. FIG. 10B illustrates the delay state of the measured pulse signal, from which a pulse rise delay time Tup2 equal to the on-change delay time Ton of the switching element Tr22 is shown.
Then, a pulse falling delay time Tdn2 obtained by adding the dead time Tdt to the OFF change delay time Toff of the switching element Tr22 is obtained. Based on these two times Tup2 and Tdn2, a pulse width error ΔT2 = Tup2−Tdn2 when the current is in FIG. 9B (I <0) is calculated.

Next, an average value ΔTav = ΔT1 + ΔT2 of the pulse width errors ΔT1 and ΔT2 is calculated, and the average value ΔTav is set as a pulse width error ΔT (I) corresponding to the current I. Such measurement of the pulse width errors ΔT1, ΔT2, and ΔTav is performed while changing the current I variously, and various pulse width errors ΔT1 (I), ΔT2 (I), and ΔTav (I) corresponding to the current I are obtained.
The graph of FIG. 11 shows the pulse width errors ΔT1 (I), ΔT2 (I), and ΔTav (I) that change according to the current I. Then, as described above, this pulse width error ΔT (I) (= ΔTav (I)) is stored in the pulse width error calculator 71 in accordance with the magnitude of the current I.

Returning to the description of FIG. 3 again, the voltage decrease calculating section 72 calculates the pulse width errors ΔT (Iu), ΔT (Iv), and ΔT (Iw) of each phase from the pulse width error calculating section 71. Based on the pulse width error voltage compensation values ΔVdtu, ΔVdtv, ΔVdt
w is calculated by executing the following mathematical operations.

[0056]

ΔVdtu = Vdc · Fc · ΔT (Iu)

[0057]

[Expression 9] ΔVdtv = Vdc · Fc · ΔT (Iv)

[0058]

ΔVdtw = Vdc · Fc · ΔT (Iw)

In the equations (8) to (10), Vdc is a predetermined constant indicating a DC input voltage value to the inverter circuit 34, and Fc is a predetermined constant indicating a carrier frequency used in the PWM voltage generator 33. It is a given constant.

These pulse width error voltage compensation values .DELTA.Vdtu, .DELTA.Vdtv, and .DELTA.Vdtw are supplied to arithmetic units 51 to 53 via arithmetic units 54 to 62. Command voltage value V of each phase from converter 31
It is added to uo *, Vvo *, and Vwo *. Therefore, it changes according to the currents Iu, Iv, Iw of each phase, and the inverter circuit 3
4 dead time Td of the switching elements Tr1 to Tr6
The voltage decrease Vdt due to t, the on-change delay time Ton, and the off-change delay time Toff is compensated (FIG. 12).
reference).

Next, the operation of the embodiment configured as described above will be described. When the driver turns the steering wheel 12, the turning operation is transmitted to the tie rod 13, and the front wheel is steered by the axial movement of the rod 13. At the same time, the steering torque sensor 15
The brushless motor 1 detects the steering torque applied to the
1 is servo-controlled by the electric control device and drives the tie rod 13 with an assist torque corresponding to the steering torque, so that the front wheels are steered while being assisted by the driving force of the brushless motor 11.

In the servo control by this electric control device, the basic assist force calculation unit 21, the return force calculation unit 22, and the calculation unit 23 are based on the detected steering torque, vehicle speed, rotor electrical angle θe and angular speed ω. In addition to calculating the command torque T *, the command current value determination unit 24 determines the command field current value Id * and the command torque current value Iq * based on the command torque T * and other various sensor values. And
The operation units 25 and 26, the proportional-integral control units 27 and 28, and the two-phase / three-phase coordinate conversion unit 31 provide a command voltage value Vu0 for each phase based on the command field current value Id * and the command torque current value Iq *.
*, Vv0 *, Vw0 *, and the feedforward compensator 32 corrects the command voltage values Vu0 *, Vv0 *, Vw0 * of each phase to the command voltage values Vu1 *, Vv1 *, Vw1 *, and then the PWM voltage The generating unit 33 and the inverter circuit 34 output the respective voltages Vu, V according to the corrected command voltage values Vu1 *, Vv1 *, Vw1 *.
v and Vw are given to each phase of the brushless motor 11 to control the rotation of the motor 11. In this case, the brushless motor 11 changes the detected field current value Id and the detected torque current value Iq to the command field current value Id * and the command torque current value Iq *, respectively.
The operation is controlled so as to coincide with
Generates a rotation torque corresponding to the command torque T *, and assists the turning operation of the steering wheel 12.

As described above, in the servo control of the brushless motor 11, the feedforward compensator 32 compensates for various voltage errors by feedforward control of the command voltage values Vu0 *, Vv0 *, Vw0 * of each phase. Specifically, the unbalanced voltage compensation value calculation unit 32A and the calculation units 51 to 51
53 adds the compensation voltage values ΔVunu, ΔVunv, ΔVunw of each phase for compensating for the imbalance of the brushless motor 11 to the command voltage values Vu0 *, Vv0 *, Vw0 * of each phase, and thereby Compensate for unbalanced components in all operating conditions. Therefore, even if the brushless motor 11 is configured to be unbalanced, desired control characteristics according to operation control parameters such as the command torque T *, the command field current value Id *, and the command torque current value Iq * can be obtained. This means that unbalanced polyphase AC motors can be used in various controls, so in the production of polyphase AC motors, simplification of the motor structure, freedom of component shapes, material This increases the degree of freedom in design, such as the room for selection, and in turn leads to lower manufacturing costs for the polyphase AC motor.

The quantization error due to the operation, the error due to the forward voltage drop of the inverter circuit 34, and the pulse width error of the inverter circuit 34 are also calculated by the quantization error compensation value calculator 32.
B, since the compensation is performed by the voltage drop compensation value calculation unit 32C, the pulse width error compensation value calculation unit 32D, and the calculation units 51 to 53, the control accuracy of the brushless motor 11 is improved, and more favorable steering assist control for the steering wheel 12 is performed. Is performed.

The unbalanced voltage compensation value calculator 32A in the above embodiment may be modified as shown in FIG. The unbalanced voltage compensation value calculation unit 32A according to this modification stores the voltage waveform ν = fm (θe, ω, Id, Iq) including the unbalanced component waveform obtained by the measurement or the simulation as described above. Then, the difference between the voltage waveform ν and the fundamental wave component ν1 = fm1 (θe, Δθe, | ν1 |) obtained by real-time calculation is obtained, and the three-phase unbalanced voltage compensation values ΔVunu, ΔVunv, ΔVunw
Is calculated.

Therefore, the unbalanced voltage compensation value calculating section 32A according to this modification converts the voltage waveform ν = fm (θe, ω, Id, Iq) including the unbalanced component waveform into the angular velocity ω and the field current value. An unbalanced voltage waveform memory 81 stored in correspondence with various values of Id and the torque current value Iq is provided. In the memory 81, in addition to the angular velocity ω, the detected field current value Id and the detected torque current value Iq, The angle θe is input as a read address signal. The voltage waveform ν read from the unbalanced voltage waveform memory 81 is supplied to the interpolation unit 82, and the interpolation unit 82 converts the voltage waveform ν
Angular velocity ω, detected field current value Id and detected torque current value Iq
And performs the same interpolation calculation as in the above-described embodiment, and supplies the result to the negative input of the calculation unit 83.

The positive input of the arithmetic unit 83 has a fundamental wave component ν
A sine wave memory 84 for calculating 1 = fm1 (θe, Δθe, | ν1 |), an amplitude calculation unit 85, and a calculation unit 86 are connected. The sine wave memory 84 stores a waveform sampling value representing a sine wave, calculates a sine wave signal sin (θe + Δθe) according to the input electrical angle θe and phase difference Δθe, and supplies the calculated signal to the calculation unit 86. The amplitude calculator 85 receives the command field voltage value Vd * and the command torque voltage value Vq *, and calculates the amplitude value of the fundamental wave component ν1 by taking the square root of the sum of squares of the two voltage values Vd * and Vq *. | Ν1 | = (Vd * ² + Vq * ² ) ^1/2 is calculated. The calculation unit 86 calculates the fundamental wave component ν1 by executing the calculation of the following Expression 11 by multiplying the calculation results of the sine wave memory 84 and the amplitude calculation unit 85.
To the positive input of

[0068]

Ν1 = (Vd * ² + Vq * ² ) ^1/2 · sin (θe + Δθe) = | ν1 | · sin (θe + Δθe) = fm1 (θe, Δθe, | ν1 |)

The calculation unit 83 calculates the compensation voltage waveform −ν = −fun (θe, ω, Id, Iq) by executing the calculation of the following equation 12 for subtracting the voltage waveform ν from the fundamental wave component ν1.

[0070]

-Δνun = ν1−ν = fm1 (θe, Δθe, | ν1 |) −fm (θe, ω, Id, Iq) = − fun (θe, ω, Id, Iq)

The calculation result by the calculation unit 83 is output to the three-phase output unit 6 configured in the same manner as in the above embodiment.
5 is supplied. Therefore, also in the unbalanced voltage compensation value calculation unit 32A according to this modified example, the three-phase unbalanced voltage compensation values ΔVunu, ΔVunv, ΔVun similar to the above-described embodiment.
Since w is calculated, the same effect as in the above embodiment can be expected in this case as well.

Further, in the compensation voltage waveform memory 63 in the above embodiment, the unbalanced voltage waveform memory 81 and the sine wave memory 84 in the above modification, the waveform sampling data is stored as it is. May be converted into a function, and a parameter representing the function may be stored, and each waveform value may be calculated and output by a function calculation with respect to the input signal.

Further, a forward voltage drop calculating section 67 between CE and
Also in the diode forward voltage drop calculator 68 and the pulse width error calculator 71, not only the measured data itself but also parameters representing the same function are stored by measuring the input / output function, and the function for the input signal is stored. Each data value may be calculated and output by calculation. In this case, FIG.
And the forward voltage drops Vce and Vd based on the graph of FIG.
And the average value ΔTav of the pulse width errors ΔT1 and ΔT2 may be converted into a function.

As for the forward voltage drops Vce and Vd, the two forward voltage drops Vce and Vd may be combined into one combined function as shown by the approximate straight line of the average value in FIG. it can. In this case, instead of the C-E forward voltage drop calculation unit 67, the diode forward voltage drop calculation unit 68, and the combining operation unit 69 of FIG. 3, the forward voltage compensation value ΔVonu, What is necessary is just to provide the synthesis function generation part which each produces (DELTA) Vonv and (DELTA) Vonw. Even in this simplification, the duty ratio of the PWM voltage pattern becomes approximately 50% at the time of low-speed driving in which the voltage of each phase significantly decreases, and the current flows between the collector and the emitter of the switching elements Tr1 to Tr6 and the freewheel diode Di1. .About.Di6 for substantially the same time, so that there is substantially no problem.

[Brief description of the drawings]

FIG. 1 is an overall schematic diagram of an electric power steering device for a vehicle to which an electric control device for a polyphase AC motor according to an embodiment of the present invention is applied.

FIG. 2 is a circuit diagram showing a part of the inside of the inverter circuit of FIG. 1;

FIG. 3 is a detailed block diagram of a feedforward compensator of FIG. 1;

FIGS. 4A to 4D are explanatory diagrams for explaining an unbalanced component generated in a three-phase AC motor and an influence of the unbalanced component.

FIG. 5 is an explanatory diagram for explaining how to extract the unbalanced component.

FIGS. 6A to 6D are circuit diagrams for measuring a forward voltage drop occurring in the inverter circuit of FIG. 1;

FIG. 7 is a graph showing a change in the forward voltage drop.

FIG. 8 is an explanatory diagram for explaining a state of occurrence of the same forward voltage drop.

FIGS. 9A and 9B are circuit diagrams for measuring a pulse width error generated in the inverter circuit of FIG. 1;

FIGS. 10A and 10B are pulse waveform diagrams showing the state of occurrence of the pulse width error.

FIG. 11 is a graph showing a change in the pulse width error.

FIG. 12 is an explanatory diagram for explaining a voltage decrease caused by the pulse width error.

FIG. 13 is a block diagram showing a modification of the unbalanced voltage compensation value calculator of FIG.

[Explanation of symbols]

11 ... brushless motor, 12 ... steering wheel, 15 ...
Steering torque sensor, 16: rotation angle sensor, 21: basic assist force calculation unit, 24: command current value determination unit, 27, 28
... Proportional integration control unit (PI control unit), 31 ... 2-phase / 3-phase coordinate conversion unit, 32 ... Feed forward compensation unit, 32A ...
Unbalanced voltage compensation value calculator, 32B: quantization error compensation value calculator, 32C: voltage drop compensation value calculator, 32D: pulse width error compensation value calculator, 33: PWM voltage generator, 34 ...
Inverter circuit, 35, 36 current sensor, 37 three-phase / two-phase coordinate conversion part, 41 electric angle conversion part, 42 angular velocity conversion part, 63 compensation voltage waveform memory, 64, 82 interpolation part, 65 ... 3-phase output unit, 81 ... unbalanced voltage waveform memory,
84: sine wave memory, 85: amplitude calculator.

 ──────────────────────────────────────────────────続 き Continued on the front page F term (reference) 3D033 CA03 CA11 CA13 CA16 CA20 CA21 5H560 AA08 BB04 BB07 BB12 DC12 EB01 EC04 GG04 JJ04 UA06 XA02 XA12

Claims (3)

[Claims] An operation command of each phase is calculated in accordance with an operation control parameter for controlling an operation state of a polyphase AC motor, and the operation of the polyphase AC motor is performed according to the calculated command voltage value of each phase. An electric control device for a polyphase AC motor for controlling a state, comprising: an operation state detecting means for detecting an operation state of the polyphase AC motor; and a compensation voltage value for each phase for compensating for imbalance of the polyphase AC motor. Compensation voltage value calculating means for calculating based on the detected operation state; and correcting the command voltage value of each phase by adding the calculated compensation voltage value of each phase to the command voltage value of each phase. An electric control device for a polyphase AC motor, comprising a correction unit. 2. The electric control device for a polyphase AC motor according to claim 1, wherein said compensation voltage value calculating means generates a compensation voltage waveform corresponding to a voltage unbalance component for one phase of the polyphase AC motor. A storage means for storing data representing the operation state of the polyphase AC motor in advance, and a compensation voltage value for each phase based on the data read from the storage means in accordance with the detected operation state. The electric control device of the polyphase AC motor to be calculated. 3. The electric control apparatus for a polyphase AC motor according to claim 1, wherein said compensation voltage value calculating means calculates an unbalanced voltage waveform for one phase including a voltage unbalance component of the polyphase AC motor. Storage means for storing data representing the operation state of the polyphase AC motor in advance, and an unbalanced voltage waveform represented by data read from the storage means in accordance with the detected operation state. Compensation voltage waveform calculation means for calculating a compensation voltage waveform corresponding to the voltage unbalance component for one phase by calculating a difference between the fundamental voltage component of the balanced voltage waveform and the compensation voltage waveform based on the calculated compensation voltage waveform. An electric control device for a multi-phase AC motor for calculating a compensation voltage value for each phase.