JP2000324827A - Switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To prevent abnormal operation of a switching element which corresponds to the reverse recovery time of a clamp diode. SOLUTION: A clamp circuit 2 that a self-excitating resonance-type converter is provided with is formed by a Schottky barrier diode DSBD and a resistor RD, thus eliminating a period as reverse recovery time from forming, when a damper period ends at the turn-off time of a switching element. Then, the switching element is controlled so as to be turned on when the damper period ends, and the occurrence of an abnormal switching operation is prevented, even in the case of a stable operation and under the conditions of low AC input voltage by a heavy load.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

[0002]

2. Description of the Related Art As a switching power supply circuit, a circuit employing a switching converter of a type such as a flyback converter or a forward converter is widely known. Since these switching converters have a rectangular switching operation waveform, there is a limit in suppressing switching noise. Also, due to its operating characteristics,
It has been found that there is a limit in improving the power conversion efficiency. Therefore, the present applicant has previously proposed various switching power supply circuits using various resonance type converters.
The resonance type converter can easily obtain high power conversion efficiency and realize low noise because the switching operation waveform is sinusoidal. It also has the advantage that it can be configured with a relatively small number of parts.

FIG. 8 is a circuit diagram showing an example of a switching power supply circuit as a prior art, which can be configured based on the invention proposed by the present applicant. This power supply circuit is provided with a voltage resonance type converter having a single switching element Q1 and performing a switching operation by a self-excited method in a so-called single-ended system.

In the power supply circuit shown in FIG. 1, a bridge rectification circuit Di is used as a rectification and smoothing circuit for receiving a commercial AC power supply (AC input voltage VAC) and obtaining a DC input voltage.
And a full-wave rectifier circuit composed of a smoothing capacitor Ci, and generates a rectified smoothed voltage Ei corresponding to a level that is one time the AC input voltage VAC. Further, in the rectifying / smoothing circuit, an inrush current limiting resistor Ri is inserted into the rectifying current path so as to suppress, for example, an inrush current flowing into the smoothing capacitor when the power is turned on.

A voltage resonance type switching converter provided in this power supply circuit has a single switching element Q1.
It has a self-excited configuration with In this case, a high withstand voltage bipolar transistor (BJT; junction transistor) is employed as the switching element Q1.

The base of the switching element Q1 is connected to a smoothing capacitor Ci1 (rectified smoothing voltage 2) via a starting resistor RS.
It is connected to the positive electrode side of Ei) so that the base current at the time of starting can be obtained from the rectification smoothing line. A resonance circuit for self-excited oscillation driving is connected between the base of the switching element Q1 and the ground on the temporary side, and includes a series connection circuit of an inductor LB, a detection drive winding NB, a resonance capacitor CB, and a base current limiting resistor RB. . A clamp diode DD inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a clamp current flowing when the switching element Q1 is turned off. , The collector of switching element Q1 is connected to one end of primary winding N1 of insulating converter transformer PIT, and the emitter is grounded.

A parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. This parallel resonance capacitor Cr
Forms a primary parallel resonance circuit of the voltage resonance type converter by its own capacitance and a leakage inductance L1 on the primary winding N1 side of the insulating converter transformer PIT described later. Although the detailed description is omitted here, when the switching element Q1 is turned off, the voltage Vcr across the resonance capacitor Cr actually becomes a sinusoidal pulse waveform due to the action of the parallel resonance circuit, and the voltage resonance type operation is performed. Is obtained.

The orthogonal control transformer PRT shown in FIG.
Is a saturable reactor on which a detection winding ND, a drive winding NB, and a control winding NC are wound. The orthogonal transformer PRT drives the switching element Q1 and is provided for constant voltage control. As the structure of the orthogonal control transformer PRT, although not shown, a three-dimensional core is formed by joining the ends of the two magnetic legs of two double U-shaped cores having four magnetic legs. I do. And
The detection winding ND and the drive winding NB are wound around the predetermined two magnetic legs of the three-dimensional core in the same winding direction, and the control winding NC is further connected to the detection winding ND and the drive winding ND. It is configured by being wound in a direction orthogonal to the line NB.

In this case, the detection winding ND of the orthogonal control transformer PRT is inserted in series between the positive electrode of the smoothing capacitor Ci and the primary winding N1 of the insulating converter transformer PIT, so that the switching of the switching element Q1 is performed. The output is transmitted to the detection winding ND via the primary winding N1.
In the orthogonal control transformer PRT, the switching output obtained on the detection winding ND is excited by the drive winding NB via the transformer coupling, so that an alternating voltage as a drive voltage is generated on the drive winding NB. . This drive voltage is applied to a series resonance circuit (NB,
CB) is output to the base of the switching element Q1 as a drive current via the base current limiting resistor RB.
As a result, the switching element Q1 performs a switching operation at a switching frequency determined by the resonance frequency of the series resonance circuit (NB, CB).

The insulated converter transformer PIT for transmitting the switching output of the switching element Q1 to the secondary side, as shown in FIG.
For example, an EE-type core in which E-type cores CR1 and CR2 made of a ferrite material are combined so that their magnetic legs are opposed to each other is provided. The wire N1 and the secondary winding N2 are wound separately. A gap G is formed for the center magnetic leg as shown in the figure. As a result, loose coupling with a required coupling coefficient can be obtained. The gap G is the E type core CR1,
The CR2 can be formed by forming the central magnetic leg shorter than the two outer magnetic legs. The coupling coefficient k is set to a loosely coupled state of, for example, k ≒ 0.85, so that a saturated state is hardly obtained.

One end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the switching element Q1, and the other end is connected to the positive terminal of a smoothing capacitor Ci through a series connection of a detection winding ND as shown in the figure. (Rectified smoothing voltage E
i).

On the secondary side of the insulated converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, the secondary parallel resonance capacitor C2 is connected in parallel to the secondary winding N2, so that the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary parallel resonance capacitor C2 are determined. A parallel resonance circuit is formed. With this parallel resonance circuit,
The alternating voltage excited by the secondary winding N2 becomes a resonance voltage.
That is, a voltage resonance operation is obtained on the secondary side.

That is, in this power supply circuit, the primary side is provided with a parallel resonance circuit for making the switching operation a voltage resonance type, and the secondary side is provided with a full-wave rectification operation (voltage resonance operation).
And a parallel resonance circuit for obtaining In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

In the secondary parallel resonance circuit formed as described above, a tap is provided for the secondary winding N2, and the rectifier diodes DO1, DO2 and the smoothing capacitors CO1, CO2 are connected. By connecting as shown in the figure, two sets of half-wave rectifier circuits each including a set of [rectifier diode DO1, smoothing capacitor CO1] and a set of [rectifier diode DO2, smoothing capacitor CO2] are provided. [Rectifier diode DO1,
The half-wave rectifier circuit composed of the set of the smoothing capacitor CO1] receives the resonance voltage supplied from the secondary parallel resonance circuit to generate a DC output voltage EO1, and generates a full-wave composed of the [rectifier diode DO2, the smoothing capacitor CO2]. Similarly, the rectifier circuit receives the resonance voltage supplied from the secondary-side parallel resonance circuit and generates a DC output voltage EO2. In this case, the DC output voltage EO1 and the DC output voltage EO2 are also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operation power supply of the control circuit 1.

Incidentally, the insulation converter transformer PIT
, The polarity (winding direction) of the primary winding N1 and the secondary winding N2 and the rectifier diode DO (DO1, DO2 / DO3, DO4)
, The mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2 may be + M or -M. For example, when the connection form shown in FIG. 10A is adopted, the mutual inductance is + M,
When the connection form shown in (b) is adopted, the mutual inductance is -M. When this is made to correspond to the operation on the secondary side shown in FIG. 8, for a half-wave rectifier circuit using a set of [rectifier diode DO1, smoothing capacitor CO1], for example, the alternating voltage obtained in the secondary winding N2 is When the polarity is positive, a rectification current flows through the rectifier diode DO1 to perform the + M operation mode (forward mode). Conversely, when the alternating voltage obtained in the secondary winding N2 has a negative polarity, the rectifier diode DO1 is turned off. As a result, no rectified current flows. That is, in this power supply circuit, the rectifying operation is performed in a mode where the mutual inductance is + M.

In such a configuration, the electric power is supplied to the load side increased by the action of the primary side parallel resonance circuit and the secondary side parallel resonance circuit, and the electric power supplied to the load side is also increased. The rate of increase in load power is also improved. This is because, as described above with reference to FIG. 10, the gap G is formed in the insulating converter transformer PIT and loose coupling is performed with a required coupling coefficient, thereby obtaining a state that is more difficult to be saturated. It is realized. For example, when the gap G is not provided for the insulating converter transformer PIT, there is a high possibility that the insulating converter transformer PIT becomes saturated during flyback operation and the operation becomes abnormal. It is difficult to hope that this is done properly.

In the control circuit 1, the level of the control current (DC current) flowing through the control winding NC is varied according to the change in the secondary side DC output voltage level (EO1), so that the winding is wound around the orthogonal control transformer PRT. The inductance LB of the mounted drive winding NB is variably controlled. Thereby, the switching element Q1 formed including the inductance LB of the drive winding NB is formed.
The resonance condition of the series resonance circuit in the self-excited oscillation driving circuit for the above changes. This is an operation for varying the switching frequency of the switching element Q1, as described below with reference to FIG. 8, and this operation stabilizes the secondary-side DC output voltage.

Here, in the case where an orthogonal control transformer PRT having a variable control structure is provided with the inductance LB of the drive winding NB as shown in FIG. 8, the switching element Q1 is turned off to change the switching frequency. Period TOFF is fixed, and the period T
N is variably controlled. In other words, in this power supply circuit, as a constant voltage control operation, the switching frequency is variably controlled to perform resonance impedance control for the switching output, and at the same time,
It can be seen that the conduction angle control (PWM control) of the switching element in the switching cycle is also performed. This complex control operation is realized by a set of control circuit systems.

FIG. 12 is a waveform diagram showing the operation of the main part in the switching cycle of the circuit shown in FIG. 8 and is a waveform at the time of stable operation. In the series resonance circuit (NB, CB) as a self-excited oscillation drive circuit, the resonance operation is performed by the alternating voltage obtained in the drive winding NB, so that the series resonance current IO via the base current limiting resistor RB is reduced. , FIG. 12 (e)
As shown in the figure, a sinusoidal waveform is obtained. Then, the series resonance current IO is combined with the current ID of FIG.
As shown in FIG. 12D, a base current (drive current) IB flows through the base of the switching element Q1. With the driving current IB, the switching element Q1 performs a switching operation. The collector current Icp of the switching element Q1 controlled by the drive current IB is:
The waveform shown in FIG. 12B is obtained. Further, at both ends of the parallel connection circuit of the switching element Q1 // parallel resonance capacitor Cr, a parallel resonance voltage Vcp is generated by the operation of the parallel resonance circuit as shown in FIG. As shown in the figure, the parallel resonance voltage Vcp has a waveform of a sine wave pulse during the period TOFF when the switching element Q1 is on and the period TOFF when the switching device Q1 is off as shown in FIG. Yes, it is. As shown in FIG. 12C, a sine-wave parallel resonance current Icr flows through the parallel resonance capacitor Cr during the period TOFF.

Here, during the period TON when the switching element Q1 is turned on, the series resonance current IO shown in FIG.
The positive current region (period T1) corresponds to the forward bias current region of the drive current IB in FIG. Also,
In the same period TON, a region (T2) where the series resonance current IO is a negative current is a reverse bias current of the drive current IB. The area of the reverse bias current of the drive current IB during this period TON is the accumulation time tstg of the switching element Q1.

A low-speed clamp diode DD whose reverse recovery time trr is increased to a predetermined value is connected between the base and the emitter of the switching element Q1. In the period TOFF during which the switching element Q1 is turned off, as shown in FIG. 12 (f), the series resonance current IO changes from the clamp diode DD to the base current limiting resistor RB to the resonance capacitor CB.
It flows through the drive winding NB. Then, when the period TON starts, the damper period immediately starts, and the charging / discharging energy of the parallel resonance capacitor Cr becomes equal to the damper current (I
As D), the current flows through the clamp diode DD → the base → the collector of the switching element Q1. In this case, as shown in FIG. 3F, the damper current (ID) has a level of 2.3 Ap. At the end of the damper period, the clamp diode DD has a reverse recovery time t.
rr region. In this reverse recovery time trr,
The clamp diode DD is conducting. And
When the driving current IB increases during the reverse recovery time trr and a current amount enough to turn on the switching element Q1 is obtained, the switching element Q1 is turned on. However,
At the time when the reverse recovery time trr is started, no current is actually supplied to the base of the switching element Q1 even if the drive current IB (base current) in FIG. Therefore, the switching element Q1 does not turn on immediately, and thereafter, the switching element Q1
When the base current level of 1 reaches a level sufficient to make the switching element Q1 conductive, the switching element Q1 transitions to the ON state at a certain timing.

[0022]

Here, FIG. 11 shows:
The operation waveforms of the primary side parallel resonance voltage Vcp, the collector current Icp, and the drive current IB are shown. FIG. 11 (a)
(B) and (c) show the operation under the condition of the maximum load power with the AC input voltage Vac = 80 V, and FIGS.
(F) shows the operation under the condition of the AC input voltage Vac = 60 V and the maximum load power.

First, as an operation under the condition of the AC input voltage Vac = 80 V and the maximum load power, FIG.
The period T3 in (b) and (c) corresponds to the damper period, and the period T4 corresponds to the reverse recovery time trr. Under such heavy load and low AC input voltage conditions, FIG. ), The drive current IB does not flow in the forward direction even during the period T4 (reverse recovery time trr), so that the switching element Q1 is not turned on. Therefore, the switching operation becomes an abnormal operation.
In such an abnormal operation, for example, as shown in FIG. 11B, high-frequency noise as parasitic oscillation is superimposed on the collector current Icp immediately after the period T4, and accordingly, FIG. Appear as a phenomenon that pulse noise occurs as shown by the parallel resonance voltage Vcp.

Under the condition of the AC input voltage Vac = 60 V and the maximum load power, this abnormal operation is further enlarged. That is, as shown in FIG.
(Reverse recovery time trr) varies. In this case, the period T4 becomes longer and the reverse current level of the drive current IB increases, so that the collector current I shown in FIG.
As in cp, the amplitude of noise superimposed on the same period T4 increases. Then, with this, FIG.
The level of the pulse noise of the parallel resonance voltage Vcp shown in FIG.

In order to cope with the condition of heavy load by reducing the phenomenon as the abnormal operation described above, in practice of the circuit shown in FIG. 8, the capacitance of the secondary side parallel resonance capacitor C2 is the required maximum load power. In order to obtain a value, a value larger than the originally optimum value is selected. However, in this case, the primary-side resonance current I1 flowing through the primary winding N1 and the secondary-side resonance current I1 in the circuit shown in FIG. The power loss increases as the secondary side resonance current I2 flowing in the line N2 increases.

In addition, in the case of the circuit configuration shown in FIG. 8 described above, the primary-side parallel resonance current Icr falls within the period TON as shown in FIG. In the start period, noise as parasitic oscillation occurs. Accordingly, although slightly, noise also occurs in the parallel resonance voltage Vcp which is a switching pulse. Also, the driving current IB shown in FIG. 11D also causes parasitic oscillation in the start period within the period TOFF.

[0027]

In view of the above-mentioned problems, the present invention is configured as a switching power supply circuit as follows. That is, a switching device including a switching device, for intermittently outputting the input DC input voltage, an insulating converter transformer for transmitting the output of the switching device to the secondary side, and the switching device are formed. A self-excited oscillation drive circuit that performs switching driving by a self-excited system, a primary-side resonance circuit inserted so that the operation of the switching means is of a resonance type, and a secondary-side resonance capacitor for a secondary winding of an insulating converter transformer Is connected, the secondary resonance circuit that forms a resonance circuit by the leakage inductance component of the secondary winding of the insulating converter transformer and the capacitance of the secondary resonance capacitor, and the alternating voltage obtained in the secondary winding It is configured to input and perform half-wave rectification operation or full-wave rectification operation to obtain a secondary-side DC output voltage. Current output voltage generation means, constant voltage control means for performing constant voltage control by varying the switching frequency of the switching element according to the voltage level obtained on the secondary side, at least no reverse recovery time And a clamp circuit formed by connecting a diode element and a resistor element in series, and provided to flow a current in a negative direction to the self-excited oscillation drive circuit during a period when the switching element is turned off. Constitute.

According to the above configuration, the clamp circuit is formed by a diode element having a characteristic that has no reverse recovery time and a resistor connected in series. With such a clamp circuit, there is no reverse recovery time as a diode element for the clamp function.

[0029]

FIG. 1 is a circuit diagram showing a configuration of a power supply circuit according to an embodiment of the present invention. In this figure, the same parts as those in FIG.

The power supply circuit shown in FIG. 1 has a self-excited oscillation driving circuit (CB-
A Schottky barrier diode DSBD and a resistor RD are provided as a clamp circuit 2 for supplying a clamp current which is a negative level current to NB). Here, the Schottky barrier diode DSBD is the reverse recovery time tr
It is assumed that it has the characteristic of r = 0. Then, the anode side of the Schottky barrier diode DSBD is connected to the primary side ground via a series connection of a resistor RD, and the cathode is inserted into the circuit so as to be connected to the base of the switching element Q1. That is, the present embodiment employs a configuration in which a series connection circuit of a Schottky barrier diode DSBD and a resistor RD is provided instead of the low-speed clamp diode DD used in the circuit of FIG.

In this circuit, the base current limiting resistor RB shown in FIG. 8 is omitted. Therefore,
The output of the self-excited oscillation circuit composed of the drive winding NB and the resonance capacitor CB connected in series is directly connected to the base of the switching element Q1 without passing through the base current limiting resistor RB. Is done.

FIG. 3 shows operation waveforms of the power supply circuit having the above-described configuration according to the switching cycle. The operation waveform shown in this figure is obtained during a stable operation under the condition of the maximum input power with the AC input voltage Vac = 100V.

Here, the parallel resonance voltage Vcp on the primary side of FIG. 3A and the collector current Icp of FIG. 3B obtained with the switching operation of the switching element Q1.
Is similar to the waveform shown in FIG.

FIG. 3F shows a current ID flowing through the Schottky barrier diode DSBD. In the Schottky barrier diode DSBD, a clamp current flowing in the negative direction to the self-excited oscillation drive circuit (CB-NB) is conducted during a period TOFF when the switching element Q1 is turned off, and the switching element Q1 is turned on. When a certain period TON starts, a damper current flows through the base-collector of the switching element Q1 in a short damper period T3. However, since the Schottky barrier diode DSBD has the reverse recovery time trr = 0, after the damper period T3 ends, the phenomenon that the reverse recovery current flows due to the reverse recovery time trr disappears, and the Schottky barrier diode DSBD immediately disappears. DSBD is turned off.

By eliminating the period as the reverse recovery time trr in this manner, the operation of drawing the resonance current IO of the self-excited oscillation driving circuit shown in FIG. 3E to the primary side ground via the clamp diode DD, for example. Disappears. For this reason, the drive current IB shown in FIG. 2D is supplied to the base of the switching element Q1 as a forward current from the time when the damper period T3 ends and the damper current becomes 0 level, The switching element Q1 is turned on. That is, the switching element Q1 is turned on from the start of the period TON shown in FIG.

FIG. 3 shows a circuit according to this embodiment.
As can be seen by comparing the waveforms shown in FIGS. 12C and 12D with the waveforms shown in FIGS.
No noise due to parasitic oscillation is generated in cr and the drive current IB.

Further, as the drive current IB shown in FIG. 3D, the waveform during the period corresponding to the accumulation time tstg is shown in FIG.
The absolute value as the reverse current is larger than in the case of the drive current IB shown in (d). This is due to the fact that the forward voltage drop of the Schottky barrier diode DSTB is smaller than that of the low-speed clamp diode DD, so that the accumulated carriers when the switching element Q1 is turned off disappear more rapidly. Therefore, it also acts to shorten the accumulation time tstg and the fall time tf of the switching element Q1.

FIG. 2A, FIG. 2B and FIG.
(D) In (e) and (f), the primary side parallel resonance voltage Vcp,
7 shows operation waveforms of the collector current Icp and the drive current IB.
Here, FIGS. 2A, 2B, and 2C show the AC input voltage V
The operation at the time of maximum load power at ac = 60V is shown,
FIGS. 2D, 2E, and 2F show the AC input voltage Vac = 80.
The operation at the maximum load power is shown at V.

Even under such heavy load and low input voltage conditions, as shown in FIG. 2C or 2F, for example, the drive current IB is already positive after the damper period T3. The level is obtained, and for example, the switching element Q1 is controlled to be turned on immediately after the period TON. That is, a stable switching operation can be obtained. Further, since the reverse recovery time after the damper period is not formed, for example, the collector current Icp shown in FIGS. 2B and 2E has noise like the collector current Icp shown in FIGS. 11B and 11E. Accordingly, the noise pulse of the parallel resonance voltage Vcp appearing during the period TON does not occur as shown in FIGS. 2A and 2D.

For example, in the power supply circuit shown in FIG.
In order to avoid the abnormal operation described in 1 and the occurrence of noise shown in FIG. 12, the reverse recovery time trr is selected for the clamp diode DD, and the current amplification factor hFE of the switching element Q1 is classified. .
Further, the value of the base current limiting resistor RB is selected so as to obtain the optimum driving condition of the switching element.

On the other hand, in the present embodiment, the reverse recovery time trr = 0, and the cause of the variation in the reverse recovery time trr is eliminated. As described above, in the present embodiment, since the reverse base current (drive current IB) in the period corresponding to the accumulation time tstg increases, the accumulation time tstg and the current amplification factor h for the switching element Q1 are increased.
The range of FE ranking is expanded. Thereby, for example, improvement in manufacturing efficiency such as parts management is achieved.

As a comparison between the circuit shown in FIG. 1 as the present embodiment and the circuit shown in FIG. 8, under the circuit configuration shown in FIG. 8, the base current limiting resistor RB = 1Ω, the secondary side When the parallel resonance capacitor C2 = 0.19 μF is selected, the primary side resonance current I1 = 5.8 Ap and the collector current I
cp = 3.3 Ap, the secondary side resonance current I2 = 8 Ap-p, and the result that the input power Pin = 166.5 W was obtained. On the other hand, under the configuration shown in FIG. 1, the resistance R D = 8.2Ω and the secondary side parallel resonance capacitor C 2 = 0.0
12 μF can be selected. Here, C2 = 0.
The value of 012 μF is the value of the secondary side parallel resonance capacitor C.
2 is a value originally intended to be suitable. When the components are selected in this manner, under the operation described with reference to FIGS. 2 and 3, the primary-side resonance current I1 = 4.4 Ap, the collector current Icp = 2.8 ap, and the secondary-side resonance current I2 = 6. .8A
pp, and a result of input power Pin = 164.0 W was obtained. That is, in the present embodiment, the current levels of the primary side resonance current I1 and the secondary side resonance current I2 are suppressed as compared with the case of the circuit shown in FIG.
The power loss is reduced by about 5 W.
In addition, the maximum load power that can be handled at the same time is about 150 W in the circuit shown in FIG. 8, while the circuit of the present embodiment is expanded by about 50 W and raised to about 200 W. .

Next, a description will be given of a modified example of the power supply circuit of the present embodiment. First, FIG. 4 shows a configuration of the power supply circuit of the present embodiment as a first modified example. In this figure, the same parts as those in FIG.

In the power supply circuit shown in this figure, a ferrite bead inductor L is provided in the clamp circuit 2A.
The point where D is added is that the clamp circuit 2 shown in FIG.
And different. In this case, the ferrite bead inductor LD
Is inserted in series between the Schottky barrier diode DSBD and the resistor RD. For example, the insertion of the ferrite bead inductor LD absorbs noise when the Schottky barrier diode DSBD is turned off and further turned on. As such a clamp circuit 2A, a resistor RD = 4.7Ω and a ferrite bead inductor LD = 0.5 μH can be selected.

Also, as shown as a clamp circuit 2B in FIG. 4, a connection form in which a ferrite bead inductor LD is inserted in parallel with the resistor RD has a function of absorbing noise. The operation of the power supply circuit including the clamp circuit 2A or the clamp circuit 2B is such that the same operation as that shown in FIGS. 2 and 3 can be obtained.

The secondary side of the power supply circuit of the present embodiment is not limited to the configuration of the secondary side parallel resonance circuit and the half-wave rectifier circuit as shown in FIG. Therefore, the configuration of the secondary side as the second to fourth modified examples is shown in FIGS.
As shown in FIG. In FIGS. 5 to 7, the configuration on the primary side may be the same as that in FIG. 1 and 8 are denoted by the same reference numerals, and description thereof is omitted.

FIG. 5 shows a configuration on the secondary side as a second modification. Also on the secondary side shown in this figure,
By providing the secondary side parallel resonance capacitor C2 for the secondary winding N2, a parallel resonance circuit is formed on the secondary side. In this case, a center tap is provided for the secondary winding N2, and the rectifier diodes DO1, DO2, D
By connecting O3, DO4 and the smoothing capacitors CO1, CO2 as shown in the figure, a set of [rectifier diodes DO1, DO2, smoothing capacitor CO1] and a set of [rectifier diodes DO3, DO4, smoothing capacitor CO2] are obtained. Two sets of full wave rectifier circuits are provided. A full-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] receives a resonance voltage supplied from a secondary parallel resonance circuit and receives a DC output voltage EO1.
Similarly, the full-wave rectifier circuit composed of [rectifier diodes DO3, DO4, smoothing capacitor CO2] also receives the resonance voltage supplied from the secondary-side parallel resonance circuit to generate the DC output voltage EO2.

When such a configuration is compared with the rectification operation on the secondary side described earlier with reference to FIG. 10, for example, when the alternating voltage obtained in the secondary winding N2 has a positive polarity, the rectification diode DO1 (DO3 The operation in which a rectified current flows through
+ M operation mode as 0 (a) (forward mode)
Conversely, when the alternating voltage obtained in the secondary winding N2 has a negative polarity, the operation of the rectifying current flowing through the rectifying diode DO2 (DO4) is the same as that of -M shown in FIG. It can be regarded as the operation mode (flyback method). That is, in this power supply circuit, each time the alternating voltage obtained in the secondary winding becomes positive / negative, the mutual inductance becomes + M
/ -M mode.

FIG. 6 shows a configuration on the secondary side as a third modification. Also on the secondary side shown in FIG.
2, a secondary parallel resonance circuit is formed by providing a secondary parallel resonance capacitor C2, and a bridge rectifier circuit DBR and a smoothing capacitor CO1 are provided for the secondary winding N2. Is provided to obtain the secondary side output voltage EO1.
That is, in this configuration, a full-wave rectification operation is obtained by the bridge rectification circuit DBR on the secondary side. In this case, on the secondary side, independent of the secondary winding N2, another secondary winding N2A is wound around the secondary winding N2A and subjected to a center tap, and then the rectifier diodes DO3 and DO4 are provided. By connecting the smoothing capacitor CO2 as shown in the figure, a secondary output voltage EO2 is obtained by full-wave rectification. However, no parallel resonance capacitor is provided for the secondary winding N2A.

FIG. 7 is a circuit diagram showing a configuration of a power supply circuit according to a fourth modification. As a configuration of the secondary side shown in this figure, one end of the secondary winding N2 is connected to the secondary side ground,
The other end is connected to a connection point between the anode of the rectifier diode DO1 and the cathode of the rectifier diode DO2 via a series connection of the series resonance capacitor Cs1. Rectifier diode D
The cathode of O1 is connected to the positive electrode of the smoothing capacitor CO1,
The anode of the rectifier diode DO2 is connected to the secondary side ground. The negative side of the smoothing capacitor CO1 is connected to the secondary side ground.

As a result, in such a connection form, a voltage doubler full-wave rectifier circuit composed of a set of [series resonance capacitor Cs1, rectifier diodes DO1, DO2, smoothing capacitor CO1] is provided. Here, the series resonance capacitor C
s1 is a rectifier diode DO1, DO2 due to its own capacitance and the leakage inductance component of the secondary winding N2.
To form a series resonance circuit corresponding to the on / off operation.
That is, the power supply circuit of the present embodiment is provided with a series resonance circuit on the primary side for making the switching operation a current resonance type, and a series resonance circuit for obtaining the voltage doubler full-wave rectification operation on the secondary side. A complex resonance type switching converter provided with a circuit is employed.

Here, the [series resonance capacitor Cs]
1, rectifier diodes DO1, DO2, smoothing capacitor CO1]
The double voltage full-wave rectification operation by the set is as follows. When a switching output is obtained on the primary winding N1 by the switching operation on the primary side, this switching output is excited by the secondary winding N2. And a rectifier diode DO
During the period when 1 is off and the rectifier diode DO2 is on, the secondary winding N1 operates in the depolarization mode in which the polarity (mutual inductance M) of the primary winding N1 and the secondary winding N2 is -M. The operation of charging the series resonant capacitor Cs1 with the rectified current IC2 rectified by the rectifier diode DO2 is obtained by the series resonance effect of the leakage inductance of the line N2 and the series resonant capacitor Cs1. And
The rectifier diode DO2 turns off, and the rectifier diode DO1
Is turned on to perform the rectification operation, the polarity (mutual inductance M) of the primary winding N1 and the secondary winding N2.
Becomes + M, and the smoothing capacitor C01 is charged in a state where series resonance occurs in which the potential of the series resonance capacitor Cs1 is added to the voltage induced in the secondary winding N2. As described above, the rectifying operation is performed by using both the positive polarity mode (+ M; forward operation) and the depolarization mode (-M; flyback operation), so that the smoothing capacitor CO1 has: A DC output voltage EO1 corresponding to almost twice the induced voltage of the secondary winding N2 is obtained.

According to the above configuration, in the present embodiment, the secondary side DC output voltage is obtained by performing the double voltage full-wave rectification utilizing the state in which the mutual inductance is in the operation mode of + M and -M. I have to. That is, since the electromagnetic energy due to the primary side current resonance action and the secondary side current resonance action is simultaneously supplied to the load side, the power supplied to the load side further increases accordingly, and the maximum load A large increase in power will be achieved.

Further, by obtaining the secondary side DC output voltage by the voltage doubler full-wave rectifier circuit, it is possible to obtain a level equivalent to the secondary side DC output voltage obtained by the unity voltage rectifier circuit, for example. In this case, the secondary winding N2 of the present embodiment requires only half the number of turns of the conventional winding.
This reduction in the number of turns leads to a reduction in the size and weight of the insulating converter transformer PIT and a reduction in cost. Also in this case, the secondary winding N2 is wound independently of the secondary winding N2,
The center tap is grounded to the secondary winding N2A, and a full-wave rectifier circuit composed of rectifier diodes DO3, DO4 and a smoothing capacitor CO2 is connected to generate a DC output voltage EO2. I have to.

The applicant of the present invention uses a secondary-side series resonance circuit as a composite resonance type switching converter.
Although a configuration having a voltage doubler rectifier circuit has already been proposed, such a configuration can be realized as a modification of the present embodiment. That is, the present embodiment is not particularly limited as the configuration of the secondary resonance circuit.

Further, in the above-described embodiment, the configuration in which the self-excited voltage resonance type converter is provided on the primary side as the composite resonance type switching converter is described. For example, the current resonance type converter is provided on the primary side. The present invention can be applied to a power supply circuit having a parallel or series resonance circuit on the secondary side. Even if the same voltage resonance type converter is provided on the primary side, not only a so-called single-ended configuration having one switching element but also two switching elements alternately as shown in FIG. The present invention can be applied to a so-called push-pull system for switching.

[0057]

As described above, the present invention provides a clamp circuit to be provided in a self-excited resonance type converter by using a diode element such as a Schottky diode and the like, which has no reverse recovery time trr and a resistance element. Form. As a result, a period as a reverse recovery time is not formed after the end of the damper period when the switching element is turned on, but as a result, the switching element is controlled to be on at the time after the end of the damper period. Things. Therefore, even during stable operation, and even under heavy load and low AC input voltage conditions,
The occurrence of an abnormal switching operation is prevented. In addition, this makes it possible to eliminate stress applied to the switching element even under conditions such as when power is turned on. Furthermore, generation of noise due to parasitic vibration during power supply operation is also eliminated.

Further, by eliminating the occurrence of the abnormal operation as described above, it is possible to select a secondary parallel resonance capacitor having a proper capacitance as the secondary side parallel resonance capacitor. The power conversion efficiency can be improved by reducing the levels of the current and the secondary-side resonance current.

Further, according to the configuration of the present invention, when the switching element is turned off, the reverse current flowing through the base of the switching element increases. For this reason, the accumulation time and the fall time of the switching element are shortened, so that the margin for the variation of the accumulation time and the current amplification factor is expanded, and accordingly, the margin of the optimum drive condition is expanded. Also, the rank range when selecting the switching element is expanded, and the working efficiency is improved. In the case of the present invention, it is needless to say that it is not necessary to select the reverse recovery time for the clamp diode.

In the clamp circuit according to the present invention, if a ferrite bead inductor is inserted, for example, in series or in parallel with a resistor element, for example, it may occur when a Schottky diode element having no reverse recovery time trr is turned on / off. It becomes possible to absorb noise.

[Brief description of the drawings]

FIG. 1 is a circuit diagram illustrating a configuration example of a power supply circuit according to an embodiment of the present invention.

FIG. 2 is a waveform chart showing an operation of a main part of the power supply circuit of the present embodiment.

FIG. 3 is a waveform chart showing an operation of a main part of the power supply circuit of the present embodiment.

FIG. 4 is a circuit diagram showing a configuration example of a power supply circuit as a first modification.

FIG. 5 is a circuit diagram showing a configuration example of a power supply circuit as a second modification.

FIG. 6 is a circuit diagram showing a configuration example of a power supply circuit as a third modification.

FIG. 7 is a circuit diagram showing a configuration example of a power supply circuit as a fourth modification.

FIG. 8 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 9 is a cross-sectional view illustrating a structure of the insulating converter transformer of the present embodiment.

FIG. 10 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.

FIG. 11 is a waveform diagram showing an operation of a main part of a power supply circuit according to the prior art.

FIG. 12 is a waveform diagram showing an operation of a main part of a power supply circuit according to the prior art.

[Explanation of symbols]

1 control circuit, 2 clamp circuit, Ci smoothing capacitor, Q1, switching element, PIT isolation converter transformer, PRT orthogonal type control (drive) transformer, Cr parallel resonance capacitor, C2 secondary side parallel resonance capacitor, NC control winding, NB Drive winding, CB resonance capacitor, DSBD Schottky barrier diode, RD resistor, LD ferrite bead inductor

Claims (5)

[Claims] A switching means for intermittently outputting an input DC input voltage, an insulating converter transformer for transmitting an output of the switching means to a secondary side, and the switching means; A self-excited oscillation drive circuit for switchingly driving the switching element by a self-excited method, a primary side resonance circuit inserted so as to make the operation of the switching means a resonance type, and a secondary winding of the insulating converter transformer. By connecting a secondary-side resonance capacitor, a secondary-side resonance circuit that forms a resonance circuit by a leakage inductance component of a secondary winding of the insulating converter transformer and a capacitance of the secondary-side resonance capacitor; Input the alternating voltage obtained to the secondary winding and perform half-wave rectification or full-wave rectification to perform secondary DC. DC output voltage generating means configured to obtain a power voltage, and a constant voltage configured to perform constant voltage control by varying a switching frequency of the switching element according to a voltage level obtained on a secondary side. Control means, formed at least by a series connection of a diode element and a resistance element having no reverse recovery time, and in a negative direction with respect to the self-excited oscillation drive circuit during a period in which the switching element is turned off. A switching power supply circuit comprising: a clamp circuit provided so as to flow a current. 2. The switching power supply circuit according to claim 1, wherein a ferrite bead inductor is inserted into the clamp circuit. 3. The switching power supply circuit according to claim 2, wherein said ferrite bead inductor is connected in series with said resistance element. 4. The switching power supply circuit according to claim 2, wherein said ferrite bead inductor is connected in parallel to said resistance element. 5. The switching power supply circuit according to claim 1, wherein a resistor for limiting current is not inserted between the self-excited oscillation drive circuit and a control input of the switching element.