FI76463B - BELASTNINGSKOMPENSERINGSKRETS FOER EN TELEVISIONSMOTTAGARE. - Google Patents

Description

1 76463

The present invention relates to a regulated power supply and deflection circuit, and comprises a deflection winding; a deflection silicon 5 connected to the deflection winding to generate a sweeping current therein; a return transformer connected to the deflection circuit; a reverse resonant circuit for generating return transformer voltages across the return transformer windings during the return interval; source of energy supply; a load silicon 10 connected to one coil of the return transformer and energized across this coil with a voltage generated to draw load current therefrom; a first inductance connected to the return transformer; switching means for switching the supply energy source to the first inductance, the conductivity of the switching means 15 varying to adjust the energy stored in the first inductance and transferred to the load circuit.

In power supply systems such as "SICOS" described in U.S. Patent Application 3,336,101, filed December 22, 1981, 20 P.E. By Haferle entitled "Regulated Deflection

Circuit "in accordance with EP 0058552, published on August 25, 1982, the control circuit can react to the peak voltage of the horizontal return pulse to achieve good control of the horizontal sweep amplitude and high voltage. When the load is due to the power consumed by high power audio power amplifiers, interfering image-to-width modulation will be observed at power levels ranging around an average of 10 W DC when using an east-west correction circuit as described in U.S. Pat. .1984 PE Ha-ferle entitled “Variable Horizontal Deflection Circuit 35 Capable of Providing East-West Pincushion Correction.” When 2 76463 such east-west correction circuits are not used, an interfering imaging disc becomes detectable even at low on which power bogs.

One feature of the invention is an adjusted power supply and offset circuit that is compensated for the sound load of the return transformer to maintain a relatively stable raster width. The circuit is characterized by a second inductance operatively connected to the return transformer and means for generating a current variation in said second inductance indicating the variation of the load current taken by the load circuit, the return transformer switching the second inductance to the return resonant circuit to control the return time of the second inductance current.

In the drawing: Fig. 1 shows a switch power supply and deflection circuit including a load compensation circuit according to the invention, and Fig. 2 shows waveforms useful in explaining the operation of the circuit of Fig. 1.

In the controlled power supply and deflection circuit shown in Fig. 1, the supply energy source 19 comprises an unregulated AC voltage source 21 connected to the input terminals 23 and 24 of the full wave rectifier bridge 22 and a main filter capacitor C 1 connected to the output terminal 25 and rectifier. An unregulated DC voltage U. is generated across capacitor C.

in 1

A switch power supply 27 is located between the supply energy source 19 and the horizontal return transformer to control the energy transfer between the source and a plurality of load circuits 30 connected to the return transformer windings w2-w4 *.

The DC level locking capacitor 38 is connected between the dotted terminal of the winding w2 and the body ground, which is electrically isolated 35 from the mains gap, which is electrically isolated from the mains supply 21.

The horizontal deflection circuit 39 comprises a horizontal oscillator and control circuit 29 connected at the base of the horizontal terminal transistor 31 to provide a transistor connection at a horizontal frequency. The horizontal end transistor 31 together with the damping diode 32 acts to generate a horizontal sweep current i in a horizontal deflection coil L 1, which is in series with the s-shaping capacitor Cg. When the horizontal end transistor 31 becomes non-conductive at the end of the horizontal line interval, the horizontal deflection coil 10 forms a deflection return resonant circuit together with the deflection return capacitor Cr to generate a deflection-return pulse voltage across the deflection winding.

The deflection return resonant circuit supplies a deflection-15 return pulse voltage to the return transformer windings to generate a return pulse voltage across the other windings of the return transformer. The return pulse voltage generated across the coil is raised by a high voltage coil to warn the high voltage circuit 33 to generate an extreme anode voltage at terminal U for a telescope receiver picture tube not shown in Figure 1.

The voltage generated across the winding in the voltage source 50 is line rectified, i.e., rectified during the horizontal line interval by a diode 34 and filtered by a capacitor 35 to generate a low voltage V of Mata-25. The voltage V acts as a supply voltage 3c for load circuits such as a horizontal deflection circuit, which is also not shown in Fig. 1, and a high-power sound circuit comprising a high-power sound stage 36 which controls the speaker arrangement 37.

The return pulse voltage generated across the coil is transformer coupled to the return transformer coil to interact with the switch power supply 27 in a direct and controlled transfer of energy from the source 19 without intermediate DC conversion. The switch power supply 27 may be similar to that described in the above-mentioned Haferle patent application.

4 76463

The power supply 27 includes controllable bidirectional conductors and Sjotka is connected together by an output terminal 40. A series connection of the inductance of the transformer T1 of the capacitor C2 »and the inductance 5 w1 of the transformer is connected across the switch. The switches and S 1 thus form an alternating phase structure together with the above-mentioned series connection.

During normal operation, the horizontal deflection circuit 39 produces a pulse voltage over the secondary winding of the return transformer during the horizontal return interval, which transformer is thus magnetically connected to a magnetically tightly connected winding. The voltage across the coil is shown in Figure 2A, the solid line of the voltage V r · winding occurring väliottoliit, with remote setpoint pulse voltage is applied to the switching power supply 27 pulssinleveysmodulaattoriohjauspiiriin 28. The control circuit 28 15 pulssinleveysmoduloi vuorovaihekytkemien and S2 activity vaakapäätemuuntajän windings over to adjust the developed paluupulssi- voltage amplitude unregulated sisääntu-lojännitteen V H of the fluctuations, and against load changes caused by load circuits connected to the transformer.

At a controlled time, during each horizontal line interval, for example, at time t in Fig. 2, the switch is made non-conductive and the switch becomes conductive. As shown in Fig. 2d, the current i, the winding w of the transformer T «and the winding of the transformer have an ascending embankment waveform between the moments t ^ and t '^ 25. At the end of the horizontal line interval, near moments t '^, the current i ^ has reached a positive peak by storing a certain amount of energy in the inductance of the winding.

At the beginning of the horizontal return interval near time t'- 1, when the horizontal end transistor 31 becomes non-conductive and when the deviation-return resonant circuit is formed, the control switch is made non-conductive by the control circuit 28 and the switch S2 becomes conductive. Energy transfer is initiated from the coil by the return transformer T1 to the deflection return resonant circuit and return controlled load circuits connected to the return transformer, such as the 35 extreme anode voltage circuit 33 connected to the high voltage coil n 5 to reach a level close to the moment t '^ which is less than the positive peak level of the current near 5 moments.

At the beginning of the horizontal line interval close to the moment t'4 or the corresponding moment t ^, the current i ^ continues to embank downwards, albeit a gentler curve than during the horizontal return interval 10 due to the supply of voltage to the winding w consisting of condenser 3.

from the algebraic sum of the voltages generated over the transducer C0 and the return transformer T. From the moment t1, the energy is transferred to the line rectified voltage source 50 by means of the return-transformer windings and w2. This transferred energy 15 is obtained from the energy previously stored in the capacitor by the control switch. Near time t1, the switch S2 is made non-conductive and the switch becomes conductive to repeat the energy transfer cycle that occurs during each horizontal deflection interval.

Any variation in the load or mains voltage which tends to cause a change in the amplitude of the return pulse voltage V causes the control circuit 28 to change the switch-off time of the switch S2 in a manner which keeps the return pulse amplitude relatively unchanged. The dashed curves in Figure 2 show the average load situation of the return transformer T 1. Switching off of switch S2 is started earlier within the horizontal line interval at time tc. An earlier switch-off of the switch S_ is required o 2 to allow the peak level of the inductance current i1 of the winding w to decrease at the beginning of the horizontal return interval close to the time t '^ so as to adapt to the reduced need for energy transfer to the load circuit. A similar situation applies to changes in the mains supply voltage, where the switch is previously switched off within the horizontal line interval when the mains voltage is high.

The circuitry described heretofore may undesirably tend to change the return time of the return pulse voltages with changes in the loads of the line rectifier voltage source 50 so that, for example, as the load of the high power sound stage 36 increases, the return time tends to increase significantly. The return time increases as the line load increases due to the flywheel effect of the energy produced by the deflection-5 circuit 39, the transformer winding w2 and the capacitor 38. The line voltages across the transformer windings are determined by the voltage across the capacitor 38. A high line load in any of the palms drops the voltage across the capacitor 38. Accordingly, the return voltage, and in particular dV ^ / dt, during the first half of the return line also decreases. This calculates the current i ^ -di / dt between the times t1 ^ and t '^ and delays the passing of the zero point of the current flowing through the winding w2 and also delays but to a lesser extent the center of the return run. As a result of the total-15, the return time has increased. The overall effect on image size is the tendency to increase as the load increases.

The switch power supply of Figure 1 includes a load compensation circuit 30 according to the invention that maintains a constant return pulse duration under varying load conditions. The compensation circuit 30 of the load 20 comprises a diode of an additional compensation coil L1 of the second winding of the transformer and a capacitor C1. A capacitor is connected between the current return terminal 26 of the full wave rectifier bridge 22 and the terminal not marked at the point of the return transformer winding. In the same way, a series arrangement of the winding w ^, the coil and the diode is connected.

Figure 2b shows the voltage V2 generated across the winding of the load compensation circuit. This voltage is a combination of the disconnected DC voltage and the return pulse voltage V at the switch and S2. The current i2 flowing in the series arrangement of the coil w,, the coil L „and the diode D 30 r d 2 1 is shown in Fig. 2c.

The current i2 charges the capacitor C3 to a positive voltage with respect to the capacitor base plate. The voltage is the gain voltage that adds up to the rectified mains voltage V across the capacitor. The switch power supply 27 35 in thus operates approximately 10% of the higher DC voltage above the basket 7 76463 and is thus capable of transferring 20% more power to the load circuits of the television receiver connected to the return transformer.

Under the average load conditions shown by the dashed curves in Figure 2, the voltage V2 increases to a positive voltage level at time tg and the current i2 begins to rise at this point in the embankment. Near time t '^, when the switch is turned off by the control circuit 28, the voltage V2 changes the polarity when the return pulse voltage V is applied to the unmarked terminal of the coil at a point of 10 w. The current i ~ reaches a 2 after reaching a peak value near the moment t '^ increases in its level under the influence of the return pulse voltage part of the voltage V2 and reaches zero near the moment t'2 ·

During the initial part of the return, during the interval t '^ - t' ^, due to the effect of the transformer, the coil 15 is connected in parallel with the winding w. The resulting inductance of the winding w is thus lower during the interval t '^ - t' ^ than during the rest of the return interval t'2 “t'4.

Since the winding w is connected to the winding of the return transformer T.

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20 min, it is in a circuit with the resonant-return circuit 39 of the horizontal deflection circuit. The lower inductance of the winding w during the interval t '^ - t'2 results in a shortened return time compared to the return time that would occur if the switch power supply 27 were without the load compensation circuit 30. This reduced return time varies with the load transformers of the return transformer. compensates for the tendency to change the return time with these variations.

For example, when the load has substantially increased, for example through the additional line load of the sound stage 36, the fixed line waveforms of Figure 2 are valid. The tendency of the return pulse amplitude V to increase is compensated by the control circuit 28 by changing the switch-off time for the time t1 in Fig. 2. The positive level of voltage V2 taken between moments t ^ and t '^ under increased load conditions is greater than during the average load conditions described previously, such as 76463 8. shown in Figure 2b. The current compensating coil L2 thus has a steeper slope, as shown in Fig. 2c, and achieves a larger peak amplitude near the beginning of the return, close to the moment During the return, it takes 5 longer times to flow down to zero, reaching zero at a later time t '

Coil L2 is connected in parallel with the effective inductance of the winding w for a longer period of time within the return interval than the average load conditions prevail. As a result, the return time tends to be shortened under increased load conditions, compensating for the return time tendency to increase with load. Thus, the different inductances are connected in parallel with the deflection return resonant circuit for the different parts within the return interval according to the variations of the return transformer load. Comparing the solid line waveforms of Figure 2c with the dashed waveforms of Figure 2c, it is observed that switching different inductances during return for different durations is automatically performed depending on the control circuit 28 which changes the switch-off time 20 of switch S2 between times t and t7.

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Any fluctuations in load or mains voltage change the switch-off time of switch $ 2 and the start of current i2 via control circuit 28. At high load, the current i2 starts at a later moment, namely t ^. As a result, the amplitude at the end of line 25 near time t 'is higher and the current returns to zero later during the return interval, returning to zero at time t' ^ · Thus, current fluctuations in the compensating coil L2 indicate current fluctuations in the return transformer T ^ load. The result of the operation of the load compensation circuit 30 is a return time which does not vary with different load conditions. In other words, the tendency of the return time duration to change with the variation of the load current is eliminated.

As a general observation, it is observed that the current i2 is a direct return current to the storage capacitor 35 of the input source 19. The loop gain of the control circuit 28 is increased by using the load compensation circuit 30 which allows the control range tg-t ^ to be smaller. Further, the amplitude of the current i at the end of the line 5 is higher when the load compensation circuit 30 is used, thus providing additional load control capability. The peak amplitude of the return pulse voltage is better controlled with respect to load variations when the compensation circuit 30 is used due to the additional control capability 10 provided by this circuit.

The control circuit 28 of Figure 1 controls the peak return voltage. Without the compensation circuit 30, an increase in the load causes an increase in the return time and also an increase in the line voltage, thus the line voltage depends on the return time. Good image stability 15 is obtained when both line and return voltages are kept constant. This is achieved by means of a compensation circuit 30 which maintains a constant return time as the loads vary.

Claims (9)

A regulated energy supply and deflection circuit including a deflection winding (Ly), a deflection circuit (39) coupled to the deflection winding for generating scanning current in the deflection winding, in order to Astadkomma transform tor wrap reversal pulse voltages over windings of sweep transverse transformers (Τχ) during a sweep operation interval, a source (19) for supply energy, a winding how winding reverse transformer (Τχ) is switched on, of the voltage generated above said winding for extracting a load current therefrom, a first switching inductor (kopp) coupled first inductance (Wa), switching means (Si, S2) coupling said source of supply energy to the first inductance, varies to control the energy stored in the first inductance o and transmitted to said load circuit, characterized by a second said inverter wiring transformer (Τχ) functionally coupled second inductance (L2) and means (Wj>, ϋχ, C3) to effect a variation of the current in said second inductance ( L2), which variation is indicative of the variation of the load current taken out by the load circuit (36), wherein the sweep return transformers (lar) couple the second inductance (L2) to the sweep and resonance circuit (Cr) to control the sweep return time in one. similar to said variation of the current in the second inductance (Ι> 2). 2. A circuit according to claim 1, characterized in that said sweeping thread transformer (T1) couples the second inductance (L2) to the sweep feedback resonant circuit (Cr) in such a way that the propensity for the duration of the sweep thread time varies with the variation of the load current. eliminated. 3. A circuit according to claim 2, characterized by a regulator control circuit (28) for controlling the operation of the switching means (S 2, S 2) such that the amplitude of the voltage across a winding (W 1) of the sweep return transformer ( Τι) is constantly poured during the sweep interval. 4. A circuit according to claim 1, characterized in that the means for effecting a current variation include a second transformer (T2) having a first winding (¾¾) connected to the second inductance and a second winding (Wa). connected to the sweep transverse transformer. 5. A circuit according to claim 1, characterized in that said supply energy source (19) is coupled to a first winding (Wj) of the sweep terminal transmitter (T T) and that the deflection circuit (39) is connected to a the second winding (W2) of the sweep return transformer, the second winding being isolated from the first winding (W)). 6. A circuit according to claim 4, characterized in that the switching means comprise first (S S) and second (S2) switches connected to the sweep return transformer (T] J in the receive circuit. 7. A circuit as claimed in claim 6, characterized by the series arrangement of a capacitor (C2), the first inductance (Wa) and a winding (V? I) of the sweep terminal transformer (Τχ) coupled over one (S2) of the Both switches in said switching means.