FI105751B - demodulation - Google Patents

Description

105751

Demodulation method - Demoduleringsforfarande

TECHNICAL FIELD OF THE INVENTION

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The invention relates to a demodulation method applied to TFM modulated signals. The invention further relates to a demodulation method according to the independent claim on the demodulation method.

BACKGROUND OF THE INVENTION

Continuous Phase Modulation (CPM) comprises a class of modulation methods that are simultaneously efficient in both power and bandwidth. In all CPM modulations, the envelope of the RF signal is constant and step 10 varies continuously. A standard envelope allows the use of non-linear amplifiers, which simplifies receiver and transmitter design. Tamed Frequency Modulation (TFM) is a CPM modulation scheme. One of the major advantages of TFM is the requirement for very narrow bandwidth compared to most other modulation schemes. CPM signals can be represented by 15 s (t) = (2Eb / Tb) ° -5cos (2jfQt + <p (i, a) + <p0) (1) where the crossing function <p (t, a) is given by nTb <t <(n + 1) Tb (2) where '{a,} are data symbols of degree M, M even, out of 20 character sets ± 1, ± 3, ...., ± (M-1), ht is modulation index, which may vary from period to period, q {t) is a phase response function, • - · - -------- w «g (t) is a frequency response,

Eb is bit energy, 25 Tb is the bit interval, 2 105751 fQ is the carrier frequency, and φ0 is the arbitrary start phase.

For TFM, M is 2, resulting in data symbols af = ± 1, and ht = 0.5. For the TFM, the bit period is as long as the symbol period.

5 TFM modulation is characterized in that the phase shift of the modulated carrier over one bit period is determined not only by the current bit, but by three binary signals in succession, the coding rule ^ (rt) = j (/ ^ + 2 / „+ /„,) O). where In represents binary data at time t = nTb, and I ′ = ± 1, i.e., for example, -110 corresponds to bit 0, and +1 corresponds to bit 1. From this coding rule, one can see that a phase change of π / 2 is obtained if for three consecutive the bit has the same polarity, and the phase remains constant if the polarities of three consecutive bits alternate. The phase shifts π / 4 are related to bit configurations ++ +, —h, and - h Figure 1 shows a signal state diagram of the TFM. When the correlation of the transmitted bits 15 is included, the TFM signal produces a narrower power spectrum than, for example, the MSK signal, since the phase changes are smoother in the TFM. TFM modulation is further explained in "Tamed Frequency Modulation, A Novel Method to Achieve Spectrum Economy in Digital Transmission", by Fank de Jager and Cornells B. Dekker, 16 Trans.on Comm. Vol. COM-26, N0.5, May 1978, pp. 534 -20,542. CPM modulation is further explained in the Digital Phase Modulation book,

John B. Anderson, Tor Aulin and Carl-Erik Sundberg, Plenum Publishing Corporation, 233 Spring Street New York, N.Y. 10013, pp. 15 - 53.

Direct conversion receivers are receivers that do not use intermediate frequencies to filter and detect received signals. The radio frequency (RF) signal received at the direct conversion receiver-25 is mixed with a local oscillator signal having a frequency corresponding to the carrier frequency of the RF signal. Straight-through * 'nosepieces have many advantages. For example, bandwidth filtering can be done at low frequencies, i.e. audio frequencies, which allow for very narrow bandwidths to be implemented at the sharpest corners. Also, intermediate frequency filters are not required. However, direct conversion receivers have not been used to receive TFM signals due to the inherent problem of the structures of their direct conversion receivers, namely the presence of a DC voltage misalignment at the output of the mixer due to imperfections in the mixer structure. The DC offset 3 105751 is typically due to the leakage of a local oscillator at the mixed RF port, and subsequently the mixing of the leaked signal to the local oscillator signal itself.

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random variations in the leakage signal lead to a relatively slow and randomly varying DC offset signal.

In addition to the DC offset problems inherent in direct downconverting applications, other types of low frequency noise often occur in the baseband signal. An example is low frequency phase noise due to transmitter phase noise or local oscillator phase noise. The problem of DC offset can be thought of as very low frequency phase noise.

10 Figure 2 illustrates the problem caused by DC offset in the detection of TFM modulated data. The signal received without DC deviation has a star pattern in white circles. Vectors S1 and s2, plotted with thick dashed lines, represent signals expressed at two consecutive sampling times. Without DC offset vectors. The expression Si and s2 is straightforward. The DC offset 15 significantly changes the situation. The DC offset appears to shift the signal's star pattern on the IQ chart as shown in black circles. The vectors s1 and s'2 represent the respective sampled signal vectors in the presence of a DC offset. Clearly, any detector optimized for detecting vectors S1 and s2 will have difficulty recognizing vectors s1 and s2. Therefore, the presence of a DC offset 20 easily leads to high error rates of detection.

The only known solution to this problem so far has been to not use a direct downconverter because this DC offset problem does not occur in heterodynamic receivers.

Various coherent detectors or differential detectors have been studied for TFM modulated signals. However, these detectors, based on the phase or phase difference of the signal over one bit period, do not work well when a DC offset occurs, since the phase of a given point in the constellation changes due to DC offset, and likewise the phase difference of two consecutive signal vectors boy because of the keamus.

30 This problem is solved for MSK type modulation. WO94 / 28662 describes the structure of a dual differential detector, which will be explained below. If rn is defined as a signal vector at nTb, then the difference vector between two received vectors over one bit period is defined as 4 105751 Δ „= r„ (4) and the phase difference δφ „between vectors Δ„ and Δ „_, is defined as <5 <ρ„ = Ζ (Δ „) - Ζ (Δ ^) (5) where the decision could be made as follows 5 if O <δφη <135 °, then the transmitted bit bn = 1; if 135 ° <δφη <-135 °, the transmitted bit bn! = b »-i; if -135 ° <δφη <0, then the transmitted bit bn-0.

This method works with MSK because in MSK the phase changes during each bit period.

10 The situation is slightly different when considering the idea of using a dual differential detector for TFM modulation. For TFM, δφη is not a sufficient measure to perform the decision, since the difference vector Δ „may be zero. As explained above, in TFM modulation, the phase does not change according to the given information bit patterns. Therefore, δφη may also be zero for some bit patterns, which makes it impossible to make a decision on the bits based on the value of δφη.

SUMMARY OF THE INVENTION

It is an object of the invention to provide a demodulation method for demodulating TFM signals. Another object of the invention is to provide a demodulation method which is capable of demodulating a TFM signal with DC offset or other types of low frequency noise. It is also an object of the invention to provide a de-modulator structure capable of demodulating a TFM signal in the presence of a DC offset.

The objectives are achieved by using the difference vector of the two received sample vectors as the decision variable to express the transmitted data symbols.

The demodulation method according to the invention is characterized by what is defined in the characterizing part of the independent method requirement for the demodulation method. The data transfer method according to the invention is characterized by what is defined in the characterizing part of the independent method claim 5 105751. The demodulator array according to the invention is characterized by what is defined in the characterizing part of the independent claim concerning the demodulator array. The mobile communication device according to the invention is characterized by what is defined in the characterizing part of the independent claim 5 relating to the mobile communication device. The base station according to the invention is characterized by what is defined in the characterizing part of the independent claim concerning the base station. The radio link system according to the invention is characterized by what is defined in the characterizing part of the independent claim of the radio link system. Other dependent embodiments of the invention are disclosed in the dependent claims.

In order to explain the inventive method, the following quantities are first defined: = i (6) where A 1 is the difference vector between two received vectors over one bit period; (7) wherein Δ2 is the difference vector between the received vectors over two bit periods; and δφ „= Ζ (Αι„) - Α ^ Γ ~ (8) where δφη is the phase difference between two difference vectors Δ1 * and Δ '„- ι. For the TFM .. 20 determines the phase change over three bits transmitted in a single bit sequence. In addition, the quantities Δ2 and δφη are determined by four successive information bits.

Figure 3 illustrates possible state changes due to two consecutive symbols. Since, according to equation (3), the phase change of one symbol can be: 0, ± π / 4, or ± π / 2, the sequential setting of two symbols results in a phase change of one of the following {0, π / 4, π / 2, 3π / 4, π) or any of the following {0, -π / 4, -π / 2, -3π / 4, -π}. Therefore, the difference between two consecutive signal vectors may be one of the vectors shown in Figure 3. Figure 3 illustrates the case {0, π / 4, π / 2, 3π / 4, π}. The length of the resolution vector may be z, s, ll12, or l3. Length z tar-30 represents the difference vector zero, length s denotes the difference vector from the point of the constellation to the adjacent point, length lj denotes the length of the difference vector passing one point of the star pattern on its way to the point of the constellation. 105751 represents the length of the difference vector passing by two points of the constellation on its way to the point of the constellation, and the length l3 denotes the length of the difference vector passing through two points of the constellation on the point of the constellation.

The following table lists all possible data patterns for four consecutive 5 data bits and the corresponding values of [A ^ J, | Δ2 „|, and δφη. As an example of how the values in the table are derived, consider bit pattern 0110. According to equation (3), bit string 011 results in a phase change π / 4. This phase change corresponds to a difference vector of length s in Figure 3, which results in a large value of s. Similarly, bit string 110 again results in a phase change of π / 4, whose product of the product becomes a large value of s. the sum is then a vector Δ2 of length lly as shown by equation (7). Then the difference between the angles of two consecutive difference vectors Δ1 ^, and Δ1η is π / 4.

b „-3 b„ .2 b „.jb„ A'n_j Δ1, Δ21 & Φη 0 0 0 0 l} hh -π / 2 0 0 0 1 hs l2 -3π / 8 ΟΟΙΟί 05 0011 5 50 π 01000 5 5 0 10 10 0 0 0 110 5 s Ij π / 4 0 111 5 U h 3π / 8 ... 1 0 0 0 5 h l2 -3π / 8 1 0 0 1 5 5 lj -π / 4 1 0 1 0 0 0 0 1 0 1 1 0 5 5 1 1 0 0 5 5 0 π 1 1 0 1 5 0 5 1110 hs l2 3π / 8 1111 hhh π / 2

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It can be seen from the above table that δφ „cannot be computed for six bit patterns {0010, 1101, 0100, 1011, 0101, 1010} because one of the vectors A′n_j and Δ1 or both are zero vectors.

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From the above table, it can be seen that the complementary bit patterns have equal lengths of difference vectors, i.e., | Δ ^, |, | Δ'Π |, and | Δ21 have the same value. For example, the difference vectors corresponding to bit pattern 0010 have the same value as the difference vectors corresponding to bit pattern 1101, as can be seen in the table above. Therefore, on the basis of complementary difference vectors alone, the receiver cannot distinguish complementary bit patterns. This problem can be overcome by using differential coding. In differential encoding, two consecutive bits having the same value correspond to bits "0" and two consecutive bits having different values correspond to bits "1". For example, 10 differentially encoded bit patterns 0010 and 1101 correspond to the same decoded bit pattern 011. In general, any complementary bit patterns produce the same differentially encoded bit patterns. It can be seen from the above table that the bit vectors 0010 and 1101 corresponding to the decoded bit pattern 011 have equal lengths of difference vectors. Similarly, when the transmitted data is encoded by differential coding, the length of the difference vectors provides sufficient data to represent the received symbols. Therefore, in preferred embodiments of the invention, the quantities | A! "| and | Δ2η | can be used as a decision variable to represent received symbols. Similarly, any quantity of | Δ! „| or | Δ21 or both substantially independent variables can be used as a decision variable.

In a preferred embodiment of the invention, | Δ21 is used as the decision variable in combination with the 4-state trellis in Figure 4. The length of the corresponding difference vector over two bit periods, i.e., the value of | Δ2 |: η is plotted for each path of the trellis. In such an embodiment of the invention, the weights of the different paths in the Viterbi algorithm can be determined as follows: 1 _ _________ 105751 W (dn-2'dn-ΐΛ) = MO, 0.0) = || Δ * | “/ 3 | * Μ0,0,1) = || δ2 | - / 2 | * »(0,1,0) = | δ * | * MO, ι, ΐ) = | δ2„ | - $ 5 Μ1Α0) = || δ2 "| - / 2 |" (9) mmu) = And * | - / i | * MU, ο) = | δ2 "| -5 Μι, ι, ι) = || δ2 | Ρ Using these formulas, the weights of the different paths should have the weight of the right path 10 close to zero, and ideally it is zero, so the correct path can be found in the Viterbi algorithm by searching for the path with the lowest cumulative weight. For example, weights may be formulated such that the weight of a given path has a maximum when the path is the correct path, in which case a cumulative sum of the path weights would be sought in the Viterbi-algo-15 rhyme.

The exponent k can be adjusted to achieve optimum performance. For example, the value of k may be 1, whereby the weights are the absolute value of the difference between the length of the difference vector received over two bit sequences and the ideal length of the vector corresponding to the path in question. The value k may additionally be, for example, 2, whereby the weights are the square of the absolute value of the difference between the length of the difference vector between the vectors received over two bit periods and the ideal length of the vector corresponding to the path in question. Other A-values are also possible, and the optimum can be found, for example, by simulation or experimentation.

V The invention is not limited to | Δ2 | to use as a decision variable and to use a 4-state trellis. In another preferred embodiment, for example, a variable is used as a decision variable along with a 2-state trellis.

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BRIEF DESCRIPTION OF THE DRAWINGS

The invention will be explained in more detail below with reference to the accompanying drawings, in which Fig. 1 illustrates a star pattern of TFM modulation symbols, 5 Fig. 2 illustrates the effect of a DC offset, Fig. 3 illustrates difference vectors computed over two bit periods, Fig. 4 illustrates a in a preferred embodiment, Figure 5 is a block diagram of a receiver structure 10 according to a preferred embodiment of the invention, Figure 6 is a block diagram of a preferred embodiment of the invention, and Figure 7 is a block diagram of another preferred embodiment of the invention.

In the pictures, the same reference numerals are used for like entities. Figures 1, 2, 3 and 4 were previously described.

15 DETAILED EXPLANATION

Fig. 5 shows a block diagram of a receiver structure according to a preferred embodiment of the invention. Such a receiver structure can be used, for example, in cellular networks and base stations of cellular networks. The receiver structure comprises a sampling section 200 and a demodulator section 300. The sampling section receives 20 RF signals and converts the received signal into baseband I and Q signals by means of mixers 210a, 210b. The local oscillator 205 produces a signal which is mixed with the received RF signal, and the phase shifter 206 provides a 90 ° phase shift to the local oscillator signal which is applied to the mixer 210b. The output signals of mixers 201a and 210b are applied to matched filters 215a, 215b, whose filtering characteristics are optimized for filtering TFM signals. The output signals of the filters 215a, 215b are applied to switch elements 225 which take samples of the I and Q signals in the 90 degree phase shift. Each corresponding pair of I and Q samples defines a signal sample vector. The switching elements 225 are controlled by a sampling oscillator 220. The sampling section 200 is an example of a structure 30 that can be used in a receiver structure according to the invention. Many other structures can be utilized in an unknowing manner by those skilled in the art to produce a down-converted, sampled signal. The invention is not limited to the use of the structure 200 of the sampling part shown in Fig. The switching elements 225 that perform the sampling typically comprise analog / digital converters or other types of sampling means.

The resulting I and Q samples are applied to demodulator section 300. Both I and Q signal paths comprise two summing units 260a, 260b, 260c, 260d and two delay units 250a, 250b, 250c, 250d. First pair of summing and delaying units 250a, 260a; 250c, 260c produces a difference vector between consecutive bit sequences, after which a second pair of summing and delay units 250b, 260b; 250d, 260d sum the two successive difference vectors to produce a difference vector Δ2 over two bit periods. The structure of the summing and delaying units shown in Figure 5 implements the combination of equations (4) and (7): Δπ = Δ '„+ Δπ-ι = rn ~ r„ -i + rn_ 1 - v2 (10) 15 Equation (10) can be simplified , so that: Δ »= rn ~ V1 + V1 - rn-2 =? '~ ^ -2 (10 so that the difference vector Δ2 over two bit periods can be computed equally using one sum block and one delay block for both I and Q- the signal path, whereby the delay block produces a delay 2 Tb, and the delayed sample stream is subtracted from the non-delayed 20 sample streams in the summing block according to equation (11).

The differences between I and Q signals over two bit periods are applied to a vector length determination block 270 which calculates a value of Δ21. the resulting array of values | Δ21 is applied to a Viterbi detector block 280, which determines the actual transmitted data bits and outputs the transmitted data bits to the output of the structure. In the example of Fig. 5, the Viterbi detector-25 block may advantageously use the 4-state gentle and path weights in Fig. 4 obtained from Equation (9).

Fig. 6 is a block diagram of a digital mobile communication device according to a preferred embodiment of the invention. The mobile communication device comprises a microphone 301, a keyboard 307, a display 306, a headset 314, an antenna duplexer or switch 308, an antenna 309, and a control unit 305, all of which are typical components of conventional mobile communication devices. The mobile communication device further typically comprises transmission and reception blocks 304, 311. The transmission block 304 comprises the functions required for speech and channel coding, encryption and modulation, as well as the necessary 11 105751 RF circuits to amplify the signal for transmission. Receiver block 311 comprises the necessary amplifier circuits and functions required for demodulating and decrypting the signal, as well as for channel and speech decoding. The signal produced by microphone 301 is amplified in amplifier stage 302 and converted to digital 5 by analog / digital converter 303, after which the signal is applied to transmitter block 304. The transmitter block encodes a digital signal and produces a modulated and amplified RF signal, or via switch 308. Receiver block 311 demodulates the received signal and decrypts and channel encodes. The resulting speech-10 signal is converted to analog form by a digital / analog converter 312, the output signal of which is amplified at amplifier stage 313, after which the amplified signal is applied to a headset 314. The control unit 305 controls the through. In the mobile communication device 15 according to the invention, the mobile communication device comprises a demodulation part 300 which performs the demodulation according to the invention. In some embodiments of the invention, the demodulation part 300 preferably has the structure shown in Figure 5. However, other structures may be used to demodulate the TFM signal according to the invention.

Figure 7 shows an example of an embodiment of the invention. In the example of Figure 7, a demodulator 300 of the invention is used to demodulate TFM signals received from mobile communication means 350 at least in some base stations 360 of the mobile network. In addition, Figure 7 illustrates a base station controller 370 that controls base stations 360, and two radio link units 371 for connecting a base station controller: 370 to another mobile communication network 380. Figure 7 also illustrates another preferred embodiment of the invention, namely, using demodulators 300 according to the invention. The demodulation method according to the invention is very advantageously used in continuous high speed communication where continuous demodulation is required. High speed radio links are one example of such a preferred embodiment of the invention.

The method according to the invention can advantageously be used to eliminate the effects of the DC deviation. In addition, the method can also be used to demodulate TFM signals in the presence of other types of low-frequency noise, such as low-frequency phase noise or low-frequency thermal noise. The invention can advantageously be applied directly to downconverting receivers.

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In light of the above description, it will be apparent to one skilled in the art that various modifications may be made within the scope of the invention. Although a preferred embodiment of the invention has been described in detail herein, it should be apparent that many variations and modifications thereto are possible, all within the real spirit and scope of the invention.

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Claims (12)

1. Demodulation method for demodulating TFM modulated signals, characterized in that the method comprises the steps of v - forming signal sample vectors of a received signal, 5. calculating difference vectors on the basis of said signal sample vectors, and - detecting received data symbols on the basis of the length of said difference . Demodulation method according to claim 1, characterized in that each said difference vector is calculated as the difference between two successive signal sample vectors. Demodulation method according to claim 1, characterized in that each said difference vector is calculated as the difference between two signal sample vectors received at two bit intervals between said two signal sample vectors. Demodulation method according to claim 1, characterized in that the transmitted data is determined on the basis of the length of said difference vectors using a Viterbi algorithm. Method for data transmission, characterized in that the method comprises the steps of: - differentially encoding data to be transmitted; 20. modulating differentially encoded data using TFM modulation; - transmitting modulated data; - receiving modulated data; calculating difference vectors on the basis of said signal sample vectors, and detecting received data symbols on the basis of the length of said difference vectors. Method for data transmission according to claim 5, characterized in that each said difference vector is calculated as the difference between two consecutive signal sample vectors. ______ 30 Method for data transmission according to claim 5, characterized in that each said difference vector is calculated as the difference between two signal sample vectors received at two bit intervals between said two signal sample vectors. 16 105751 Method for data transfer according to claim 5, characterized in that transmitted data is determined on the basis of the length of said difference vectors using a Viterbi algorithm. A demodulator system for demodulating TFM modulated signals, characterized in that the system comprises - a block for calculating difference vectors, - a block for determining the length of difference vectors, and - a Viterbi detector block. Mobile communication means, characterized in that the mobile communication means comprises a demodulator system comprising - blocks for calculating difference vectors, - a block for determining the length of difference vectors, and - a Viterbi detector block. 11. Base station for a mobile communication network, characterized in that the base station 15 comprises a demodulator system comprising - blocks for calculating difference vectors, - a block for determining the length of difference vectors, and - a Viterbi detector block. 12. Radio link system, characterized in that the radio link system comprises a demodulator system comprising - blocks for calculating difference vectors, - a block for determining the length of difference vectors, and - a Viterbi detector block. * · ^ I