EP2007028B1 - Recepteur - Google Patents

Recepteur Download PDF

Info

Publication number
EP2007028B1
EP2007028B1 EP07741040.5A EP07741040A EP2007028B1 EP 2007028 B1 EP2007028 B1 EP 2007028B1 EP 07741040 A EP07741040 A EP 07741040A EP 2007028 B1 EP2007028 B1 EP 2007028B1
Authority
EP
European Patent Office
Prior art keywords
matrix
receiver
transmission path
common factor
formula
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Not-in-force
Application number
EP07741040.5A
Other languages
German (de)
English (en)
Other versions
EP2007028A9 (fr
EP2007028A4 (fr
EP2007028A2 (fr
Inventor
Allen Yuan
Holly He
Thanh Bui
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Lenovo Innovations Ltd Hong Kong
Original Assignee
NEC Corp
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by NEC Corp filed Critical NEC Corp
Publication of EP2007028A2 publication Critical patent/EP2007028A2/fr
Publication of EP2007028A9 publication Critical patent/EP2007028A9/fr
Publication of EP2007028A4 publication Critical patent/EP2007028A4/fr
Application granted granted Critical
Publication of EP2007028B1 publication Critical patent/EP2007028B1/fr
Not-in-force legal-status Critical Current
Anticipated expiration legal-status Critical

Links

Images

Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/69—Spread spectrum techniques
    • H04B1/707—Spread spectrum techniques using direct sequence modulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L25/03012—Arrangements for removing intersymbol interference operating in the time domain
    • H04L25/03019—Arrangements for removing intersymbol interference operating in the time domain adaptive, i.e. capable of adjustment during data reception
    • H04L25/03038—Arrangements for removing intersymbol interference operating in the time domain adaptive, i.e. capable of adjustment during data reception with a non-recursive structure
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03891—Spatial equalizers
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L2025/03592—Adaptation methods
    • H04L2025/03598—Algorithms
    • H04L2025/03605—Block algorithms

Definitions

  • the present invention relates to an equalizer and a receiver including such an equalizer, and more particularly to an equalizer used in a radio communication system utilizing a W-CDMA (Wideband Code Division Multiple Access) method, a receiver including such an equalizer, and an arithmetic method.
  • W-CDMA Wideband Code Division Multiple Access
  • this type of W-CDMA methods is a radio communication method in which different codes and a plurality of carrier waves are multiplied together and multiplexed, and a spread spectrum is then performed, and is used in radio communication such as cellular phones.
  • radio communication such as cellular phones.
  • global standards used in the third-generation mobile communication are defined for the W-CDMA methods by 3GPP (3 rd Generation Partnership Project).
  • the STTD Space Time block coding based Transmit antenna Diversity
  • the TSTD Time Switched Transmit Diversity
  • the closed loop transmit diversity mode I (CLM1) method are defined as transmit diversity methods by the 3GPP.
  • the STTD method is a method of encoding the same transmitting data and transmitting them simultaneously from two antennas. This method can reduce the level fluctuation of receiving data.
  • the TSTD method is a method of switching transmitting antennas at each slot of a radio frame.
  • the CLM1 method is a method of operating based on downlink information fed back to a base station from a mobile station.
  • this type of communication systems includes a system in which base stations that transmit and receive data by either one of the aforementioned transmit diversity methods and base stations that transmit and receive data by another transmit diversity method are mixed. Furthermore, in another system, a base station does not use any of the aforementioned transmit diversity methods.
  • Patent Document 1 discloses a receiving apparatus having a plurality of CPICH (Common Pilot Channel) receiving parts corresponding to a plurality of antennas, a plurality of SCH (Synchronization Channel) receiving parts for demodulating SCH with use of propagation-path estimates obtained by the CPICH receiving parts, and a judging part for judging with use of the demodulated SCH whether or not the STTD method is used for CCPCH (Common Control Physical Channel) of the received signals.
  • CPICH Common Pilot Channel
  • SCH Synchrononization Channel
  • CCPCH is demodulated with use of a propagation-path estimate obtained by each CPICH receiving part.
  • STTD method is used for CCPCH, a STTD demodulation process is performed on the CCPCH.
  • Patent Document 2 discloses a radio transmitting and receiving apparatus having a sampling device, a channel estimation device, a channel equalizer, and a despreader . Furthermore, the channel equalizer equalizes a received vector with use of a channel response matrix (H) and a noise variance ( ⁇ 2 ) obtained from the channel estimation device and outputs a spread symbol vector. The despreader despreads the spread symbol vectors obtained from the channel equalizer with using codes of the received signals and produces estimated symbols.
  • H channel response matrix
  • ⁇ 2 noise variance
  • Patent Document 3 discloses a filter coefficient arithmetic method capable of reducing the number of operations relating a filter coefficient of an FIR filter.
  • Patent Document 1 can improve demodulation characteristics of SCH by selecting a proper input signal in an SCH demodulation process system, irrespective of whether or not the TSTD method is used for SCH, after slot synchronization, frame synchronization, and scrambling code identification.
  • Patent Document 1 only judges the use of the STTD method and fails to disclose processes subsequent to the selection of the input signal. Furthermore, Patent Document 1 also fails to describe an arithmetic method for a filter coefficient of an FIR filter.
  • a code block diagonal matrix is obtained by processing a signal code using a filter block Fourier transform (FT) described in Patent Document 2, a channel response of the received signal is estimated and extended to produce a channel response block diagonal matrix. Moreover, the received signal sampled is processed with use of a combination of the channel response block diagonal matrix and the aforementioned code block diagonal matrix with a Cholesky algorithm. Furthermore, a block inverse FT is performed based on the result of the Cholesky algorithm to produce spread symbols. The spread symbols are despread to recover the received signals.
  • FT filter block Fourier transform
  • Patent Document 2 discloses that an inverse FT is performed to recover the received signals but fails to disclose any means for coping with different transmit diversity methods or any chip equalizer including an FIR filter. Therefore, Patent Document 2 fails to describe an increase of the computational complexity caused by computation for a filter coefficient of an FIR filter and fails to show any method for simplifying computation for a filter coefficient.
  • this type of CDMA receivers uses a chip level equalizer (CLE) for equalizing and detecting PN sequence of spread sequence.
  • CLE chip level equalizer
  • This chip equalizer produces a chip estimate from input data by controlling a coefficient of an FIR filter.
  • Matrix inverse operation is used to compute a coefficient of an FIR filter.
  • the matrix inverse operation requires operations such as matrix decomposition operation, forward substitution, and backward substitution. Although those operations are relatively easy in a case of a single transmitting antenna, the computation becomes very complicated in a case of the transmit diversity or the like, where a plurality of antenna are used.
  • Patent Document 3 discloses an arithmetic method capable of reducing the amount of computation required for a coefficient of an FIR filter. However, Patent Document 3 does not consider the use of transmit diversity methods.
  • An object of the present invention is to provide a receiver capable of reducing the amount of computation required for coefficients of FIR filters in a case of a transmit diversity method.
  • the present invention seeks to provide a receiver capable of symbol recovery by performing different operations in cases of an STTD method and a CLM1 method.
  • the present invention seeks to provide an arithmetic method capable of reducing the amount of computation required for filter coefficients of FIR filters in a case of a transmit diversity method.
  • a receiver having a plurality of FIR filters and being capable of communicating with a transmitting part using a transmit diversity method through a transmission path with use of the plurality of FIR filters, characterized by comprising a processing arithmetic circuit operable to compute a filter coefficient w of each of the FIR filters with a common factor (c 0 ) and output the filter coefficients w to the plurality of FIR filters.
  • a receiver characterized in that the common factor c 0 is computed by performing forward substitution and backward substitution on a lower triangular matrix L obtained by performing Cholesky decomposition on the formula (2).
  • the filter coefficient w is derived by performing Hamilton transpose on the common factor c 0 obtained by the formula (5) to obtain c 0 H and then performing the computation of the formula (1).
  • the present invention makes it possible to obtain a receiver capable of computing filter coefficients of a plurality of FIR filters with a relatively small amount of computation. As a result, this type of receivers can be put into practice.
  • FIG. 1 there is shown an equivalent of a communication system according to the present invention, i.e., a communication system using a transmit diversity method.
  • the illustrated communication system is formed by a base station or the like.
  • the communication system includes a transmitting part 11 having a plurality of transmitting antennas (two transmitting antennas in this example) Tx1 and Tx2, a receiving part 12 formed by a mobile station such as a cellular phone, and a transmission path 13 between the transmitting part 11 and the receiving part 12.
  • the illustrated transmitting part 11 is characterized by an encoding function block 21 for encoding a transmitting symbol S in accordance with the STTD method or adding a weight to a transmitting symbol S in accordance with the CLM1 method and outputting the encoded signals and two spread function blocks 22 and 23 for spreading the outputs of the encoding function block 21 with a spread code.
  • This embodiment only describes a case in which the outputs of the encoding function block 21 are spread with a PN sequence.
  • the signals spread with chip signals in the spread function blocks 22 and 23 are outputted from the two transmitting antennas Tx1 and Tx2 to the transmission path 13.
  • the transmitting part 11 shown in FIG. 1 performs transmission via the two transmitting antennas Tx1 and Tx2 with the transmit diversity method.
  • the illustrated transmitting part 11 will be described as performing transmission with selectively using the STTD method and the CLM1 method. Nevertheless, a transmitting part 11 using a different transmit diversity method may be present in the communication system.
  • the TSTD method it is not necessary to employ the present invention because its transmission rate is low. Therefore, this embodiment describes a case in which the STTD method and the CLM1 method are selectively used.
  • the transmission path 13 can be illustrated equivalently as shown in the drawing.
  • the transmission path 13 using a transmit diversity method can be represented by a first multipath for propagating a transmitting signal from the transmitting antenna Tx1, a second multipath for propagating a transmitting signal from the transmitting antenna Tx2, and a noise added in those multipaths.
  • the first and second multipaths are equivalently represented by spread channel response matrices H 1 and H 2 , a noise added in the transmission path 13 (a variance a 2 of a noise in this example), the spread channel response matrices H 1 and H 2 , and the noise.
  • the input data r inputted to the receiving part 12 can be demodulated and recovered by forming an inverse system to the transmission path 13.
  • the illustrated receiving part 12 is characterized solely by an equalizing device 40 according to the present invention.
  • the equalizing device 40 recovers a symbol S from the input data r received via the transmission path 13.
  • the equalizing device 40 according to the present invention utilizes direct matrix inverse approach and computes filter coefficients of FIR filters by a direct matrix inverse method.
  • the equalizing device 40 according to the present invention has a configuration capable of recovering and demodulating signals encoded by a plurality of different transmit diversity methods (the STTD method and the CLM1 method in this example) so as to correspond to those transmit diversity methods and can compute filter coefficients necessary for the plurality of transmit diversities with a relatively small amount of computation.
  • the equalizing device 40 according to the present invention will be described in detail with reference to FIG. 2 .
  • the equalizing device 40 shown in FIG. 2 is supplied with the input data r and also with channel estimates from a channel estimation device, which is not shown.
  • the input data r are normal communication data after performing cell search operations such as slot synchronization, frame synchronization, code group identification, and scrambling code identification.
  • the equalizing device 40 estimates a channel response of the transmission path 13 to a signaling pulse, produces an inverse system to the transmission path 13, and connects the inverse system to the transmission path 13 in series, thereby compensating the interference between the multipaths.
  • the equalizing device 40 implements an inverse system by FIR filters and operates as a chip equalizer configured to accurately reproduce a chip signal corresponding to a PN sequence in the input data r.
  • the channel estimation device provided on the receiving part 12 estimates channel response matrices H 1 and H 2 of the two multipaths forming the transmission path 13.
  • the estimated channel response matrices H 1 and H 2 as results of the estimation are supplied as channel estimates to the equalizing device 40.
  • an estimate ⁇ of a noise figure (scalar) ⁇ which represents a noise, is assumed to be given.
  • the computation for an estimate of a noise figure is described by Japanese laid-open patent publication No. 2006-54900 and is not be described herein in detail.
  • the two channel response matrices H 1 and H 2 corresponding to the two multipaths which have been estimated by the channel estimation device are represented by H g .
  • H g is 1 or 2.
  • the input data r are supplied through L multipaths from the gth transmitting antenna and that its estimate is represented by h l g .
  • / 0, 1, 2,... L-1.
  • a channel estimate can be represented by the following channel response matrix H g.
  • H ⁇ g h 0 g 0 ⁇ 0 h 1 g h 0 g ⁇ ⁇ ⁇ h 1 g ⁇ 0 h L - 1 g ⁇ ⁇ h 0 g 0 h L - 1 g ⁇ h 1 g ⁇ ⁇ ⁇ 0 ⁇ 0 h L - 1 g
  • the channel response matrix H g estimated by the channel estimation device is supplied to a channel matrix arithmetic function block 42 of the illustrated equalizing device 40.
  • a gain matrix G corresponding to the channel response matrix H g is computed in the channel matrix arithmetic function block 42.
  • H H is a correlation matrix of the estimated channel response matrix H (i.e., a Hamilton transposed matrix of H) and I is a unit matrix.
  • a gain matrix is usually computed individually for a multichannel corresponding to each transmitting antenna.
  • a gain matrix is individually computed for each multichannel to obtain an inverse matrix, the amount of computation becomes so large as to make the practical application to cellular phones or the like difficult.
  • the transmission path 13 is represented as shown in FIG. 1 , and a single gain matrix G is used to form an inverse system.
  • G H ⁇ 1 H ⁇ H ⁇ 1 + H ⁇ 2 H ⁇ H ⁇ 2 + ⁇ ⁇ ⁇ I
  • the superscript letter H means a Hamilton transposed matrix of a matrix.
  • H 1 and H 2 are estimates of channel response matrices of the two multipaths
  • H 1 H and H 2 H are Hamilton transposed matrices of H 1 and H 2 , respectively
  • I is a unit matrix
  • ⁇ is an estimate of a noise figure.
  • the estimate of a noise figure can be computed by the aforementioned method.
  • the channel response matrix arithmetic function block 42 of the equalizing device 40 shown in FIG. 2 receives estimates H 1 and H 2 of the channel response matrix and an estimate ⁇ of a noise figure which are provided by the channel estimation device, computes H 1 H and H 2 H , and computes a gain matrix G in accordance with the formula (8).
  • the channel response matrix arithmetic function block 42 is a block for performing the above operations and outputs the Hamilton transformation matrices H 1 H , H 2 H , and a gain matrix G, which have been computed by the block 42.
  • the gain matrix G is supplied to a Cholesky decomposition function block 44, which performs a process of Step 1.
  • the Cholesky decomposition function block 44 performs Cholesky decomposition on the gain matrix G (Step 1) to compute a lower triangular matrix L and an upper triangular matrix U of the gain matrix G.
  • the lower triangular matrix L computed by the Cholesky decomposition function block 44 is supplied to a forward substitution function block 46, which performs a process of Step 2, and a backward substitution function block 48, which performs a process of Step 3.
  • forward substitution (Step 2) and backward substitution (Step 3) as described later are performed to compute solutions ( d and c 0 ) of the system equation.
  • Step 4 the solution c 0 obtained by the backward substitution function block 48 (i.e., Step 3) and matrices H 1 H and H 2 H computed by the channel matrix arithmetic function block 42 are supplied to a filter coefficient computation function block 50.
  • This function block 50 computes weight vectors (w 1 and w 2 in this example) representing filter coefficients by using the aforementioned solution c 0 and matrices H 1 H and H 2 H .
  • the equalizing device 40 shown in FIG. 2 includes a plurality of FIR filters (first and second FIR filters 52 and 54 in this example) corresponding to a plurality of transmitting antennas Tx1 and Tx2 used in a transmit diversity method.
  • Each of the FIR filters 52 and 54 is supplied with the input data r and is also supplied with the weight vector w 1 or w 2 as a filter coefficient from the filter coefficient computation function block 50.
  • An inverse system of the transmission path 13 is implemented by suitably changing the filter coefficients.
  • the optimum weight vectors w 1 and w 2 for the first and second FIR filters 52 and 54 are represented by the middle column of the following weight matrix W g .
  • W ⁇ g G - 1 ⁇ H ⁇ g H
  • the input data r supplied to the first and second FIR filters 52 and 54 are equalized by the FIR filters 52 and 54 and then supplied to first and second despread function blocks 56 and 58, respectively.
  • first and second despread function blocks 56 and 58 respectively.
  • the filter coefficients of the FIR filters 52 and 54 are optimized by the optimum weight vectors w 1 and w 2 , respectively, then despreading operation is performed while chip equalization is performed.
  • Despread symbols S1 and S2 are outputted from the despread function blocks 56 and 58.
  • the despread symbols S1 and S2 outputted from the despread function blocks 56 and 58 are supplied to a processing part 60.
  • the processing part 60 performs a process corresponding to the STTD method or the CLM1 method to produce a demodulation symbol S .
  • the gain matrix G is a Hamilton matrix and is a real number.
  • G -1 is computed using the following formulas (9) and (10) by procedures of steps a, b, and c.
  • a matrix G and its inverse matrix G -1 have the following relationship.
  • I is a unit matrix and D is represented by the following formula.
  • L H ⁇ G - 1 D
  • the computation is performed using the formulas (9) and (10) by the following steps a, b, and c.
  • Step a Cholesky decomposition is performed on a channel response matrix G to obtain a lower triangular matrix L.
  • Step b A solution for the formula (9) is computed.
  • forward substitution is performed on the lower triangular matrix L to obtain a matrix D. Since a unit matrix I is given, the matrix D can be computed.
  • Step c The solution for the formula (10) is computed. In this case, backward substitution is performed on the matrices D and L H to obtain an inverse channel response matrix G -1 .
  • the equalizing device 40 according to the present invention as shown in FIG. 2 performs the following arithmetic operations so as to reduce the amount of computation in the steps b and c from O(N 3 ) to O(N 2 ) and is thus applicable to a practical communication system.
  • the equalizing device 40 has the following basic function blocks as described above.
  • a row vector d is obtained.
  • the computation for the row vector d can remarkably be simplified.
  • FIG. 3 is a conceptual diagram showing a forward substitution step performed in this block. It can be seen that the computation to obtain a row vector d is simplified by using the formula (11) instead of the formula (9).
  • the equalizing device 40 equalizes transmitting data from the two transmitting antennas Tx1 and Tx2. Therefore, the illustrated equalizing device 40 has two FIR filters 52 and 54.
  • the filter coefficients (w 1 , w 2 : w g ) of those two FIR filters 52 and 54 are computed by the filter coefficient computation block 50 and supplied to the FIR filters 52 and 54, respectively (Step 5).
  • the common c 0 H is used to compute the two filter coefficients.
  • the computation for the filter coefficients is remarkably simplified.
  • the FIR filters 52 and 54 filter the input data r with the filter coefficients w g updated temporally (Step 6).
  • each of the despread function blocks 56 and 58 performs despreading to obtain an estimated symbol value (Sg) corresponding to the gth transmitting antenna (Step 7) and outputs the estimated symbol value (S1 and S2 in this example) to the processing part 60 (Step 8).
  • the processing part 60 operates as a STTD decoder or a CLM1 phase compensator. Specifically, it is assumed that an output symbol (HS-DSCH/HS-SCCH) corresponding to the gth transmitting antenna in a single slot is expressed by ⁇ Sg(0),Sg(1),Sg(2),... ⁇ .
  • the output of the processing part 60 is S (0), S (1), S (2)...
  • a recovery symbol is outputted in accordance with the following formulas.
  • the computation for filter coefficients of a plurality of FIR filters used in a transmit diversity method can remarkably be simplified as compared to a case where an inverse matrix G -1 is used.
  • the present invention can be applied to receivers, equalizers, and the like which have a plurality of FIR filters and computes filter coefficients of the FIR filters with an inverse matrix. Furthermore, the present invention is applicable to systems using not only an STTD method and a CLM1 method but also various types of transmit diversity methods.

Landscapes

  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Power Engineering (AREA)
  • Radio Transmission System (AREA)
  • Cable Transmission Systems, Equalization Of Radio And Reduction Of Echo (AREA)
  • Mobile Radio Communication Systems (AREA)

Claims (9)

  1. Récepteur comprenant :
    une pluralité de filtres FIR (52, 54), et pouvant communiquer avec une partie émettrice en utilisant un procédé de diversité de transmission à travers un chemin de transmission au moyen de la pluralité de filtres FIR, caractérisé en ce qu'il comprend en outre :
    un circuit arithmétique de traitement (42 à 50) exploitable de manière à calculer un coefficient de filtre w de chacun des filtres FIR avec un facteur commun (co) et à générer en sortie les coefficients de filtre w vers la pluralité de filtres FIR ;
    dans lequel le coefficient de filtre est représenté par la formule suivante : w g = c 0 H ⁢ H ̲ g H
    Figure imgb0038
    où g est égal à 1 ou 2, l'exposant H représente une transposée d'Hamilton et Hg est une matrice de réponse de canal estimée du chemin de transmission ; et
    le facteur commun c0 est dérivé en représentant une matrice de gain G du chemin de transmission par la formule suivante, en utilisant des matrices de réponse de canal H 1 et H 2 correspondant au procédé de diversité de transmission : G = H ̲ 1 H ⁢ H ̲ 1 + H ̲ 2 H ⁢ H ̲ 2 + βI
    Figure imgb0039
    où β est un facteur de bruit ajouté dans le chemin de transmission, et I est une matrice unitaire.
  2. Récepteur selon la revendication 1, dans lequel le facteur commun c0 est calculé en mettant en oeuvre une substitution avant et une substitution arrière sur une matrice triangulaire inférieure L obtenue en mettant en oeuvre une décomposition de Cholesky sur la formule (2).
  3. Récepteur selon la revendication 2, dans lequel la substitution avant est mise en oeuvre en calculant un vecteur de rangée d selon la formule suivante : Ld = e N + 1 / 2 = e 1 , e 2 , … e N T
    Figure imgb0040
    où N est le nombre de composantes vectorielles d'une matrice, ei est un vecteur de colonne présentant 1 dans les cas de i = (N+1)/2 et 0 dans les autres cas.
  4. Récepteur selon la revendication 3, dans lequel d = d[(N-1)/2, (N-2)/2, ..., N-1] est utilisé en qualité de vecteur de rangée d pour la substitution arrière subséquente.
  5. Récepteur selon la revendication 4, dans lequel la substitution arrière comporte le calcul du facteur commun c0 selon la formule suivante (4) : L ̲ H ⁢ c ̲ 0 = d ̲
    Figure imgb0041
    où L H[i,j] = L H[i+(N-1)/2, j+(N-1)/2] ∀ 0 ≤ i, j ≤ (N-1)/2.
  6. Récepteur selon la revendication 5, dans lequel le facteur commun c0 est représenté par la formule suivante (5) : c 0 ⁢ N - 1 / 2 + k = c ̲ 0 k , c 0 k = c 0 N - 1 - k * , k = 0 , … , N - 1 / 2
    Figure imgb0042
  7. Récepteur selon la revendication 6, dans lequel le coefficient de filtre w est dérivé en mettant en oeuvre une transposée d'Hamilton sur le facteur commun c0 obtenu par la formule (5) pour obtenir c0 H, et en mettant ensuite en oeuvre le calcul de la formule (1).
  8. Récepteur selon la revendication 7, dans lequel le récepteur est utilisé en vue de communiquer avec une partie émettrice (11) qui transmet des symboles Sg(0), Sg(1), Sg(2), ... (g = 1 ou 2) au canal HS-DSCH/HS-SCCH par l'intermédiaire de deux antennes émettrices (Tx1, Tx2) en utilisant sélectivement un procédé STTD et un procédé CLM1 en tant que le procédé de diversité de transmission, et le récepteur présente une partie de traitement (60) exploitable de manière à générer en sortie des estimations de symboles S(0), S(1), S(2), ..., selon la formule (6) dans le cas du procédé STTD, et selon la formule (7) dans le cas du procédé CLM1 ; S ̲ 2 ⁢ i = S ⁢ 1 2 ⁢ i + S ⁢ 2 ⁢ 2 ⁢ i + 1 * , S ̲ ⁢ 2 ⁢ i + 1 = S ⁢ 1 ⁢ 2 ⁢ i + 1 - S ⁢ 2 ⁢ 2 ⁢ i *
    Figure imgb0043
     S ̲ i = S ⁢ 1 i + S ⁢ 2 i x W ⁢ 2
    Figure imgb0044
    où i = 0, 1, 2, ... et W2 est une pondération correspondant à une seconde antenne d'émission.
  9. Procédé de réception d'un signal transmis en utilisant un procédé de diversité de transmission, comprenant les étapes ci-dessous consistant à :
    fournir des données d'entrée r à une pluralité de filtres FIR ;
    fournir à chaque filtre FIR un vecteur de pondération wi sous la forme d'un coefficient de filtre ;
    calculer un coefficient de filtre w de chacun des filtres FIR, en supposant qu'une matrice de gain G d'un chemin de transmission est G = H ̲ 1 H ⁢ H ̲ 1 + H ̲ 2 H ⁢ H ̲ 2 + βI
    Figure imgb0045
    où l'exposant H représente une matrice de transposée d'Hamilton, H 1 et H 2 sont des matrices de réponses de canal estimées du chemin de transmission, β est un facteur de bruit dans le chemin de transmission, et I est une matrice unitaire ; et
    calculer un facteur commun (co) au moyen de la matrice de gain, et obtenir les coefficients de filtre w à partir du facteur commun (co).
EP07741040.5A 2006-03-31 2007-03-29 Recepteur Not-in-force EP2007028B1 (fr)

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
JP2006097704 2006-03-31
PCT/JP2007/057604 WO2007114478A1 (fr) 2006-03-31 2007-03-29 récepteur

Publications (4)

Publication Number Publication Date
EP2007028A2 EP2007028A2 (fr) 2008-12-24
EP2007028A9 EP2007028A9 (fr) 2009-07-29
EP2007028A4 EP2007028A4 (fr) 2013-07-31
EP2007028B1 true EP2007028B1 (fr) 2015-07-01

Family

ID=38563738

Family Applications (1)

Application Number Title Priority Date Filing Date
EP07741040.5A Not-in-force EP2007028B1 (fr) 2006-03-31 2007-03-29 Recepteur

Country Status (5)

Country Link
US (1) US8189654B2 (fr)
EP (1) EP2007028B1 (fr)
JP (1) JP5077578B2 (fr)
CN (1) CN101411088B (fr)
WO (1) WO2007114478A1 (fr)

Families Citing this family (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2008081715A1 (fr) * 2006-12-28 2008-07-10 Nec Corporation Égalisation de données dans un récepteur de communication avec diversité en émission et en réception
EP2100388A4 (fr) * 2006-12-28 2014-04-09 Nec Corp Égalisation de données dans un récepteur de communication avec diversité en réception
US9166663B2 (en) * 2012-12-14 2015-10-20 Futurewei Technologies, Inc. System and method for open-loop MIMO communications in a SCMA communications system
CN105703815B (zh) * 2014-11-28 2018-09-14 辰芯科技有限公司 基于闭环发送分集模式下的均衡方法、均衡设备及相应系统

Family Cites Families (15)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2000315966A (ja) * 1999-02-25 2000-11-14 Texas Instr Inc <Ti> Ｔｄｄ／ｗｃｄｍａのための空間時間送信ダイバーシチ
US6707864B2 (en) 2001-01-25 2004-03-16 Interdigital Technology Corporation Simplified block linear equalizer with block space time transmit diversity
US6466611B1 (en) * 2001-02-07 2002-10-15 Motorola, Inc. Multi-user detection using a finite-impulse-response matrix filter
EP1552405B1 (fr) * 2002-08-20 2009-06-03 Interdigital Technology Corporation Detection conjointe puissante
CN1723629A (zh) 2003-01-10 2006-01-18 美商内数位科技公司 通用二阶段数据估测
JP2005064581A (ja) * 2003-08-12 2005-03-10 Fujitsu Ten Ltd ダイバーシティ受信回路
JP2005079836A (ja) 2003-08-29 2005-03-24 Sharp Corp Ｗ−ｃｄｍａ方式の無線通信に用いる受信装置及び受信方法
US7324583B2 (en) * 2004-02-13 2008-01-29 Nokia Corporation Chip-level or symbol-level equalizer structure for multiple transmit and receiver antenna configurations
US7450924B1 (en) * 2004-03-25 2008-11-11 At&T Mobility Ii Llc Interference cancellation and receive diversity for single-valued modulation receivers
AU2005203016A1 (en) 2004-07-20 2006-02-09 Nec Australia Pty Ltd Method of noise factor computation for a chip equaliser in a spread spectrum receiver
AU2005203278A1 (en) 2004-08-12 2006-03-02 Nec Australia Pty Ltd Method for calculating filter coefficients for an equaliser in a communication receiver
US7483480B2 (en) * 2004-11-24 2009-01-27 Nokia Corporation FFT accelerated iterative MIMO equalizer receiver architecture
US20070076791A1 (en) * 2005-07-26 2007-04-05 Interdigital Technology Corporation Approximate cholesky decomposition-based block linear equalizer
US7684479B2 (en) * 2005-08-12 2010-03-23 Broadcom Corporation Methods and systems for soft-bit demapping
US7929597B2 (en) * 2005-11-15 2011-04-19 Qualcomm Incorporated Equalizer for a receiver in a wireless communication system

Also Published As

Publication number Publication date
EP2007028A9 (fr) 2009-07-29
JP5077578B2 (ja) 2012-11-21
JPWO2007114478A1 (ja) 2009-08-20
EP2007028A4 (fr) 2013-07-31
WO2007114478A1 (fr) 2007-10-11
CN101411088B (zh) 2012-12-05
US20090252214A1 (en) 2009-10-08
CN101411088A (zh) 2009-04-15
EP2007028A2 (fr) 2008-12-24
US8189654B2 (en) 2012-05-29

Similar Documents

Publication Publication Date Title
US8018903B2 (en) Closed-loop transmit diversity scheme in frequency selective multipath channels
US7339980B2 (en) Successive interference cancellation in a generalized RAKE receiver architecture
KR100647505B1 (ko) 다중 경로 페이딩 채널에 효율적인 다중 입출력 시스템
US7623572B2 (en) Noise variance estimation for frequency domain equalizer coefficient determination
US7570689B2 (en) Advanced receiver with sliding window block linear equalizer
EP1727297A1 (fr) Procédé et terminal pour réduire l&#39;interférence dans un système de radio communication
EP1686696B1 (fr) Détection mono-utilisateur
EP1579599B1 (fr) Solution a antennes multiples pour telephone mobile
US20080019431A1 (en) Groupwise successive interference cancellation for block transmission with reception diversity
EP1987599B1 (fr) Suppression d&#39;interférence à complexité réduite pour des communications sans fil
US20040047403A1 (en) Adaptive interference suppression receiving apparatus for space-time block coded direct sequence/code division multiple access communication system
US20070076791A1 (en) Approximate cholesky decomposition-based block linear equalizer
US20030142762A1 (en) Wireless receiver method and apparatus using space-cover-time equalization
EP2100389B1 (fr) Égalisation de données dans un récepteur de communication avec diversité en émission et en réception
US20030072282A1 (en) Code division multiple access downlink receiver
EP2007028B1 (fr) Recepteur
US20030095585A1 (en) Method and apparatus for downlink joint detection in a communication system
US20090323874A1 (en) Channel estimation using common and dedicated pilots
KR20040094443A (ko) 수신기 기능부를 사용한 전송 처리
EP2153540B1 (fr) Procédé et appareil de démodulation à réseau réduit
US20030026345A1 (en) Multipath equalization for MIMO multiuser systems
CN101128993B (zh) 信道估计装置、码分多址接收装置以及信道估计方法
AU2007249091A1 (en) Data equalisation in a communication receiver with receive diversity
EP1351426B1 (fr) Récepteur et procédé pour un système de communication sans fil codé en temps et en espace avec résolution des trajets multiples
Pacheco et al. Semi-blind interference suppression for frequency selective DS-CDMA systems employing space-time block codes

Legal Events

Date Code Title Description
PUAI Public reference made under article 153(3) epc to a published international application that has entered the european phase

Free format text: ORIGINAL CODE: 0009012

PUAB Information related to the publication of an a document modified or deleted

Free format text: ORIGINAL CODE: 0009199EPPU

17P Request for examination filed

Effective date: 20081020

AK Designated contracting states

Kind code of ref document: A2

Designated state(s): DE FR GB IT

RBV Designated contracting states (corrected)

Designated state(s): DE FR GB IT

DAX Request for extension of the european patent (deleted)
A4 Supplementary search report drawn up and despatched

Effective date: 20130628

RIC1 Information provided on ipc code assigned before grant

Ipc: H04B 1/707 20110101AFI20130624BHEP

Ipc: H04B 7/02 20060101ALI20130624BHEP

Ipc: H04B 7/005 20060101ALI20130624BHEP

Ipc: H04L 1/06 20060101ALI20130624BHEP

Ipc: H04L 25/03 20060101ALI20130624BHEP

Ipc: H04B 7/08 20060101ALI20130624BHEP

Ipc: H04B 3/06 20060101ALI20130624BHEP

REG Reference to a national code

Ref country code: DE

Ref legal event code: R079

Ref document number: 602007041940

Country of ref document: DE

Free format text: PREVIOUS MAIN CLASS: H04B0007080000

Ipc: H04B0001707000

GRAP Despatch of communication of intention to grant a patent

Free format text: ORIGINAL CODE: EPIDOSNIGR1

RIC1 Information provided on ipc code assigned before grant

Ipc: H04B 1/707 20110101AFI20141211BHEP

Ipc: H04L 25/03 20060101ALI20141211BHEP

Ipc: H04B 3/06 20060101ALI20141211BHEP

Ipc: H04B 7/08 20060101ALI20141211BHEP

Ipc: H04B 7/02 20060101ALI20141211BHEP

Ipc: H04B 7/005 20060101ALI20141211BHEP

Ipc: H04L 1/06 20060101ALI20141211BHEP

INTG Intention to grant announced

Effective date: 20150113

GRAS Grant fee paid

Free format text: ORIGINAL CODE: EPIDOSNIGR3

GRAA (expected) grant

Free format text: ORIGINAL CODE: 0009210

AK Designated contracting states

Kind code of ref document: B1

Designated state(s): DE FR GB IT

REG Reference to a national code

Ref country code: GB

Ref legal event code: FG4D

REG Reference to a national code

Ref country code: DE

Ref legal event code: R096

Ref document number: 602007041940

Country of ref document: DE

REG Reference to a national code

Ref country code: FR

Ref legal event code: PLFP

Year of fee payment: 10

REG Reference to a national code

Ref country code: DE

Ref legal event code: R097

Ref document number: 602007041940

Country of ref document: DE

PLBE No opposition filed within time limit

Free format text: ORIGINAL CODE: 0009261

STAA Information on the status of an ep patent application or granted ep patent

Free format text: STATUS: NO OPPOSITION FILED WITHIN TIME LIMIT

PGFP Annual fee paid to national office [announced via postgrant information from national office to epo]

Ref country code: FR

Payment date: 20160208

Year of fee payment: 10

RAP2 Party data changed (patent owner data changed or rights of a patent transferred)

Owner name: LENOVO INNOVATIONS LIMITED (HONG KONG)

26N No opposition filed

Effective date: 20160404

PGFP Annual fee paid to national office [announced via postgrant information from national office to epo]

Ref country code: IT

Payment date: 20160324

Year of fee payment: 10

PGFP Annual fee paid to national office [announced via postgrant information from national office to epo]

Ref country code: DE

Payment date: 20170321

Year of fee payment: 11

PGFP Annual fee paid to national office [announced via postgrant information from national office to epo]

Ref country code: GB

Payment date: 20170329

Year of fee payment: 11

REG Reference to a national code

Ref country code: FR

Ref legal event code: ST

Effective date: 20171130

PG25 Lapsed in a contracting state [announced via postgrant information from national office to epo]

Ref country code: FR

Free format text: LAPSE BECAUSE OF NON-PAYMENT OF DUE FEES

Effective date: 20170331

PG25 Lapsed in a contracting state [announced via postgrant information from national office to epo]

Ref country code: IT

Free format text: LAPSE BECAUSE OF NON-PAYMENT OF DUE FEES

Effective date: 20170329

REG Reference to a national code

Ref country code: DE

Ref legal event code: R119

Ref document number: 602007041940

Country of ref document: DE

GBPC Gb: european patent ceased through non-payment of renewal fee

Effective date: 20180329

PG25 Lapsed in a contracting state [announced via postgrant information from national office to epo]

Ref country code: DE

Free format text: LAPSE BECAUSE OF NON-PAYMENT OF DUE FEES

Effective date: 20181002

PG25 Lapsed in a contracting state [announced via postgrant information from national office to epo]

Ref country code: GB

Free format text: LAPSE BECAUSE OF NON-PAYMENT OF DUE FEES

Effective date: 20180329