Classifications

• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04B—TRANSMISSION
  • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
  • H04B1/06—Receivers
  • H04B1/16—Circuits
  • H04B1/26—Circuits for superheterodyne receivers
  • H04B1/28—Circuits for superheterodyne receivers the receiver comprising at least one semiconductor device having three or more electrodes
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
  • H04L27/22—Demodulator circuits; Receiver circuits
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
  • H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
  • H04L27/38—Demodulator circuits; Receiver circuits
  • H04L27/3845—Demodulator circuits; Receiver circuits using non - coherent demodulation, i.e. not using a phase synchronous carrier
  • H04L27/3854—Demodulator circuits; Receiver circuits using non - coherent demodulation, i.e. not using a phase synchronous carrier using a non - coherent carrier, including systems with baseband correction for phase or frequency offset
  • H04L27/3863—Compensation for quadrature error in the received signal
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/0014—Carrier regulation
  • H04L2027/0016—Stabilisation of local oscillators
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/0014—Carrier regulation
  • H04L2027/0083—Signalling arrangements
  • H04L2027/0089—In-band signals
  • H04L2027/0093—Intermittant signals
  • H04L2027/0095—Intermittant signals in a preamble or similar structure

Definitions

• This invention relates to In phase (I) and Quadrature (Q) path imbalance compensation. More particularly, but not exclusively, the invention relates to a receiver, such as a direct down conversion radio receiver, incorporating I/Q path imbalance compensation.
• a receiver such as a direct down conversion radio receiver
• a direct down conversion receiver 1 has an antenna 2 for receiving a signal.
• the antenna 2 is connected to output the received signal to a Radio Frequency (RF) filter 3, which is a band-pass filter for filtering the received signal.
• RF filter 3 is connected to output the filtered signal to a Low Noise Amplifier (LNA) 4 for amplifying the filtered signal.
• LNA Low Noise Amplifier
• the LNA 4 is connected to output the amplified signal to both an In-phase (I) path 5 and a Quadrature (Q) path 6 of the receiver 1.
• the I path 5 comprises an I mixer 7, an analogue low-pass filter 8, and Automatic Gain Controller (AGC) 9, an Analogue to Digital Converter (ADC) 10 and a digital low pass filter 11 connected in series in that order.
• the Q path 6 is identical to the I path, except that it has a Q mixer 12 in place of the I mixer 7. So, the Q path 6 comprises a Q mixer 12, an analogue low-pass filter 13, an AGC 14, an ADC 15 and a digital low pass filter 16.
• Signals output by the I and Q paths 5, 6 pass to a demodulation and decoding stage 17 for demodulating and decoding the signals to produce a data output 18.
• the I mixer 7 and Q mixer 12 each mix the amplified signal output by the LNA 4 with a local oscillator signal.
• the local oscillator signals should have substantially the same frequency as the carrier frequency of the wanted signal modulated in the signal received at the antenna 2. This allows the mixers 7, 12 to down convert the wanted signal to baseband.
• the local oscillator signals used by the I and Q mixers are shifted from one another by 90° in phase so that the wanted signal can be properly demodulated from the signals on the I path 5 and Q path 6.
• phase imbalance This variation is known as phase imbalance.
• the gains over the I path 5 and Q path 6 may not be the same.
• the signals output by the I and Q paths 5, 6 may therefore have unintended differences in amplitude. This is known as amplitude imbalance.
• Phase and amplitude imbalance can cause errors in demodulation. It can also cause the aliasing of image interference into the band of the wanted signal.
• phase and amplitude imbalance Many methods have been suggested for correcting phase and amplitude imbalance, most of which involve calibrating a receiver using a calibration signal.
• the calibration signal has a known frequency, phase and amplitude.
• the signals output by the I and Q paths can be analysed to determine the phase and amplitude imbalances between them.
• the calibration signal is most often generated locally, by the receiver to be calibrated, so that its frequency, phase and amplitude can be carefully controlled.
• this tends to require the provision of a dedicated circuit for generating the calibration signal, which can by undesirably expensive and complex.
• Some methods avoid this by receiving an external calibration signal via the antenna via which the receiver usually receives signals.
• a receiver for receiving a signal including a training sequence and a payload, the receiver comprising an In-phase path and a Quadrature path for converting the received signal to In-phase and Quadrature signals for output to a demodulator; processing means for determining an imbalance compensation parameter from the In-phase and Quadrature signals representing the training sequence; and an imbalance compensation circuit for compensating for phase and amplitude imbalance between the In-phase and Quadrature signals representing the payload using the determined imbalance compensation parameter.
• a method of receiving a signal including a training sequence and a payload comprising converting the received signal to In-phase and Quadrature signals for output to a demodulator; determining an imbalance compensation parameter from the In-phase and Quadrature signals representing the training sequence; and compensating for phase and amplitude imbalances between the In- phase and Quadrature signals representing the payload using the determined imbalance compensation parameter.
• phase and amplitude imbalance compensation during reception of a signal's payload can be based on the signal's training sequence.
• Training sequences are an inherent part of the signals of many current communication systems. So, the invention has the significant advantage of allowing compensation for phase and amplitude imbalance without the use of a dedicated calibration signal. No calibration signal needs to be generated locally at the receiver. Similarly, communication capacity does not need to be set aside for the reception of a dedicated calibration signal.
• the training sequence may be also used for a purpose in addition to determining the imbalance compensation parameter.
• This might include additional signal processing.
• the processing means may use the training sequence for additional signal processing.
• the method may include using the training sequence for additional signal processing.
• the additional signal processing may comprise training an equaliser.
• the additional signal processing may comprise any one of synchronisation, channel quality estimation, signal level adjustment and so on.
• the training sequence may take a variety of forms.
• the training sequence is usually a portion of a received signal that is known, e.g. the signal expected to be received during the training sequence may be known.
• the training sequence might comprise a preamble, pilot sequence, access code, synchronisation word, synchronisation sequence or such like.
• the training sequence is usually adapted for a particular purpose in addition to amplitude and phase imbalance compensation. For example, it may meet the requirements of a particular conventional communication standard.
• the training sequence may be suitable for training an equaliser or for enabling synchronisation.
• this means that the training sequence may comprise a sequence having a prominent peak in its autocorrelation function..
• the training sequence and the payload may be contained in a data packet.
• the compensation of the signals representing the payload of the data packet may therefore be performed using the imbalance compensation parameter determined from the signals representing the training sequence of that data packet.
• imbalance compensation can be achieved from the signals representing a single data packet.
• imbalance compensation could be carried out on a packet by packet basis, if desired.
• the compensation parameters are determined in the time domain rather than the frequency domain, as calculations in the time domain are quicker and easier to implement, avoiding the use of Fourier transforms.
• receiver comprises a selector for selecting portions of the In-phase and Quadrature signals of the training sequence at two different times.
• the imbalance compensation parameter can be determined from the selected portions of the In-phase and Quadrature signals at the two different times. These times are illustrated as times t ⁇ and t 2 in the description of the preferred embodiments of the invention below.
• the portions of the In-phase and Quadrature signals are selected from a part of the training sequence that is also used for the further processing described above. In other words, there need not be a specific part of the training sequence used only for amplitude and phase imbalance compensation.
• the determination of the imbalance compensation parameter is also useful for the determination of the imbalance compensation parameter to be based on signals representing the training sequence as expected to be received. These are illustrated by the baseband signals baseband signals A(t ⁇ ) , A(t 2 ) , BQ 1 ) , B(t 2 ) in the description of the preferred embodiments of the invention below.
• An imbalance compensation parameter may be determined based on the In-phase signal and another imbalance compensation parameter is determined based on the Quadrature signal. Indeed, it is preferred that a pair of imbalance compensation parameters are determined from the In-phase signal and another pair of imbalance compensation parameters are determined from the Quadrature signal. These are illustrated by imbalance compensation parameters g 7 COs(Cp 1 ) , g 7 Sm(Cp 1 ) and imbalance compensation parameters g e cos( ⁇ 3 ) , g e sin( ⁇ 3 ) respectively in the description of the preferred embodiments of the invention below. It is also preferred that a further imbalance compensation parameter is determined from mean values of these imbalance compensation parameters. This is illustrated by further imbalance compensation parameter g j g Q SUi(Cp 3 -(P 1 ) in the description of the preferred embodiments of the invention below.
• the phase and amplitude imbalance compensation can use these parameters. More specifically, the compensation can involve multiplying the In-phase and Quadrature signals by the determined imbalance compensation parameter(s). It is preferred that the compensation involves multiplying the In- phase signal by the imbalance compensation parameter(s) determined from the Quadrature signal and Quadrature signals by imbalance compensation parameter(s) determined from the In-phase signal. It is also preferred that the compensation involves multiplying the In-phase and Quadrature signals by the inverse of the determined further imbalance compensation parameter. This is illustrated by equations (6) and (7) in the description of the preferred embodiments of the invention below.
• processing means means
• circuit means
• switch means
• circuit means
• switch means
• circuit means
• switch means
• circuit means
• switch means
• computer program code adapted to carry out the method described above when processed by a processing means.
• the computer software or computer program code can be carried by a computer readable medium.
• the medium may be a physical storage medium such as a Read Only Memory (ROM) chip. Alternatively, it may be a disk such as a Digital Versatile Disk (DVD-ROM) or Compact Disk (CD-ROM). It could also be a signal such as an electronic signal over wires, an optical signal or a radio signal such as to a satellite or the like.
• ROM Read Only Memory
• DVD-ROM Digital Versatile Disk
• CD-ROM Compact Disk
• the invention also extends to a processor running the software or code, e.g. a computer configured to carry out the method described above.
• Figure 1 is a schematic illustration of a direct down conversion receiver according to the prior art
• Figure 2 is a schematic illustration of a direct down conversion receiver according to a first preferred embodiment of the present invention
• Figure 3 is a schematic illustration of an imbalance compensation parameter determining stage of the direct down conversion receiver illustrated in Figure 2;
• Figure 4 is a schematic illustration of a phase and amplitude compensation stage of the direct down conversion receiver illustrated in Figure 2;
• FIG. 5 is a schematic illustration of a heterodyne receiver according to a second preferred embodiment of the present invention.
• a direct down conversion receiver 19 according to a first preferred embodiment of the invention has many components similar to those of the receiver 1 of the prior art described above with reference to Figure 1 and the same reference numerals are used for the similar components.
• the receiver 19 according to the first preferred embodiment of the invention has an imbalance compensation parameter determining stage 20 and a phase and amplitude imbalance compensation stage 21.
• the imbalance compensation parameter determining stage 20 is connected to receive I and Q signals output from the I and Q paths 5, 6 of the receiver 19 after filtering by the analogue low pass filters 8, 13, conversion from analogue to digital by the ADCs 10, 15 and filtering by the digital low pass filters 11 , 16.
• the determining stage 20 is shown in more detail in Figure 3, from which it can be seen that it includes a selector 22 for selecting portions of the received I and Q signals I'(t) and Q ⁇ t) output by the
• I and Q paths 5, 6 and a processor 23 for processing the selected portions of the signals to determine and calculate imbalance compensation parameters g / cos( ⁇ 1 ) , g 7 sin( ⁇ j ) , g e cos( ⁇ 3 ) , g e sin( ⁇ 3 ) and
• A(t) and B(t) are baseband signals modulated onto a carrier signal cos(w c t+ ⁇ 1 ) and s ⁇ n(w J + ⁇ 1 )
• the I and Q mixers 7, 12 of the receiver 19 mix
• I(t) S(t) * ⁇ gl ⁇ s(w c t) ⁇
• the phase variation ⁇ is not zero, i.e. ⁇ ⁇ 0
• the gain factors g 7 and g Q are not equal, i.e. g 7 ⁇ g Q .
• the components of the signals I(t) , Q(t) on I path 5 and Q path 6 after mixing that have a frequency around 2w c are relatively easily removed by the low pass filters 8, 11 , 13, 16.
• the signals , QXt) output from the I and Q paths 5, 6 can therefore be expressed more simply as
• Equations (4) and (5) can be rearranged to give
• the selector 22 selects portions
• the training sequence is received at the beginning of a data packet that also contains a payload and is used to train an equalizer (not shown) of the receiver 19.
• the imbalance compensation parameters g / cos( ⁇ 1 ), g j sin((p j ) are determined from the signal portions /'(V 1 ), 1 Xh) selected by the selector 22 from the signal I'Q) output from the I path 5 and the known baseband signals AQ 1 ), AQ 2 ), BQ 1 ), B(t 2 ) of the training sequence at the two different times t ⁇ and t 2 using equations (8) and (9) below (on condition that A(I 1 )BQ 2 ) ⁇ A(I 2 )B(I 1 )).
• the compensation parameters g g cos( ⁇ 3 ) g e sin( ⁇ 3 ) are determined from signal portions Q 1 Q 1 ) , Q 1 Q 2 ) selected by the selector 22 from the signal Q'Q) output from the Q path 6 and the known baseband signals AQ 1 ), AQ 2 ), BQ 1 ), BQ 2 ) of the training sequence at the two different times t ⁇ and t 2 using equations (10) and (11) below (on the same condition that AQ 1 )BQ 2 ) ⁇ AQ 2 )BQ 1 )).
• the imbalance compensation parameters g / cos( ⁇ 1 ), g j sint ⁇ j ) g Q cos( ⁇ 3 ) and g Q sin( ⁇ 3 ) determined using equations (8), (9), (10) and (11) can be averaged over a given time, in this embodiment during the whole training sequence, to give mean values of each parameter g 7 COs(Cp 1 ) , g j Sm(Cp 1 ) g ⁇ cos( ⁇ 3 ) and g Q sin( ⁇ 3 ) . These can then be used to calculate a further imbalance compensation parameter
• this allows the imbalance compensation parameters g / cos( ⁇ 1 ) , g 7 Sm(Cp 1 ) , g e cos(cp 3 ) , g ⁇ sin( ⁇ 3 ) and g 7 g e sin(cp 3 -Cp 1 ) determined and calculated from the signals I ⁇ i) , Q ⁇ t) output by the I and Q paths 5, 6 at times when the baseband signals A(t) , B(t) used to modulate the carrier signal cos(w c t+ ⁇ 1 ) and are known (e.g.
• the phase and amplitude imbalance compensation stage 21 uses the imbalance compensation parameters g 7 cos( ⁇ 1 ) , g / sin( ⁇ 1 ) , g e cos( ⁇ 3 ) , g ⁇ sin( ⁇ 3 ) and g 7 g e SUi(Cp 3 -Cp 1 ) to compensate for phase and amplitude imbalance in the signals I ⁇ i) , Q ⁇ t) using equations (6) and (7).
• the phase and amplitude imbalance compensation stage 21 comprises a pair of I signal multipliers 24, 25, to each of which the I signal I'(t) is input, and a pair of Q signal multipliers 26, 27, to each of which the Q signal Q ⁇ t) is input.
• One multiplier 24 of the pair of I signal multipliers 24, 25 multiplies the I signal I'(t) with a first of the imbalance compensation parameters, g Q cos( ⁇ 3 ) , determined from the Q signal Q ⁇ t) .
• the other multiplier 25 of the pair of I signal multipliers 24, 25 multiplies the I signal with a second of the imbalance compensation parameters, g ⁇ sin( ⁇ 3 ) , determined from the Q signal Q ⁇ t) .
• one multiplier 26 of the pair of Q signal multipliers 26, 27 multiplies the Q signal Q ⁇ t) with a first of the imbalance compensation parameters, g 7 COs(Cp 1 ) , determined from the I signal .
• the other multiplier 27 of the pair of Q signal multipliers 26, 27 multiplies the Q signal Q ⁇ t) with a second of the imbalance compensation parameters, g 7 Sm(Cp 1 ) , determined from the I signal I'(t) .
• the phase and amplitude imbalance compensation stage 21 also has a pair of adders 28, 29 and a further pair of multipliers 30, 31.
• a first adder 28 of the pair of adders 28, 29 receives the signal output by the first I signal multiplier 24 and the first Q signal multiplier 26.
• a second of the pair of adders 28, 29 receives the signal output by the second I signal multiplier 25 and the second Q signal multiplier 27.
• the adders 28, 29 add the signals they each receive together and output the resulting added signals to respective multipliers 30, 31 of the pair of multipliers 30, 31.
• a heterodyne receiver 32 according to a second preferred embodiment of the invention has many components similar to those of the direct down conversion receiver 19 of the first preferred embodiment of the invention described above with reference to Figures 2 to 4 and the same reference numerals are used for the similar components.
• the receiver 32 has a conversion stage 33 comprising a mixer 34 for mixing the amplified filtered received signal with a local oscillator signal at a frequency fo, an intermediate frequency filter 35 for filtering the signal after mixing and an AGC 36 for adjusting the gain of the signal for output to the I and Q mixers 7, 12.
• the local oscillator signal used by the I and Q mixers 7, 12 is at an intermediate frequency (IF).
• IF intermediate frequency
• the combination of the mixing with the local oscillator frequency at the frequency fo and the local oscillator signal at the intermediate frequency (IF) converts the received signal to baseband.
• the wanted signal is therefore at baseband in the I and Q paths 5, 6.

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• Engineering & Computer Science (AREA)
• Computer Networks & Wireless Communication (AREA)
• Signal Processing (AREA)
• Digital Transmission Methods That Use Modulated Carrier Waves (AREA)
• Superheterodyne Receivers (AREA)

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• 2007
  • 2007-02-08 WO PCT/IB2007/050412 patent/WO2007096800A2/en active Application Filing
  • 2007-02-08 CN CNA200780006274XA patent/CN101390360A/zh active Pending
  • 2007-02-08 JP JP2008555910A patent/JP2009527968A/ja active Pending
  • 2007-02-08 EP EP07705818A patent/EP1989851A2/de not_active Withdrawn
  • 2007-02-08 KR KR1020087022851A patent/KR100977938B1/ko not_active IP Right Cessation
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