EP1935213B1 - Method for operating an induction heating device - Google Patents

Abstract

The invention relates to a method for operating an induction heating device. The induction heating device comprises an induction coil and a frequency converter for producing a control voltage for the induction coil. The frequency converter comprises a rectifier rectifying an alternating supply voltage (UN), an intermediate circuit capacitor, looped in between output terminals of the rectifier and equalizing the rectified voltage (UG), and at least one controllable switching element, looped in between the output terminals of the rectifier. According to the invention, in a predetermined discharge interval (INT) before a zero crossing (ND) of the alternating supply voltage (UN), the intermediate circuit capacitor is discharged to a threshold value by controlling the at least one switching element before the induction coil is controlled in order to produce an adjustable heating capacity.

Description

The invention relates to a method for operating an induction heating device according to the preamble of claim 1.

In induction heating devices, an induction coil is subjected to an alternating voltage or an alternating current, whereby eddy currents are induced in a cookware to be heated, which is magnetically coupled to the induction coil. The eddy currents cause heating of the cookware.

For controlling the induction coil, different circuit arrangements and control methods are known. All circuit or process variants have in common that they produce a high-frequency drive voltage for the induction coil from a low-frequency power input voltage. Such circuits are referred to as converters.

For conversion or frequency conversion, the mains input voltage is usually first rectified by means of a rectifier into a DC supply voltage or intermediate circuit voltage and then processed to generate the high-frequency drive voltage by means of one or more switching means, generally insulated gate bipolar transistors (IGBT) , At the output of the rectifier, ie between the intermediate circuit voltage and a reference potential, a so-called DC link capacitor for buffering the DC link voltage is usually provided.

A first converter variant forms a converter in full bridge circuit, in which the induction coil and a capacitor are connected in series between two so-called half bridges. The half bridges are each looped between the intermediate circuit voltage and the reference potential. The induction coil and the capacitor form a series resonant circuit.

Another converter variant forms a half-bridge circuit of two IGBTs, wherein the induction coil and two capacitors, which are connected in series between the intermediate circuit voltage and the reference potential, form a series resonant circuit. The induction coil is connected to a connection to a connection point of the two capacitors and to its other connection to a connection point of the two IGBTs forming the half-bridge.

However, both the variant with full bridge and the variant with half bridge are relatively expensive due to the large number of components required, in particular of IGBTs.

Therefore, a variant optimized from a cost point of view uses only one switching means or an IGBT, wherein the induction coil and a capacitor form a parallel resonant circuit. Between the output terminals of the rectifier, parallel to the DC link capacitor, the parallel resonant circuit of induction coil and capacitor are connected in series with the IGBT.

Allen mentioned inverter variants have in common that the link capacitor charges during a first half-wave to an open circuit voltage with an amount of peak value of the AC line voltage, for example, to 325V at a mains AC voltage of 230V, as soon as they are supplied with mains voltage.

If no drive voltage for generating power of the induction coil is generated, that is to block the switching means or IGBTs, the voltage present at the intermediate circuit capacitor remains approximately constant. When starting the inverter, that is, when the induction coil is driven to generate an adjustable heating power or applied with an AC voltage flows when turning on the IGBTs or first, a high current from the DC link capacitor in the resonant circuit and by the IGBTs or. This causes audible noise in a cookware heated by the induction heater, for example, in a pot bottom. Furthermore, reduces the life of the acted upon by the high inrush current components.
US4438311 discloses a method according to the prior art.

Task and solution

The invention is therefore based on the object to provide a method for operating an induction heater with a converter available that allows reliable, component-saving and quiet operation of the induction heater with low noise radiation.

The invention solves this problem by a method having the features of claim 1. Advantageous and preferred embodiments of the invention are the subject of the other claims and are explained in more detail below. The wording of the claims is incorporated herein by express reference.

According to the invention, the intermediate circuit capacitor is discharged to a threshold value by driving the switching means in a time range before a zero crossing of the AC mains voltage before the induction coil is driven to produce an adjustable heating power, which already during discharge a small heating power supply in an optionally existing cookware. The discharge of the intermediate circuit capacitor causes that at a start of a heating process, ie when the induction coil is to deliver heating power to a cookware, the intermediate circuit capacitor is substantially discharged. If at this time the switching means is turned on or conductive, there is no or only a small current pulse through the switching means and the resonant circuit of induction coil and capacitor. Consequently, there is no switch-on noise and the pulse current load of the power components is reduced, which increases their life. After discharging the DC link capacitor, the actual heating process can be carried out in a conventional manner, for example, the or the switching means can be controlled with a square wave signal with a working frequency and an associated Arbeitstastverhältnis. The inverter is consequently started up with small currents or voltages in the area of the zero crossing. With the rise of the half-wave after the zero crossing, the inverter can adjust to its, the set heating power corresponding operating point with a working frequency and a duty cycle.

In a further development, the converter is a single-transistor converter. The at least one switching means preferably forms the switching means of the single-transistor converter. Alternatively, the inverter is designed in full-bridge circuit or half-bridge circuit, wherein the at least one switching means is part of a bridge.

In a further development, the time range starts from 1 ms to 5 ms, preferably 2.5 ms, before the zero crossing of the mains alternating voltage. This allows a reliable discharge of the DC link capacitor, with comparatively low loss power generation in the switching means by the discharge.

In a further development, the threshold is in a range of 0V to 20V. Preferably, the intermediate circuit capacitor is discharged to 0V. This allows a practically impuls current-free starting of the inverter.

In a development, the at least one switching means is a transistor, in particular an IGBT. Preferably, the transistor for discharging the intermediate circuit capacitor is driven during the discharge such that a linear operating state of the transistor is established. Since the transistor does not completely switch through in this operating mode or this operating state, the DC link capacitor is discharged slowly along the mains half-cycle. The resulting currents through the parallel resonant circuit and the transistor remain relatively low, whereby noise is avoided or significantly reduced.

In a development, the switching means for discharging the DC link capacitor is driven with a pulse width modulated square wave signal. The square-wave voltage signal preferably has a frequency in the range from 20 kHz to 50 kHz, in particular 39 kHz, and / or an on / off ratio in the range from 1/300 to 1/500, in particular 1/378. In this way, a controlled discharge of the DC link capacitor can be effected without an excessive discharge current flows. The frequency and / or the on / off ratio is preferably adapted to a used IGBT type, its drive voltage, a driver circuit used for generating the drive voltage and / or to a capacitance value of the DC link capacitor.

In a development, the adjustable heating power is generated by means of a half-wave pattern, wherein the intermediate circuit capacitor is discharged before activation of a half-wave. In a heating power generation With the aid of the half-wave pattern, individual half-waves of the mains alternating voltage are completely blanked out or deactivated, ie not used for heating power generation. In a so-called 1/3-Netzhalbwellenbetrieb example, only one of three consecutive half-waves for power supply to the resonant circuit or the induction coil is used or activated. During the remaining two half-cycles, the switching means remains open, ie no power is fed into the resonant circuit. In a 2/3 mains half-wave operation, two out of three consecutive half-waves are used or activated for supplying power to the oscillating circuit or the induction coil. During an active half cycle, power adjustment is done in a conventional manner. Line half-wave operation allows finer resolution of power levels over a wide power setting range. Such a power setting is particularly advantageous for single-transistor converters. When a half-wave operation is used for power adjustment in a conventional operating method of the single-transistor converter, an open-circuit voltage, for example, 325V at 230V mains voltage, turns on the intermediate circuit capacitor during an inactive half-wave, during which no power is fed into the resonant circuit.

Therefore, when the switching means is turned on for the first time during the transition from a non-active to an active half-wave, a high current flows through the oscillating circuit and the switching means for a short time, as a result of which a noise is caused. With 1/3 and 213 half-wave operation, this produces a noise every 30ms. This is unreasonable for a user. Therefore, conventional single-transistor inverters typically do not use half-wave control for power adjustment. When using the inventive discharge of the DC link capacitor before activating a half-wave, ie the transition from a deactivated to an activated half-wave, resulting in a transition no high inrush current, ie it can also be used in Eintransistorumrichter a half-wave controller for power adjustment. Preferably one of three or two out of three half-waves is activated, ie the 1/3 or 2/3 half-wave operation is set.

These and other features will become apparent from the claims and from the description and drawings, wherein the individual features in each case alone or more in the form of sub-combinations in an embodiment of the invention and in other fields be realized and advantageous and protectable Represent embodiments for which protection is claimed here. The subdivision of the application into individual sections and intermediate headings does not limit the statements made thereunder in their generality.

Brief description of the drawings

Fig. 1

ein Schaltbild eines Eintransistorumrichters, der mit dem erfindungsgemäßen Betriebsverfahren betrieben wird,

Fig. 2

Zeitablaufdiagramme von Signalen des Eintransistorumrichters von Fig. 1,

Fig. 3

ein Schaltbild eines Umrichters in Halbbrückenschaltung, der mit dem erfindungsgemäßen Betriebsverfahren betrieben wird, und

Fig.4

ein Schaltbild eines Umrichters in Vollbrückenschaltung, der mit dem erfindungsgemäßen Betriebsverfahren betrieben wird.

Embodiments of the invention are shown schematically in the drawings and are explained in more detail below. Hereby show:

Fig. 1

a circuit diagram of a Eintransistorumrichters, which is operated with the operating method according to the invention,

Fig. 2

Timing diagrams of signals of the Eintransistorumrichters of Fig. 1 .

Fig. 3

a circuit diagram of a converter in half-bridge circuit, which is operated with the operating method according to the invention, and

Figure 4

a circuit diagram of an inverter in full bridge circuit, which is operated with the operating method according to the invention.

Detailed description of the embodiments

Fig. 1 shows a circuit diagram of an induction heater in the form of a Eintransistorumrichters EU. The induction heating device may also include further, not shown, identically constructed single-transistor converter EU and additional conventional components, such as control elements for power adjustment, etc.

The single-transistor converter EU comprises a bridge rectifier GL, which generates a DC link voltage UG from an AC input voltage UN of 230V and 50Hz, a buffer or DC link capacitor C1 for stabilizing or buffering the intermediate DC voltage UG connected between output terminals N1 and N2 of the rectifier GL Induction coil L1 and a capacitor C2, which are connected in parallel and form a parallel resonant circuit, a controllable switching means in the form of an IGB transistor T1, which is looped in series with the resonant circuit between the output terminals N1 and N2 of the rectifier GL, a freewheeling diode D1, the is connected in parallel with a collector-emitter path of the IGB transistor T1, and a control unit SE, for example in the form of a microprocessor or a digital signal processor.

The control unit SE carries out the invention, hereinafter with reference to Fig. 2 described operating method for operating the Eintransistorumrichters EU and may include other, not shown actuators and / or sensors, for example, for mains voltage monitoring, include or be coupled with these.

Fig. 2 does not show true-to-scale timing diagrams of signals from the single-pole converter EU of Fig. 1 , Due to the mains frequency of the input mains AC voltage UN of 50 Hz, a zero crossing takes place every 10 ms between adjacent mains half-waves H1 to H3 of the input mains AC voltage UN. The single-transistor converter EU is operated in 2/3 mains half-wave operation, ie power is fed into the parallel resonant circuit or into the induction coil L1 only during two out of three mains half-cycles. In Fig. 2 For example, the half-waves H2 and H3 are the active half-waves during which power is applied, and the mains half-wave H1 is the inactive half-wave during which no power feed takes place. During the inactive half cycle H1, the IGB transistor T1 blocks, except for a transition region or predefinable discharge time range INT, during which the intermediate circuit capacitor C1 is discharged.

UC is a voltage at the collector of the IGB transistor T1 with respect to a reference potential applied to the terminal N1 of the rectifier GL. During inactive half-cycles, when the IGBT transistor T1 is turned off, an open circuit voltage with an amount of a peak value of the mains AC voltage UN at the collector, i. in the illustrated embodiment about 325V.

During the active half waves H2 and H3 power is fed into the induction coil L1. This can be effected in a conventional manner, for example by driving the IGB transistor T1 with a square-wave voltage signal having a frequency and a duty cycle, which are adjusted depending on the power to be injected during the half-wave.

In order to prevent a switch-on current pulse during the transition from the half-wave H1 to the half-wave H2, during the discharge time range or time interval INT, starting at a time T0, about 2.5 ms a zero crossing ND between the half-wave H1 and H2 and the zero crossing ND of the DC link capacitor C1 continuously discharged to about 0V by driving the IGB transistor T1. For this purpose, the IGB transistor T1 is driven by a rectangular voltage signal, not shown, with a frequency of about 39 kHz and an on / off ratio of about 1/378. The drive pulses are so short that they are insufficient to clear the charge on the IGB transistor gate. The IGB transistor T1 is therefore not completely turned on, but goes into a linear operation mode. The voltage UC at the collector of the IGB transistor T1, which in this case corresponds to the voltage UG at the intermediate circuit capacitor C1, thereby drops as shown slowly along the network half-wave as an envelope to about 0V. In the in Fig. 2 shown section enlargement, the signal UC is shown with greater temporal resolution. From this the switching frequency of the IGBT of approx. 39kHz becomes visible during the discharging process.

Since the IGBT T1 is not completely conducted or switched through, only a small current through the induction coil L1 results. Noise caused by the coil current is thus prevented or significantly reduced.

During the half-waves H2 and H3, the IGB transistor T1 is driven in a conventional manner with a square-wave voltage signal, not shown. In Fig. 2 is the envelope of the resulting voltage UC and a partial enlargement of the signal UC shown with greater temporal resolution. The voltage UC increases due to the vibration in the parallel resonant circuit to values well above the open circuit voltage. The envelope has a sinusoidal shape, which follows the rectified input AC voltage UN. The course of the voltage UC shown repeats during the half wave H3. The frequency of the drive signal of the IGBT T1 in this operating state is approximately 22 kHz.

In a half-wave, not shown, following the half wave H3, the IGB transistor T1 is deactivated, whereby the voltage UC rises again to its no-load value of approximately 325V. During the transition to a subsequent, active half cycle, the discharge process is repeated, as shown for the half-wave H1. The processes described are repeated periodically.

Consequently, the converter circuit can start with small voltages and currents and adjust with the increase of the mains half-wave to its actual operating point of suitable frequency and duty cycle.

Depending on the IGB transistor used, a drive voltage used to drive it, the capacitance of the link capacitor and the resonant circuit sizing, the discharge frequency and duty cycle can be adjusted to operate the IGB transistor in linear mode during discharge.

As a result of the discharge according to the invention of the intermediate circuit capacitor, power control with half-wave patterns of the single-turn converter EU is possible, without any noise being caused. If, in this case, power is to be output in a half-wave, the DC link capacitor is discharged at the end of the previous, non-active half-cycle. This allows a large power adjustment range without inrush current peaks overstressing the IGB transistor T1. Overall, therefore, increases the life of the components.

Fig. 3 shows a circuit diagram of an inverter HU in half-bridge circuit, which is operated with the operating method according to the invention. Components with compared to Fig. 1 identical function are the same Provided with reference numerals. Regarding their functional description is on Fig. 1 directed.

A half-bridge is formed of IGBTs T2 and T3, which are serially connected between the output terminals N1 and N2 of the rectifier GL. Freewheeling diodes D2 and D3 are connected in parallel to an associated collector-emitter path of the IGBTs T2 and T3, respectively. Capacitors C3 and C4 are also serially connected between the output terminals N1 and N2. Between a connection node N3 of the IGBTs T2 and T3 and a connection node N4 of the capacitors C3 and C4, the induction coil L1 is looped. It forms a series resonant circuit together with the capacitors C3 and C4.

The IGBTs T2 and T3 are driven by the control unit SE. A power setting can be done in a conventional manner, for example by a frequency adjustment of the control signals generated by the control unit SE IGBTs.

After switching on the converter HU and before generating a heating power, the intermediate circuit capacitor C1 and the capacitors C3 and C4 are discharged by driving the IGBTs T2 and T3. This is done analogously to that with reference to Fig. 2 described method by driving the IGBTs T2 and T3 with rectangular voltage signals with a suitable frequency and suitable on / off ratio. Again, the drive pulses are so short that they are insufficient to clear the charge at the respective IGB transistor gate. The IGB transistors T2 and T3 are therefore not completely turned on, but go into a linear operation mode.

In this way, even with a converter in a half-bridge circuit, annoying cracking noises can be effectively prevented during a switch-on process or after a deactivation of the heating power and subsequent renewed activation.

Fig. 4 shows a circuit diagram of an inverter VU in full bridge circuit, which is operated with the operating method according to the invention. Components with compared to Fig. 1 identical function are provided with the same reference numerals. Regarding their functional description is on Fig. 1 directed.

A first half-bridge is formed of IGBTs T4 and T5 and a second half-bridge of IGBTs T6 and T7, which are respectively connected in series between the output terminals N1 and N2 of the rectifier GL. Free-wheeling diodes D4 to D7 are connected in parallel with in each case one associated collector-emitter path of the IGBTs T4 to T7. Between a connection node N5 of the IGBTs T4 and T5 and a connection node N6 of the IGBTs T6 and T7, the induction coil L1 and a capacitor C5 are connected in series. The inductor L1 and the capacitor C5 form a series resonant circuit.

The IGBTs T4 to T7 are driven by the control unit SE. A power setting can be done in a conventional manner, for example by a frequency adjustment of the control signals generated by the control unit SE IGBTs.

After switching on the converter VU and before a heating power generation, the DC link capacitor C1 is discharged by driving the IGBTs T4 to T7. This is done analogously to that with reference to Fig. 2 described method by driving the IGBTs T4 to T7 with square wave signals with a suitable frequency and suitable on / off ratio. The drive pulses are again so short that they are not sufficient to clear the charge at the respective IGB transistor gate. The IGB transistors T4 to T7 are therefore not completely turned on, but go into a linear operation mode.

For discharging the DC link capacitor C1, all the IGBTs T4 to T7 or only certain IGBTs can be driven in such a way that a current path for discharging the DC link capacitor C1 is formed. For example, only T4 and T5, only T6 and T7, only T4 and T7 or only T6 and T5 for discharge can be controlled.

In this way, annoying cracking noises can be effectively prevented at a power-up or after a deactivation of the heating power and subsequent re-activation even in a converter in full bridge circuit.

In the embodiments shown, the mains voltage is 230V and the mains frequency is 50Hz. Of course, the operating method shown can be adapted to other mains voltages and mains frequencies.

Claims (12)

1. Method for operating an induction heating device comprising:

   - an induction coil (L1), and

   - a frequency converter (ET, HU, VU) for generating a control voltage for the induction coil (L1) comprising:

   - a rectifier (GL) which rectifies an alternating supply voltage (UN),

   - an intermediate circuit capacitor (C1) coupled between output terminals (N 1, N2) of the rectifier (GL) and buffering the rectified voltage (UG), and

   - at least one controllable switching element (T1 to T7) coupled between the output terminals (N1, N2) of the rectifier (GL),

   characterized in that

   - in a predeterminable discharge time range (INT) prior to a zero passage (ND) of the alternating supply voltage (UN) the intermediate circuit capacitor (C1) is discharged to a threshold value by controlling the at least one switching element (T1 to T7) before the induction coil (L1) is controlled for generating an adjustable heating power.
2. Method according to claim 1, characterized in that the frequency converter is a single transistor converter (EU).
3. Method according to claim 1, characterized in that the frequency converter is a converter in full bridge circuit (VU) or half-bridge circuit (HU), the at least one switching element (T1 to T7) forming part of the bridge.
4. Method according to one of the preceding claims, characterized in that the discharge time range (INT) commences 1 ms to 5 ms prior to the zero passage (ND) of the alternating supply voltage (UN).
5. Method according to one of the preceding claims, characterized in that the threshold value is between 0 and 20 V.
6. Method according to one of the preceding claims, characterized in that the at least one switching element is a transistor, particularly an IGB transistor (T1 to T7).
7. Method according to claim 6, characterized in that, for discharging the intermediate circuit capacitor (C1) during the discharge the IGB transistor (T1 to T7) is controlled such that the IGB transistor (T1 to T7) operates in a linear operating state.
8. Method according to one of the preceding claims, characterized in that the at least one switching element (T1 to T7) is controlled by a pulse-width modulated square-wave voltage signal for discharging the intermediate circuit capacitor (C1).
9. Method according to claim 8, characterized in that the square-wave voltage signal has a frequency of 20 to 50 kHz.
10. Method according to claim 8 or 9, characterized in that the square-wave voltage signal has an on/off ratio in the range of 1/300 to 1/500.
11. Method according to one of the preceding claims, characterized in that the adjustable heating power is generated using a half-wave pattern, the intermediate circuit capacitor (C1) being discharged prior to an activation of a half-wave.
12. Method according to claim 11, characterized in that one of three or two of three half-waves are activated.

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