EP1485984B1 - Regulateur multiphase a commutation - Google Patents

Regulateur multiphase a commutation Download PDF

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Publication number
EP1485984B1
EP1485984B1 EP02789920A EP02789920A EP1485984B1 EP 1485984 B1 EP1485984 B1 EP 1485984B1 EP 02789920 A EP02789920 A EP 02789920A EP 02789920 A EP02789920 A EP 02789920A EP 1485984 B1 EP1485984 B1 EP 1485984B1
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Prior art keywords
output
circuit
summing
voltage
inductors
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German (de)
English (en)
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EP1485984A2 (fr
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Tod Schiff
Joseph C. Buxton
Richard Elfi S.A. REDL
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Analog Devices Inc
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Analog Devices Inc
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H02M3/1584—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load with a plurality of power processing stages connected in parallel

Definitions

  • This invention relates to the field of switching voltage regulators, and particularly to multi-phase switching voltage regulators.
  • Switching voltage regulators provide a predetermined and substantially constant output voltage from a source voltage that may be fluctuating, or that may be at an inappropriate amplitude for the load.
  • Such regulators typically employ one or more switching elements.
  • the switching elements may be, for example, field-effect transistor (FET) switches.
  • Control circuitry regulates the current supplied to the load by varying the ON-OFF times of the switching elements (i.e., the regulator's duty cycle, which is the percentage of time that a switch is ON during a cycle of operation).
  • Inductors and capacitors are typically used to convert the switched current pulses into a steady flow of load current. Switching regulators are frequently used in portable battery-powered electronic products, as they provide high operating efficiency and long battery life with little heat generation.
  • a switching voltage regulator may be arranged to operate using one or more different regulation modes. For example, for voltage-mode regulation, a feedback signal V fb which varies with the regulator's output voltage is fed to a comparator along with a periodic sawtooth waveform; the comparator's output controls the duty cycle of the switching elements. For current-mode regulation, a voltage V i is generated which is proportional to the current in the output inductor. V i and V fb are provided as inputs to a comparator, the output of which controls the duty cycle of the switching elements.
  • Advanced microprocessors require power supplies that provide a low voltage (e.g., ⁇ 2 volts) at high current.
  • a low voltage e.g., ⁇ 2 volts
  • One way in which high currents are provided is with a multi-phase switching regulator, in which the components of a number (N) of single-phase regulators are repeated to produce N output currents, which are summed together to provide the total output current.
  • N number
  • implementing any of the regulators cited above as a multi-phase regulator requires summing and/or sampling data in the controller IC, which can introduce unwanted complexity and processing delays into the IC design.
  • AVP adaptive voltage positioning
  • Non-patent publication entitled " A new control for multi-phase buck converter with fast transient response” (Wenkang Huang; XP010536007; ISBN:0-7803-6618-2 ) relates to a three-phase constant frequency synchronous buck controller.
  • the three phases are interleaved, and each phase is delayed by 120° from the previous phase. Every phase has a current sense amplifier.
  • Current signals from sensing resistors are connected to three current sense amplifiers, the outputs of which are summed with the fast feedback voltage before connecting to the inputs of pulse width modulation (PWM) comparators.
  • PWM pulse width modulation
  • a multi-phase switching voltage regulator is presented which overcomes the problems noted above, enabling precise multiple-loop control which is particularly well suited to AVP power supplies.
  • the invention provides a "total current circuit", which makes use of an amplifier circuit to sum together individual phase currents to generate a signal representing the total instantaneous output current in the inductors I out .
  • N current sensing elements - which can be the output inductors themselves or current sense resistors connected in series with the inductors - carry respective phase currents.
  • a summing circuit preferably made from N resistors each having a resistance R p , is connected to sum the voltages present at the switch node sides of the sensing elements, and to provide the summed voltages at a summing node.
  • a single amplifier circuit is connected to the summing node at a first input and to the regulator's output terminal at a second input, and produces an output voltage V cs .
  • V out When V out is subtracted from V cs , the result is proportional to I out .
  • the amplifier circuit preferably comprises an operational amplifier, with a feedback resistor having a resistance R cs , and a filter capacitor having a capacitance C cs connected across the feedback resistor.
  • the invention also provides a unique means for direct insertion of total inductor output current information into a switching regulator's voltage-mode control loop, to provide AVP for the output voltage.
  • a desired output voltage is specified by providing a reference voltage V ref , and a current circuit - preferably the total current circuit described above - produces an output voltage V droop which varies with the the regulator's total filtered instantaneous output current I out (fltr) .
  • V set and V out are presented to the inputs of a voltage control error amplifier circuit, which produces an output V err that varies with the difference between V set and V out .
  • a control circuit operates the regulator's switches in response to V err .
  • the control circuit, total current circuit, summation circuit, and voltage control error amplifier circuit are arranged such that the regulator's output has a desired droop impedance R o , to provide AVP of V out as a function of the total filtered instantaneous inductor output current I out(fltr) .
  • This serves to directly insert instantaneous inductor output current information into the reference voltage input of the regulator's voltage control error amplifier, which enables fast, accurate control of output voltage for dynamic changes in load.
  • FIG. 1 A schematic/block diagram illustrating the principles of a total current circuit for a multi-phase switching regulator in accordance with the present invention is shown in FIG. 1 .
  • the regulator includes a switching element (10, 12) made up of one or more switching transistors, with each switching element connected to the "switch node" side of a respective output inductor (14, 16).
  • the opposite, "output" sides of the inductors are connected together at a common output terminal 18.
  • the N switching elements are operated by a controller 20, which operates the elements so as to provide an output voltage V out at output terminal 18 and a total inductor output current I out to a load 22.
  • Load 22 normally includes a parallel filter capacitor (not shown) to reduce the ripple voltage component of the output voltage V out .
  • the present total current circuit includes N "current sensing elements".
  • the current sensing elements may be the output inductors themselves, or current sense resistors (23, 24) connected in series between each inductor and output terminal 18.
  • Each current sensing element carries a respective phase current I px .
  • a Summing circuit 25 is connected to sum the N voltages present at the switch node sides of the circuit's current sensing elements: when the sensing elements are the inductors, circuit 25 is connected to the inductors' switch node sides; when sense resistors are used, circuit 25 is connected to the resistors' inductor sides. The summed voltages are provided at a summing node 26.
  • the present total current circuit also includes an amplifier circuit 28, which is connected to summing node 26 at a first input and to the regulator's output voltage V out .
  • Amplifier circuit 28 is arranged to sum together the individual phase currents to generate an output voltage (V cs ) which directly represents the total instantaneous inductor output current I out .
  • Output voltage V cs can then be used, for example, as an input to controller 20 to provide current-mode control of the regulator, or as an input to a current limiting circuit.
  • Summing circuit 25 is preferably implemented with N resistors, each of which has a resistance R p and is connected to the switch node side of a respective current sensing element at one end and to summing node 26 at its other end.
  • Amplifier circuit 28 preferably comprises an operational amplifier A1, having its inverting input connected to summing node 26 and its non-inverting input connected to V out .
  • a feedback resistor having a resistance R cs is connected between the A1's output and inverting input; R cs sets the A1's gain, and the load line required by the system being powered.
  • a filter capacitor having a capacitance C cs is preferably connected across R cs .
  • V cs V out - R cs ⁇ R l ⁇ 1 + s / R l L R p ⁇ 1 + s ⁇ C cs ⁇ R cs ⁇ I out , where R 1 and L are the resistance and inductance values, respectively, for output inductors 14 and 16. Equation 1 presumes that R 1 and L are approximately equal for all output inductors and that all summing resistors have a resistance approximately equal to R p ; those skilled in the art can calculate V cs precisely for non-equal values.
  • V out is subtracted from V cs , the result is proportional to the regulator's total instantaneous inductor output current I out , with the proportionality varying with frequency.
  • the time constant C cs R cs is made substantially equal to the time constant L/R 1 of each output inductor (again presuming that R 1 and L are approximately equal for all output inductors)
  • V droop is directly proportional to total inductor output current I out .
  • V droop can be set by properly selecting the value of R cs , which is also used to establish the regulator's droop impedance R o .
  • Signals V cs and/or V droop can be used, for example, as a feedback signal provided to controller 20 for one or more of the regulator's control loops, or for monitoring the output current with respect to a current limit.
  • V cs and/or V droop could also be used as a feedback signal in a switching regulator which employs AVP. Note, however, that the present total current circuit can be used with any single or multi-phase switching regulator, with or without AVP.
  • the V droop signal can be used to implement AVP, giving the regulator a finite output impedance.
  • the present total current circuit offers high speed and accurate continuous sensing of total inductor output current. Sensing the total inductor current at the output is an inherently more accurate method than other current sensing methods such as peak current detection in the high side MOSFET or sampling the current across a sense element such as the low side MOSFET.
  • the resistance R 1 of the output inductors is due to their copper windings.
  • R 1 has an associated temperature dependence, represented herein with a temperature coefficient t c .
  • feedback resistance R cs can be selected to have a temperature coefficient which is approximately equal to -t c .
  • Making the temperature coefficient of R cs equal to -t c also provides continuous time constant tracking; i.e., time constant C cs R cs remains substantially equal to time constant L/R 1 over temperature.
  • Amplifier circuit 28 can be realized with, for example, a current feedback or voltage feedback op amp,a transconductance amplifier, or any number of other implementations.
  • Amplifier circuit 28 preferably has a monotonic relationship between its output voltage and the difference between its two inputs.
  • the amplifier also preferably has a low offset input voltage to provide good accuracy, and a high gain (if it is an op amp) or a constant gain (if it is a transconductance amplifier).
  • the regulator includes a switching element (100, 102) made up of one or more switching transistors, with each switching element connected to the switch node side of a respective output inductor (104, 106). The opposite, “output" sides of the inductors are connected together at a common output terminal 108.
  • the switching elements are operated by a control circuit 110, which operates the elements so as to provide an output voltage V out at output terminal 108 and a total inductor output current I out to a load 112.
  • AVP is conventionally accomplished by inserting total inductor output current information into the output voltage feedback loop as an error signal.
  • This approach can result in sluggish and imprecise control of the output voltage for dynamic changes in the load.
  • the invention overcomes this by inserting total inductor output current information directly into the reference voltage input of the regulator's voltage control error amplifier 114.
  • total current signal V tc offsets reference voltage V ref to reposition output voltage V out based on the total inductor output current I out , thereby providing AVP.
  • This approach provides faster feedback of the total inductor output current information into the voltage-mode control loop, when compared with prior art methods which insert the total inductor output current information into the output voltage feedback as an error signal.
  • FIG. 4 is a diagram of a preferred embodiment of a voltage-mode control loop for a multi-phase switching regulator per the present invention.
  • Voltage control error amplifier circuit 114 preferably comprises an operational amplifier A2, which receives voltage V set from summation circuit 116 at its non-inverting input, and a voltage which represents V out at its inverting input.
  • a loop compensation network 118 is connected between A2's output and its inverting input. Since the reference voltage provided to summation circuit 116 is often provided by a digital-to-analog converter (DAC), V ref is identified as "V dac " in FIG. 4 .
  • DAC digital-to-analog converter
  • V out V dac - I out fltr ⁇ R o , where I out(fltr) is the total inductor current without its ripple component, i.e. the filtered total inductor current.
  • the desired output voltage V dac is offset by the output droop impedance (R o ) times the filtered output current (I out(fltr) ), to provide AVP of output voltage V out .
  • A2's inverting input may be connected to V out directly for a transconductance amplifier, or through a series resistance R b for an operational amplifieras shown in FIG. 4 .
  • the resistor R b and a current source 120 which causes a current I fb to flow through R b sets the no-load offset voltage from the reference voltage (V dac ) in AVP converters, such that the regulated output voltage V out is given by:
  • the novel voltage-mode control loop preferably receives voltage V tc from the total current circuit described above. This is illustrated with reference to the multi-phase switching regulator shown in FIG. 5 .
  • the regulator includes a switching element (200, 202) made up of one or more switching transistors, with each switching element connected to the switch node side of a respective output inductor (204, 206). The opposite sides of the inductors are connected together at a common output terminal 208.
  • the switching elements are operated by a control circuit 210, which operates the elements so as to provide an output voltage V out at output terminal 208 and a total inductor output current I out to a load 212.
  • a total current circuit in accordance with the present invention produces voltage V droop , which is proportional to the regulator's total inductor output current I out .
  • Summing circuit 25 sums the voltages at the switch node sides of output inductors 204 and 206 at a summing node 26. The summed voltages are fed to amplifier circuit 28, preferably implemented with op amp A1 and feedback resistance and capacitance R cs and C cs , respectively.
  • Op amp A1 also receives output voltage V out at an input, and produces voltage V cs at its output.
  • Subtractor circuit 30 subtracts V out from V cs to produce V droop .
  • V droop is provided to the summation circuit 30 of a voltage-mode control loop per the present invention, along with reference voltage V ref which represents the desired output voltage. V droop thus offsets V ref , with voltage V set resulting. V set is provided to voltage control error amplifier 114, which is preferably implemented with op amp A2, loop compensation network 118, current source 120, resistance R b , and capacitance C b . The output of A2 is provided to control circuit 210 to provide the regulated output voltage with AVP.
  • V out V dac - I out fltr ⁇ R o - I fb ⁇ R b
  • This arrangement provides a novel method for creating a signal (V droop ) proportional to total filtered inductor output current (I out(fltr) ), as well as a unique method of inserting the output current information into the voltage control loop for providing AVP of the output.
  • FIG. 5 depicts a two-phase regulator
  • the invention is intended to encompass all single and multi-phase converters, with or without AVP.
  • FIG. 6 One possible implementation of a control circuit 210, suitable for use with the present invention to form a multi-phase switching regulator, is shown in FIG. 6 .
  • the output inductors, total current circuit, and voltage-mode control loop circuit are as shown in FIG. 5 .
  • the control circuit includes a pair of gate driver circuits 300, 302 which provide the signals that operate the switching elements (200, 202). Additional control circuitry processes the feedback signal produced by voltage control error amplifier A2 and drives the gate driver circuits.
  • the regulator shown in FIG. 6 provides multiple feedback loop control with AVP for a two-phase converter; note that this configuration can be extended to any number of phases.

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  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)

Claims (10)

  1. Circuit de courant total pour un régulateur de commutation à N phases qui fournit une tension de sortie Vout au niveau d'une borne de sortie (18) et qui comporte N bobines d'induction de sortie (14, 16) portant des courants de phase respectifs qui sont additionnés ensemble pour fournir un courant de sortie de bobines d'induction instantané total Iout à une charge (22) connectée à ladite borne de sortie, ledit circuit de courant total comprenant :
    N éléments de détection de courant (14, 16, 23, 24) qui portent des courants respectifs desdits courants de phase, chacun desdits éléments ayant un côté noeud de commutation et un côté sortie, les côtés sortie desdits éléments étant connectés à la borne de sortie dudit régulateur,
    un circuit de sommation (25) qui additionne les tensions présentes au niveau des côtés noeud de commutation desdits éléments de détection et fournit lesdites tensions additionnées au niveau d'un noeud de sommation (26), et
    un circuit d'amplificateur unique (28) qui est connecté audit noeud de sommation au niveau d'une première entrée et à ladite borne de sortie au niveau d'une seconde entrée et est agencé pour produire une tension de sortie Vcs de telle sorte que Vcs - Vout soit proportionnelle à Iout.
  2. Circuit de courant total selon la revendication 1, dans lequel ledit circuit de sommation comprend N résistances de sommation (Rp), qui sont chacune connectées au niveau d'une extrémité au côté noeud de commutation d'un élément respectif desdits éléments de détection de courant, et connectées au niveau de leur autre extrémité audit noeud de sommation.
  3. Circuit de courant total selon la revendication 1, dans lequel lesdits N éléments de détection de courant sont lesdites N bobines d'induction de sortie (14, 16).
  4. Circuit de courant total selon la revendication 1, dans lequel lesdits N éléments de détection de courant sont N résistances de détection de courant (23, 24) connectées entre des bobines d'induction respectives desdites N bobines d'induction de sortie et ladite borne de sortie et ledit circuit de sommation additionne les tensions présentes au niveau des côtés bobines d'induction de sortie desdites résistances de détection de courant.
  5. Circuit de courant total selon la revendication 1, dans lequel ledit circuit amplificateur comprend :
    un amplificateur opérationnel (A1), lesdites première et seconde entrées étant les entrées inverseuse et non inverseuse dudit amplificateur opérationnel, respectivement, et
    une résistance de contre-réaction ayant une résistance Rcs connectée entre la sortie dudit amplificateur opérationnel et ladite entrée inverseuse qui règle le gain dudit amplificateur opérationnel.
  6. Circuit de courant total selon la revendication 5, dans lequel lesdits N éléments de détection de courant sont lesdites N bobines d'induction de sortie (14, 16), lesquelles ont chacune un premier coefficient de température tc, ladite résistance de contre-réaction Rcs ayant un second coefficient de température qui est approximativement égal à -tc.
  7. Circuit de courant total selon la revendication 5, dans lequel chacune desdites N bobines d'induction de sortie a une inductance L et une résistance R1 et ledit circuit de sommation comprend N résistances de sommation, lesquelles ont chacune une résistance Rp, et sont connectées au niveau d'une extrémité au côté noeud de commutation d'une bobine d'induction respective desdites bobines d'induction de sortie, et connectées au niveau de leur autre extrémité audit noeud de sommation,
    comprenant en outre un condensateur de filtrage ayant une capacité Ccs connecté aux bornes de ladite résistance de contre-réaction, de telle sorte que la tension de sortie Vcs dudit amplificateur opérationnel soit donnée par : V cs = V out - R cs ⁢ R l ⁢ 1 + s / R l L R p ⁢ 1 + s ⁢ C cs ⁢ R cs ⁢ I out .
    Figure imgb0016
  8. Circuit de courant total selon la revendication 7, dans lequel chacune desdites N bobines d'induction a une constante de temps L/R1 et le produit CcsRcs est rendu sensiblement égal à L/R1 de telle sorte que la tension de sortie Vcs dudit amplificateur opérationnel soit donnée par : V cs = V out - R l ⁢ R cs R p ⁢ I out .
    Figure imgb0017
  9. Circuit de courant total selon la revendication 1, comprenant en outre un circuit soustracteur (30) qui soustrait ladite tension de sortie Vout de Vcs.
  10. Circuit de courant total selon la revendication 1, dans lequel lesdits N éléments de détection de courant sont lesdites N bobines d'induction de sortie (14, 16), le circuit de sommation (25) consistant en N résistances de sommation (Rp), lesquelles sont chacune connectées au niveau d'une extrémité au côté noeud de commutation d'une bobine d'induction respective desdites bobines d'induction de sortie et connectées au niveau de leur autre extrémité à un noeud de sommation (26),
    ledit circuit d'amplificateur unique comprenant un amplificateur opérationnel (A1) qui est connecté audit noeud de sommation au niveau de son entrée inverseuse et à la borne de sortie dudit régulateur au niveau de son entrée non inverseuse, ledit circuit de courant total comprenant en outre :
    une résistance de contre-réaction ayant une résistance Rcs connectée entre la sortie dudit amplificateur opérationnel et ladite entrée inverseuse qui règle le gain dudit amplificateur opérationnel, et
    un condensateur de filtrage ayant une capacité Ccs connectée aux bornes de ladite résistance de contre-réaction,
    dans lequel chacune desdites N résistances de sommation a une résistance Rp, et chacune desdites N bobines d'induction de sortie a une inductance L et une résistance R1, de telle sorte que la tension de sortie Vcs dudit amplificateur opérationnel soit donnée par : V cs = V out - R cs ⁢ R l ⁢ 1 + s / R l L R p ⁢ 1 + s ⁢ C cs ⁢ R cs ⁢ I out .
    Figure imgb0018
EP02789920A 2001-11-26 2002-11-25 Regulateur multiphase a commutation Expired - Fee Related EP1485984B1 (fr)

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US33340401P 2001-11-26 2001-11-26
US333404P 2001-11-26
US284222 2002-10-30
US10/284,222 US6683441B2 (en) 2001-11-26 2002-10-30 Multi-phase switching regulator
PCT/US2002/038148 WO2003049266A2 (fr) 2001-11-26 2002-11-25 Regulateur multiphase a commutation

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EP1485984A2 (fr) 2004-12-15
US20030102849A1 (en) 2003-06-05
WO2003049266B1 (fr) 2004-05-13
USRE39976E1 (en) 2008-01-01
US6683441B2 (en) 2004-01-27
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