EP1352302A1 - Voltage regulator with static gain in reduced open loop - Google Patents

Voltage regulator with static gain in reduced open loop

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Publication number
EP1352302A1
EP1352302A1 EP01995774A EP01995774A EP1352302A1 EP 1352302 A1 EP1352302 A1 EP 1352302A1 EP 01995774 A EP01995774 A EP 01995774A EP 01995774 A EP01995774 A EP 01995774A EP 1352302 A1 EP1352302 A1 EP 1352302A1
Authority
EP
European Patent Office
Prior art keywords
output
operational amplifier
transistors
voltage regulator
transistor
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Withdrawn
Application number
EP01995774A
Other languages
German (de)
French (fr)
Inventor
Cécile HAMON
Christophe Bernard
Alexandre Pons
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
STMicroelectronics SA
Original Assignee
STMicroelectronics SA
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Filing date
Publication date
Application filed by STMicroelectronics SA filed Critical STMicroelectronics SA
Publication of EP1352302A1 publication Critical patent/EP1352302A1/en
Withdrawn legal-status Critical Current

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Classifications

    • G—PHYSICS
    • G05—CONTROLLING; REGULATING
    • G05F—SYSTEMS FOR REGULATING ELECTRIC OR MAGNETIC VARIABLES
    • G05F1/00—Automatic systems in which deviations of an electric quantity from one or more predetermined values are detected at the output of the system and fed back to a device within the system to restore the detected quantity to its predetermined value or values, i.e. retroactive systems
    • G05F1/10—Regulating voltage or current
    • G05F1/46—Regulating voltage or current wherein the variable actually regulated by the final control device is dc
    • G05F1/56—Regulating voltage or current wherein the variable actually regulated by the final control device is dc using semiconductor devices in series with the load as final control devices
    • G05F1/565—Regulating voltage or current wherein the variable actually regulated by the final control device is dc using semiconductor devices in series with the load as final control devices sensing a condition of the system or its load in addition to means responsive to deviations in the output of the system, e.g. current, voltage, power factor
    • G—PHYSICS
    • G05—CONTROLLING; REGULATING
    • G05F—SYSTEMS FOR REGULATING ELECTRIC OR MAGNETIC VARIABLES
    • G05F1/00—Automatic systems in which deviations of an electric quantity from one or more predetermined values are detected at the output of the system and fed back to a device within the system to restore the detected quantity to its predetermined value or values, i.e. retroactive systems
    • G05F1/10—Regulating voltage or current
    • G05F1/46—Regulating voltage or current wherein the variable actually regulated by the final control device is dc
    • G05F1/56—Regulating voltage or current wherein the variable actually regulated by the final control device is dc using semiconductor devices in series with the load as final control devices
    • G05F1/575—Regulating voltage or current wherein the variable actually regulated by the final control device is dc using semiconductor devices in series with the load as final control devices characterised by the feedback circuit

Definitions

  • the present invention relates to the field of voltage regulators and in particular that of regulators with low waste voltage.
  • a low drop out regulator in the form of an integrated circuit can be used to supply a predetermined potential with low noise to a set of electronic circuits from a supply potential supplied by a rechargeable battery. .
  • a supply potential decreases over time and is likely to include noise caused by the action of neighboring electromagnetic radiation on the battery / regulator links.
  • the regulator is said to have a low waste voltage because it provides a potential close to the supply potential.
  • Figure 1 shows schematically an example of a conventional low-voltage waste regulator.
  • the regulator has an output terminal S intended to be connected to a load R.
  • the essentially resistive load R represents the sum of the input impedances of the circuits supplied by the regulator. For simplicity, it is subsequently considered that the load R is a resistance.
  • the regulator comprises an operational amplifier 4 of which a non-inverting input IN + is connected to a positive reference potential Vref and of which an inverting input IN ⁇ is connected to the terminal S by a feedback loop.
  • the potential Vref is produced in a known manner by a constant voltage source (not shown) with high output impedance.
  • the operational amplifier 4 is supplied between a positive supply potential Vbat supplied by the battery and a ground potential GND.
  • An inverter stage 6, supplied between the potentials Vbat and GND, receives the output of the operational amplifier 4 and its output is connected to the gate of a power MOS transistor T1, with P channel, the drain of which is connected to the output terminal S and whose source is connected to the potential Vbat.
  • the transistor T1 is of the MOS type rather than bipolar in particular to minimize the difference between the output potential Vout of the terminal S and the supply potential Vbat.
  • a charge capacitor C is disposed between the output terminal S and the potential GND.
  • FIG. 2 schematically represents an exemplary embodiment of the operational amplifier 4 of FIG. 1.
  • Two MOS transistors T2, T3, with P channel, have their • sources connected to each other and their gates respectively connected to the inputs IN ⁇ and IN +.
  • a bias current source CS1 is arranged between the potential Vbat and the sources of the transistors T2 and T3.
  • the transistors T2 and T3 form a differential pair.
  • Two N-channel MOS transistors T4 and T5 have their sources connected to the GND potential and their gates connected to each other.
  • the drains of the transistors T4 and T5 are respectively connected to the drains of the transistors T2 and T3.
  • the drain of transistor T3 is connected to the gates of transistors T4 and T5.
  • the transistors T4 and T5 form an active charge of the differential pair formed by the transistors T2 and T3.
  • the drain of transistor T2 constitutes the output of amplifier 4.
  • the voltage regulator of FIG. 1 maintains the potential Vout of the output terminal S at a value equal to the reference potential Vref. Any variation in the potential Vbat results in a variation in the potential Vout, which is transmitted by the feedback loop on the input IN " . When the regulator functions correctly, the variation of the potential of the input IN " causes the potential Vout to return to the potential Vref.
  • the regulator circuit which forms a looped system between the input IN " and the terminal S must be a stable system. For the system to be stable when it is looped, its gain in open loop must not be greater to 1 when the phase shift is less than -180 ° (when there is phase opposition between the input and the output of the system).
  • Figure 3 illustrates, according to the frequency f, the variation of the gain G and the phase shift ⁇ of the open loop regulator taken between input IN " and terminal S.
  • the gain G is equal to the static gain Gs of the regulator in open loop.
  • the elements that make up the regulator each have a gain which varies according to the frequency.
  • the cut-off frequency of an element whose gain decreases when the frequency increases constitutes a "pole" of the transfer function of the regulator in open loop.
  • Each pole of the transfer function of the open loop regulator introduces a fall of 20 dB per decade of gain G.
  • each pole of the transfer function of the open loop regulator introduces a phase shift ⁇ of 90 °.
  • the transfer function of the open loop regulator comprises only a main pole PO and a secondary pole PI.
  • the frequency at which the main pole PO is located depends in particular on the inverse of the product of the values of the load resistance R and the capacitor C.
  • the frequency at which the secondary pole PI is located depends in particular on the impedance of the grid of the transistor T1.
  • the inverting stage 6 is an ideal stage which does not introduce any pole.
  • the characteristics of the elements that make up the regulator are chosen so that, when the phase shift ⁇ becomes equal to -180 °, the gain G is less than the unit gain (0 dB).
  • the PO pole is located at a low frequency and the PI pole is located at a frequency higher than the frequency of the PO pole.
  • the gain is equal to the static gain Gs of the open loop regulator. Between the PO and PI poles, the gain drops by 20 decibels per decade. Beyond the PI pole, the gain drops by 40 decibels per decade. The phase shift drops from 0 to -90 ° at the PO pole and from -90 ° to - 180 ° at the PI pole.
  • the static gain Gs of the regulator is equal to Gs4.Gs6.Gsl, where Gs4 is the static gain of the operational amplifier 4, Gs6 is the static gain of the inverter stage 6, and Gsl is the static gain of the transistor Tl.
  • the static gain of the operational amplifier 4 is of the form:
  • GS4 Grr-2.
  • (R2.R4) / (R2 + R4) ( ⁇ 2. Zout where .3 ⁇ 2 is the transconductance of the transistor T2, and R2, R4 are the conduction resistors, called “Early" resistors, of the transistors T2 and T4
  • the ratio (R2.R4) / (R2 + R4) constitutes the output impedance Zout of the operational amplifier.
  • the "Early" resistances of the transistors T2 and T4 are high, and the output impedance Zout as well as the static gain Gs4 of the amplifier 4 have a high value.
  • a strong Gs4 gain makes the static Gs gain high, which shifts the gain curve upwards and makes the stability of the regulator difficult to obtain.
  • FIG. 3 illustrates a gain curve G 'of an open loop regulator having the two poles PO, PI above and having a static gain Gs' greater than the previous static gain Gs.
  • the gain G ' is greater than 1 (0 dB) when the phase shift ⁇ reaches the value of -180 °, which makes the regulator unstable.
  • a conventional way of solving this problem is to increase the capacitance of the capacitor C, which reduces the frequency at which the main pole PO is located.
  • the use of a large capacitor C is not desirable.
  • An object of the present invention is to provide a stable voltage regulator with a large passband while using a low value output capacitor. To achieve this object, the present invention provides for reducing the apparent output resistance of the operational amplifier of a regulator.
  • the present invention provides a voltage regulator having an output terminal suitable for being connected to a load, comprising an operational amplifier whose non-inverting input is connected to a first reference potential, and whose inverting input is connected at the output terminal, an inverter stage the input of which is connected to the output of the operational amplifier, a power switch controlled by the output of the inverter stage, disposed between the output terminal and a supply potential , and a charge capacitor disposed between the output terminal and a supply reference potential, comprising means for reducing the effective output impedance of the operational amplifier.
  • the means of reduction of -pumpedance comprises a first resistor, a first terminal of which is connected to the output of the operational amplifier, a MOS transistor connected as a diode, the drain of which is connected to a second terminal of the first resistor and whose source is connected to the second reference potential, and means for biasing the transistor connected as a diode in the on state.
  • the first resistance has a value much lower than the output impedance of the operational amplifier.
  • the operational amplifier comprises first and second MOS transistors, of a first type, the sources of which are connected to each other, and the gates of which are respectively connected to the inputs inverting and non-inverting, a current source disposed between the supply potential and the sources of the first and second transistors, of the third and fourth MOS transistors, of a second type, the sources of which are connected to the first reference potential, of which the gates are connected to each other, and the drains of which are respectively connected to the drains of the first and second transistors, the drain of the first transistor being connected to the output of the operational amplifier and the drain and the gate of the fourth transistor being connected to each other.
  • the inverter stage comprises a fifth MOS transistor, of the type of the third and fourth transistors, the gate and the drain of which are respectively connected to the input and to the output of the inverter stage , and the source of which is connected to the first reference potential, an impedance arranged between the output of the inverter stage and the supply potential, and a capacitor and a second resistor arranged in series between the input and the output of 1 'reverse stage.
  • the power switch is a sixth MOS transistor of the type of the first and second transistors.
  • the first, second and sixth transistors are P-channel MOS
  • the third, fourth and fifth transistors are N-channel MOS
  • FIG. 4 schematically represents an embodiment of a regulator according to the present invention
  • FIG. 5 schematically represents an embodiment of an inverter usable according to the present invention. Only the elements allowing the understanding of the present invention have been represented in the various figures. The same references represent the same elements in the different figures.
  • FIG. 4 schematically represents an embodiment of a regulator.
  • the regulator comprises the elements already described of a conventional regulator and a circuit 7 for reducing the output impedance connected to the output of the operational amplifier 4.
  • a resistor RI has a first terminal connected to the output of the operational amplifier 4.
  • An MOS transistor 8 with N channel, has its drain connected to a second terminal of the resistor RI and its source connected to the potential GND. The drain and the gate of transistor 8 are connected to each other so that transistor 8 is connected as a diode.
  • a current source CS2 for biasing the transistor 8 connected as a diode is connected between the potential Vbat and the drain of the transistor 8.
  • the current source CS2 is chosen so that the transistor 8 connected as a diode conducts permanently.
  • the transistor 8 is chosen so that the voltage drop between its drain and its source is equal to the voltage existing between the input of the inverter stage 6 and the ground potential GND. It follows that the voltage drop across the resistor RI is substantially zero, and that the operation of the operational amplifier 4 is not unbalanced by a current flowing through the resistor RI.
  • the impedance Z of the transistor 8 connected in diode and of the resistance RI connected in series is equal to: where G m 8 is the transconductance of the transistor 8.
  • the resistance RI and the transistor 8 are chosen so that the impedance Z is much less than the output impedance Zout of the operational amplifier.
  • the static gain Gs4 of the operational amplifier 4 whose output OUT is connected in parallel on the impedance Z is equal to Gs4 ***** G ⁇ . (Zout .Z) / (Zout + Z), i.e. substantially G ⁇ . .
  • the present invention makes it possible to reduce the static gain introduced by the operational amplifier 4, and thereby the static gain of the open loop voltage regulator.
  • the reduction in the apparent output impedance of the operational amplifier 4 corresponds to a reduction in the gain of this amplifier.
  • the present invention has been described in relation to an ideal inverter stage 6 which does not introduce any pole in the transfer function of the open loop voltage regulator.
  • the inverter stage 6 is not an ideal amplifier stage, but is, for example, a so-called "Miller" amplifier stage.
  • the function of such an amplifier stage is notably to increase the frequency at which the secondary pole PI is located in order to increase the bandwidth of the open loop voltage regulator.
  • a Miller stage notably introduces a pole P2 and a zero Zl into the transfer function of the open loop voltage regulator.
  • FIG. 5 schematically represents an embodiment of a voltage regulator according to the present invention, in which the inverter stage 6 of the amplification circuit 2 'is a Miller stage.
  • the inverter stage 6 comprises a transistor T7, with an N channel, the gate and the drain of which are respectively connected to the input and to the output of the stage 6.
  • the source of the transistor T7 is connected to the potential GND.
  • An impedance 10 is disposed between the output of stage 6 and the potential Vbat.
  • a capacitor C1 and a resistor R2 are arranged in series between the input and the output of the amplifier stage. The value of the capacitor Cl, of the resistor R2, and the gain of the transistor T7 make it possible in particular to adjust the frequencies at which the poles PI, P2 are located.
  • the voltage drop across the terminals of the transistor 8 connected as a diode is in this case chosen to be equal to the gate / source voltage of the transistor T7.
  • the reduction in the output impedance connected at the input of the inverter stage 6 also has the effect of increasing the frequency at which the pole P2 introduced by stage 6 is situated, which represents an additional advantage of the present invention. .
  • the present invention is susceptible of various variants and modifications which will appear to one skilled in the art.
  • the present invention has been described in relation to a particular operational amplifier, but a person skilled in the art will easily adapt the present invention to a voltage regulator using other types of operational amplifiers.
  • the present invention has been described in relation to a voltage regulator using a power transistor T1, but a person skilled in the art will easily adapt the present invention to a voltage regulator using another type of voltage-controlled power switch.
  • the present invention has been described in relation to positive Vbat and Vref potentials, but those skilled in the art will easily adapt the present invention to negative Vbat and Vref potentials, by reversing the types of transistors. tors MOS described and the connection of the transistor 8 connected in diode.
  • the present invention has for reasons of simplicity been described in relation to a voltage regulator using a non-resistive feedback loop and supplying a voltage equal to a reference voltage Vref received.
  • a person skilled in the art will easily adapt the present invention to a voltage regulator, the feedback loop of which comprises a resistive bridge, and which supplies an output voltage different from the voltage Vref received.

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  • Engineering & Computer Science (AREA)
  • Physics & Mathematics (AREA)
  • Electromagnetism (AREA)
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  • Automation & Control Theory (AREA)
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Abstract

The invention concerns a voltage regulator having an output terminal (S) designed to be connected to a load (R), comprising an operational amplifier (4) whereof the non-inverting input is connected to a first reference potential (Vref), and whereof the inverting input is connected to the output terminal (S), and inverting stage (6) whereof the input is connected to the operational amplifier output, a power switch (T1) controlled by the inverter stage (6) output, arranged between the output terminal (S) and a supply potential (Vbat), and a load capacitor (C) arranged between the output terminal (S) and a supply reference potential (GND), including means (7) for reducing the effective output impedance of the operational amplifier (4).

Description

REGULATEUR DE TENSION A GAIN STATIQUE EN BOUCLE OUVERTE REDUIT REDUCED OPEN LOOP STATIC GAIN VOLTAGE REGULATOR
La présente invention concerne le domaine des régulateurs de tension et en particulier celui des régulateurs à faible tension de déchet.The present invention relates to the field of voltage regulators and in particular that of regulators with low waste voltage.
Un régulateur à faible tension de déchet (Lo Drop Out) réalisé sous forme de circuit intégré peut être utilisé pour fournir un potentiel prédéterminé avec un faible bruit à un ensemble de circuits électroniques à partir d'un potentiel d'alimentation fourni par une pile rechargeable. Un tel potentiel d'alimentation décroît avec le temps et est susceptible de comporter du bruit causé par l'action de radiations électromagnétiques voisines sur les liaisons pile/régulateur. Le régulateur est dit à faible tension de déchet car il permet de fournir un potentiel proche du potentiel d ' alimentation. La figure 1 représente schémati uement un exemple de régulateur à faible tension de déchet classique. Le régulateur comporte une borne de sortie S prévue pour être reliée à une charge R. La charge R, essentiellement résistive, représente la somme des impédances d'entrée des circuits alimentés par le régulateur. Par simplicité, on considère par la suite que la charge R est une résistance. Le régulateur comprend un amplificateur opérationnel 4 dont une entrée non inverseuse IN+ est reliée à un potentiel de référence positif Vref et dont une entrée inverseuse IN~ est reliée à la borne S par une boucle de contre réaction. Le potentiel Vref est produit de manière connue par une source de tension constante (non représentée) à forte impédance de sortie. L'amplificateur opérationnel 4 est alimenté entre un potentiel d'alimentation Vbat positif fourni par la pile et un potentiel de masse GND. Un étage inverseur 6, alimenté entre les potentiels Vbat et GND, reçoit la sortie de l'amplificateur opérationnel 4 et sa sortie est connectée à la grille d'un transistor MOS de puissance Tl, à canal P, dont le drain est relié à la borne de sortie S et dont la source est reliée au potentiel Vbat. Le transistor Tl est de type MOS plutôt que bipolaire notamment pour minimiser la différence entre le potentiel de sortie Vout de la borne S et le potentiel d'alimentation Vbat. Un condensateur de charge C est disposé entre la borne de sortie S et le potentiel GND.A low drop out regulator in the form of an integrated circuit can be used to supply a predetermined potential with low noise to a set of electronic circuits from a supply potential supplied by a rechargeable battery. . Such a supply potential decreases over time and is likely to include noise caused by the action of neighboring electromagnetic radiation on the battery / regulator links. The regulator is said to have a low waste voltage because it provides a potential close to the supply potential. Figure 1 shows schematically an example of a conventional low-voltage waste regulator. The regulator has an output terminal S intended to be connected to a load R. The essentially resistive load R represents the sum of the input impedances of the circuits supplied by the regulator. For simplicity, it is subsequently considered that the load R is a resistance. The regulator comprises an operational amplifier 4 of which a non-inverting input IN + is connected to a positive reference potential Vref and of which an inverting input IN ~ is connected to the terminal S by a feedback loop. The potential Vref is produced in a known manner by a constant voltage source (not shown) with high output impedance. The operational amplifier 4 is supplied between a positive supply potential Vbat supplied by the battery and a ground potential GND. An inverter stage 6, supplied between the potentials Vbat and GND, receives the output of the operational amplifier 4 and its output is connected to the gate of a power MOS transistor T1, with P channel, the drain of which is connected to the output terminal S and whose source is connected to the potential Vbat. The transistor T1 is of the MOS type rather than bipolar in particular to minimize the difference between the output potential Vout of the terminal S and the supply potential Vbat. A charge capacitor C is disposed between the output terminal S and the potential GND.
La figure 2 représente schématiquement un exemple de réalisation de l'amplificateur opérationnel 4 de la figure 1. Deux transistors MOS T2, T3, à canal P, ont leurs • sources reliées l'une à l'autre et leurs grilles respectivement reliées aux entrées IN~ et IN+. Une source de courant de polarisation CS1 est disposée entre le potentiel Vbat et les sources des transistors T2 et T3. Les transistors T2 et T3 forment une paire différentielle. Deux transistors MOS T4 et T5, à canal N, ont leurs sources reliées au potentiel GND et leurs grilles reliées l'une à l'autre. Les drains des transistors T4 et T5 sont respectivement reliés aux drains des transistors T2 et T3. Le drain du transistor T3 est relié aux grilles des transistors T4 et T5. Les transistors T4 et T5 forment une charge active de la paire différentielle formée par les transistors T2 et T3. Le drain du transistor T2 constitue la sortie de l'amplificateur 4.FIG. 2 schematically represents an exemplary embodiment of the operational amplifier 4 of FIG. 1. Two MOS transistors T2, T3, with P channel, have their • sources connected to each other and their gates respectively connected to the inputs IN ~ and IN +. A bias current source CS1 is arranged between the potential Vbat and the sources of the transistors T2 and T3. The transistors T2 and T3 form a differential pair. Two N-channel MOS transistors T4 and T5 have their sources connected to the GND potential and their gates connected to each other. The drains of the transistors T4 and T5 are respectively connected to the drains of the transistors T2 and T3. The drain of transistor T3 is connected to the gates of transistors T4 and T5. The transistors T4 and T5 form an active charge of the differential pair formed by the transistors T2 and T3. The drain of transistor T2 constitutes the output of amplifier 4.
Le régulateur de tension de la figure 1 maintient le potentiel Vout de la borne de sortie S à une valeur égale au potentiel de référence Vref. Toute variation du potentiel Vbat se traduit par une variation du potentiel Vout, qui est transmise par la boucle de contre réaction sur 1 'entrée IN" . Lorsque le régulateur fonctionne correctement, la variation du potentiel de 1 ' entrée IN" entraîne le retour du potentiel Vout au potentiel Vref. Pour cela, le circuit régulateur, qui forme un système bouclé entre 1 'entrée IN" et la borne S doit être un système stable. Pour que le système soit stable lorsqu'il est bouclé, son gain en boucle ouverte ne doit pas être supérieur à 1 lorsque le déphasage est inférieur à -180° (lorsqu'il y a opposition de phase entre l'entrée et la sortie du système) . La figure 3 illustre, en fonction de la fréquence f, la variation du gain G et du déphasage φ du régulateur en boucle ouverte pris entre l'entrée IN" et la borne S. Pour des fréquences f faibles, le gain G est égal au gain Gs statique du régulateur en boucle ouverte. Les éléments qui composent le régulateur ont chacun un gain qui varie en fonction de la fréquence. La fréquence de coupure d'un élément dont le gain décroît lorsque la fréquence augmente constitue un "pôle" de la fonction de transfert du régulateur en boucle ouverte. Chaque pôle de la fonction de transfert du régulateur en boucle ouverte introduit une chute de 20 dB par décade du gain G. En outre, chaque pôle de la fonction de transfert du régulateur en boucle ouverte introduit un déphasage φ de 90°. Par simplicité, on considère par la suite que la fonction de transfert du régulateur en boucle ouverte comprend seulement un pôle principal PO et un pôle secondaire PI. La fréquence à laquelle se situe le pôle principal PO dépend notamment de l'inverse du produit des valeurs de la résistance de charge R et du condensateur C. La fréquence à laquelle se situe le pôle secondaire PI dépend notamment de l'impédance de la grille du transistor Tl. On considère que l'étage inverseur 6 est un étage idéal qui n'introduit aucun pôle. Les caractéristiques des éléments qui composent le régulateur sont choisies de manière que, lorsque le déphasage φ devient égal à -180°, le gain G est inférieur au gain unitaire (0 dB) . En figure 3, le pôle PO est situé à une fréquence peu élevée et le pôle PI est situé à une fréquence supérieure à la fréquence du pôle PO. Pour une fréquence inférieure à la fréquence du pôle PO, le gain est égal au gain statique Gs du régulateur en boucle ouverte. Entre les pôles PO et PI, le gain chute de 20 décibels par décade. Au-delà du pôle PI, le gain chute de 40 décibels par décade. Le déphasage chute de 0 à -90° au niveau du pôle PO et de -90° à - 180° au niveau du pôle PI. Le gain statique Gs du régulateur est égal à Gs4.Gs6.Gsl, où Gs4 est le gain statique de l'amplificateur opérationnel 4, Gs6 est le gain statique de l'étage inverseur 6, et Gsl est le gain statique du transistor Tl. Le gain statique de l'amplificateur opérationnel 4 est de la forme :The voltage regulator of FIG. 1 maintains the potential Vout of the output terminal S at a value equal to the reference potential Vref. Any variation in the potential Vbat results in a variation in the potential Vout, which is transmitted by the feedback loop on the input IN " . When the regulator functions correctly, the variation of the potential of the input IN " causes the potential Vout to return to the potential Vref. For this, the regulator circuit, which forms a looped system between the input IN " and the terminal S must be a stable system. For the system to be stable when it is looped, its gain in open loop must not be greater to 1 when the phase shift is less than -180 ° (when there is phase opposition between the input and the output of the system). Figure 3 illustrates, according to the frequency f, the variation of the gain G and the phase shift φ of the open loop regulator taken between input IN " and terminal S. For low frequencies f, the gain G is equal to the static gain Gs of the regulator in open loop. The elements that make up the regulator each have a gain which varies according to the frequency. The cut-off frequency of an element whose gain decreases when the frequency increases constitutes a "pole" of the transfer function of the regulator in open loop. Each pole of the transfer function of the open loop regulator introduces a fall of 20 dB per decade of gain G. In addition, each pole of the transfer function of the open loop regulator introduces a phase shift φ of 90 °. For simplicity, it is subsequently considered that the transfer function of the open loop regulator comprises only a main pole PO and a secondary pole PI. The frequency at which the main pole PO is located depends in particular on the inverse of the product of the values of the load resistance R and the capacitor C. The frequency at which the secondary pole PI is located depends in particular on the impedance of the grid of the transistor T1. We consider that the inverting stage 6 is an ideal stage which does not introduce any pole. The characteristics of the elements that make up the regulator are chosen so that, when the phase shift φ becomes equal to -180 °, the gain G is less than the unit gain (0 dB). In Figure 3, the PO pole is located at a low frequency and the PI pole is located at a frequency higher than the frequency of the PO pole. For a frequency lower than the frequency of the PO pole, the gain is equal to the static gain Gs of the open loop regulator. Between the PO and PI poles, the gain drops by 20 decibels per decade. Beyond the PI pole, the gain drops by 40 decibels per decade. The phase shift drops from 0 to -90 ° at the PO pole and from -90 ° to - 180 ° at the PI pole. The static gain Gs of the regulator is equal to Gs4.Gs6.Gsl, where Gs4 is the static gain of the operational amplifier 4, Gs6 is the static gain of the inverter stage 6, and Gsl is the static gain of the transistor Tl. The static gain of the operational amplifier 4 is of the form:
GS4 = Grr-2. (R2.R4) / (R2+R4) = (^2 . Zout où .3^2 est la transconductance du transistor T2, et R2, R4 sont les résistances en conduction, dites résistances "Early", des transistors T2 et T4. Le rapport (R2.R4) / (R2+R4) constitue l'impédance de sortie Zout de 1 ' amplificateur opérationnel .GS4 = Grr-2. (R2.R4) / (R2 + R4) = (^ 2. Zout where .3 ^ 2 is the transconductance of the transistor T2, and R2, R4 are the conduction resistors, called "Early" resistors, of the transistors T2 and T4 The ratio (R2.R4) / (R2 + R4) constitutes the output impedance Zout of the operational amplifier.
Les résistances "Early" des transistors T2 et T4 sont élevées, et 1 ' impédance de sortie Zout ainsi que le gain statique Gs4 de l'amplificateur 4 ont une valeur élevée. Un fort gain Gs4 rend le gain statique Gs élevé, ce qui décale la courbe de gain vers le haut et rend la stabilité du régulateur difficile à obtenir.The "Early" resistances of the transistors T2 and T4 are high, and the output impedance Zout as well as the static gain Gs4 of the amplifier 4 have a high value. A strong Gs4 gain makes the static Gs gain high, which shifts the gain curve upwards and makes the stability of the regulator difficult to obtain.
Avec 1 ' amélioration des technologies, les carac- teristiques d'un amplificateur opérationnel s'améliorent et notamment son gain Gs4 tend à augmenter.With the improvement of technologies, the characteristics of an operational amplifier improve and in particular its Gs4 gain tends to increase.
La figure 3 illustre une courbe de gain G' d'un régulateur en boucle ouverte ayant les deux pôles PO, PI précédents et ayant un gain statique Gs' supérieur au gain statique Gs précédent. Le gain G' est supérieur à 1 (0 dB) lorsque le déphasage φ atteint la valeur de -180°, ce qui rend le régulateur instable.FIG. 3 illustrates a gain curve G 'of an open loop regulator having the two poles PO, PI above and having a static gain Gs' greater than the previous static gain Gs. The gain G 'is greater than 1 (0 dB) when the phase shift φ reaches the value of -180 °, which makes the regulator unstable.
Une façon classique de résoudre ce problème consiste à augmenter la capacité du condensateur C, ce qui réduit la fréquence à laquelle se situe le pôle principal PO. Toutefois, l'utilisation d'un condensateur C de grande taille n'est pas souhaitable. De plus, il n'est pas souhaitable de dégrader les caractéristiques des transistors d'un amplificateur opérationnel, étant donné que ces transistors doivent de préférence être identiques aux autres transistors du circuit intégré contenant le régulateur.A conventional way of solving this problem is to increase the capacitance of the capacitor C, which reduces the frequency at which the main pole PO is located. However, the use of a large capacitor C is not desirable. In addition, it is not desirable to degrade the characteristics of the transistors of an operational amplifier, since these transistors must preferably be identical to the other transistors of the integrated circuit containing the regulator.
Un objet de la présente invention est de prévoir un régulateur de tension stable et à grande bande passante tout en utilisant un condensateur de sortie de valeur faible. Pour atteindre cet objet, la présente invention prévoit de réduire la résistance de sortie apparente de l'amplificateur opérationnel d'un régulateur.An object of the present invention is to provide a stable voltage regulator with a large passband while using a low value output capacitor. To achieve this object, the present invention provides for reducing the apparent output resistance of the operational amplifier of a regulator.
Plus particulièrement la présente invention prévoit un régulateur de tension ayant une borne de sortie propre à être reliée à une charge, comprenant un amplificateur opérationnel dont l'entrée non inverseuse est reliée à un premier potentiel de référence, et dont l'entrée inverseuse est reliée à la borne de sortie, un étage inverseur dont l'entrée est reliée à la sortie de l'amplificateur opérationnel, un commutateur de puissance commandé par la sortie de l'étage inverseur, disposé entre la borne de sortie et un potentiel d'alimentation, et un condensateur de charge disposé entre la borne de sortie et un potentiel de référence d'alimentation, comprenant un moyen de réduction de l'impédance de sortie effective de l'amplificateur opérationnel .More particularly, the present invention provides a voltage regulator having an output terminal suitable for being connected to a load, comprising an operational amplifier whose non-inverting input is connected to a first reference potential, and whose inverting input is connected at the output terminal, an inverter stage the input of which is connected to the output of the operational amplifier, a power switch controlled by the output of the inverter stage, disposed between the output terminal and a supply potential , and a charge capacitor disposed between the output terminal and a supply reference potential, comprising means for reducing the effective output impedance of the operational amplifier.
Selon un mode de réalisation de la présente invention, le moyen de réduction d' -umpédance comprend une première résistance dont une première borne est reliée à la sortie de l'ampli icateur opérationnel, un transistor MOS connecté en diode dont le drain est relié à une seconde borne de la première résistance et dont la source est reliée au deuxième potentiel de référence, et un moyen pour polariser le transistor connecté en diode à l'état passant. Selon un mode de réalisation de la présente invention, la première résistance a une valeur très inférieure à l'impédance de sortie de l'amplificateur opérationnel.According to an embodiment of the present invention, the means of reduction of -pumpedance comprises a first resistor, a first terminal of which is connected to the output of the operational amplifier, a MOS transistor connected as a diode, the drain of which is connected to a second terminal of the first resistor and whose source is connected to the second reference potential, and means for biasing the transistor connected as a diode in the on state. According to an embodiment of the present invention, the first resistance has a value much lower than the output impedance of the operational amplifier.
Selon un mode de réalisation de la présente invention, 1 ' amplificateur opérationnel comporte des premier et deuxième transistors MOS, d'un premier type, dont les sources sont reliées l'une à l'autre, et dont les grilles sont respectivement reliées aux entrées inverseuse et non inverseuse, une source de courant disposée entre le potentiel d'alimentation et les sources des premier et deuxième transistors, des troisième et quatrième transistors MOS, d'un second type, dont les sources sont reliées au premier potentiel de référence, dont les grilles sont reliées l'une à l'autre, et dont les drains sont respectivement connectés aux drains des premier et deuxième transistors, le drain du premier transistor étant relié à la sortie de 1 ' amplificateur opérationnel et le drain et la grille du quatrième transistor étant connectés l'un à l'autre.According to an embodiment of the present invention, the operational amplifier comprises first and second MOS transistors, of a first type, the sources of which are connected to each other, and the gates of which are respectively connected to the inputs inverting and non-inverting, a current source disposed between the supply potential and the sources of the first and second transistors, of the third and fourth MOS transistors, of a second type, the sources of which are connected to the first reference potential, of which the gates are connected to each other, and the drains of which are respectively connected to the drains of the first and second transistors, the drain of the first transistor being connected to the output of the operational amplifier and the drain and the gate of the fourth transistor being connected to each other.
Selon un mode de réalisation de la présente invention, l'étage inverseur comporte un cinquième transistor MOS, du type des troisième et quatrième transistors, dont la grille et le drain sont respectivement reliés à l'entrée et à la sortie de l'étage inverseur, et dont la source est reliée au premier potentiel de référence, une impédance disposée entre la sortie de l'étage inverseur et le potentiel d'alimentation, et un condensateur et une seconde résistance disposés en série entre 1 ' entrée et la sortie de 1 'étage inverseur.According to an embodiment of the present invention, the inverter stage comprises a fifth MOS transistor, of the type of the third and fourth transistors, the gate and the drain of which are respectively connected to the input and to the output of the inverter stage , and the source of which is connected to the first reference potential, an impedance arranged between the output of the inverter stage and the supply potential, and a capacitor and a second resistor arranged in series between the input and the output of 1 'reverse stage.
Selon un mode de réalisation de la présente invention, le commutateur de puissance est un sixième transistor MOS du type des premier et deuxième transistors. Selon un mode de réalisation de la présente invention, les premier, deuxième et sixième transistors sont des MOS à canal P, et les troisième, quatrième et cinquième transistors sont des MOS à canal N.According to an embodiment of the present invention, the power switch is a sixth MOS transistor of the type of the first and second transistors. According to an embodiment of the present invention, the first, second and sixth transistors are P-channel MOS, and the third, fourth and fifth transistors are N-channel MOS
Ces objets, caractéristiques et avantages, ainsi que d' autres de la présente invention seront exposés en détail dans la description suivante de modes de réalisation particuliers faite à titre non-limitatif en relation avec les figures jointes parmi lesquelles : la figure 1, précédemment décrite, représente schéma- tiquement un régulateur de tension classique ; la figure 2, précédemment décrite, représente schéma- tiquement un mode de réalisation d'amplificateur opérationnel ; la figure 3, précédemment décrite, illustre le gain et le déphasage en fonction de la fréquence du régulateur de la figure 1 en boucle ouverte ; la figure 4 représente schématiquement un mode de réalisation d'un régulateur selon la présente invention ; et la figure 5 représente schématiquement un mode de réalisation d'un inverseur utilisable selon la présente invention. Seuls les éléments permettant la compréhension de la présente invention ont été représentés aux différentes figures. De mêmes références représentent de mêmes éléments aux différentes figures.These objects, features and advantages, as well as others of the present invention will be explained in detail in the following description of particular embodiments given without limitation in relation to the appended figures among which: FIG. 1, previously described, schematically represents a conventional voltage regulator; FIG. 2, previously described, schematically represents an embodiment of an operational amplifier; FIG. 3, previously described, illustrates the gain and the phase shift as a function of the frequency of the regulator of FIG. 1 in open loop; FIG. 4 schematically represents an embodiment of a regulator according to the present invention; and FIG. 5 schematically represents an embodiment of an inverter usable according to the present invention. Only the elements allowing the understanding of the present invention have been represented in the various figures. The same references represent the same elements in the different figures.
La figure 4 représente schématiquement un mode de réa- lisation d'un régulateur. Le régulateur comprend les éléments déjà décrit d'un régulateur classique et un circuit 7 de réduction d' impédance de sortie connecté à la sortie de 1 ' amplificateur opérationnel 4.FIG. 4 schematically represents an embodiment of a regulator. The regulator comprises the elements already described of a conventional regulator and a circuit 7 for reducing the output impedance connected to the output of the operational amplifier 4.
Une résistance RI a une première borne connectée à la sortie de l'amplificateur opérationnel 4. Un transistor MOS 8, à canal N, a son drain relié à une deuxième borne de la résistance RI et sa source reliée au potentiel GND. Le drain et la grille du transistor 8 sont reliés l'un à l'autre de manière que le transistor 8 est connecté en diode. Une source de courant CS2 de polarisation du transistor 8 connecté en diode est reliée entre le potentiel Vbat et le drain du transistor 8.A resistor RI has a first terminal connected to the output of the operational amplifier 4. An MOS transistor 8, with N channel, has its drain connected to a second terminal of the resistor RI and its source connected to the potential GND. The drain and the gate of transistor 8 are connected to each other so that transistor 8 is connected as a diode. A current source CS2 for biasing the transistor 8 connected as a diode is connected between the potential Vbat and the drain of the transistor 8.
La source de courant CS2 est choisie de manière que le transistor 8 connecté en diode conduit en permanence. Le transistor 8 est choisi de manière que la chute de tension entre son drain et sa source est égale à la tension existant entre l'entrée de l'étage inverseur 6 et le potentiel de masse GND. Il en découle que la chute de tension aux bornes de la résistance RI est sensiblement nulle, et que le fonctionnement de l'amplificateur opérationnel 4 n'est pas déséquilibré par un courant circulant à travers la résistance RI. L'impédance Z du transistor 8 connecté en diode et de la résistance RI connectés en série est égale à : où Gm8 est la transconductance du transistor 8. La résistance RI et le transistor 8 sont choisis de manière que l'impédance Z est très inférieure à 1 ' impédance de sortie Zout de 1 ' amplificateur opérationnel . Le gain statique Gs4 de 1 'amplificateur opérationnel 4 dont la sortie OUT est connectée en parallèle sur l'impédance Z est égal à Gs4 ***** G^. (Zout .Z) / (Zout+Z) , c'est-à- dire sensiblement G^. . La présente invention permet de réduire le gain statique introduit par l'amplificateur opérationnel 4, et par là le gain statique du régulateur de tension en boucle ouverte. Ainsi, la réduction de l'impédance de sortie apparente de 1 ' amplificateur opérationnel 4 correspond à une réduction du gain de cet amplificateur. On pourra ajuster ce gain pour conserver un système stable à grande bande passante, avec un condensateur C de petite valeur.The current source CS2 is chosen so that the transistor 8 connected as a diode conducts permanently. The transistor 8 is chosen so that the voltage drop between its drain and its source is equal to the voltage existing between the input of the inverter stage 6 and the ground potential GND. It follows that the voltage drop across the resistor RI is substantially zero, and that the operation of the operational amplifier 4 is not unbalanced by a current flowing through the resistor RI. The impedance Z of the transistor 8 connected in diode and of the resistance RI connected in series is equal to: where G m 8 is the transconductance of the transistor 8. The resistance RI and the transistor 8 are chosen so that the impedance Z is much less than the output impedance Zout of the operational amplifier. The static gain Gs4 of the operational amplifier 4 whose output OUT is connected in parallel on the impedance Z is equal to Gs4 ***** G ^. (Zout .Z) / (Zout + Z), i.e. substantially G ^. . The present invention makes it possible to reduce the static gain introduced by the operational amplifier 4, and thereby the static gain of the open loop voltage regulator. Thus, the reduction in the apparent output impedance of the operational amplifier 4 corresponds to a reduction in the gain of this amplifier. We can adjust this gain to keep a stable system with high bandwidth, with a capacitor C of small value.
La présente invention a été décrite en relation avec un étage inverseur 6 idéal qui n'introduit aucun pôle dans la fonction de transfert du régulateur de tension en boucle ouverte. En pratique, l'étage inverseur 6 n'est pas un étage amplificateur idéal, mais est par exemple un étage amplificateur dit "de Miller" . Un tel étage amplificateur a notamment pour fonction d'accroître la fréquence à laquelle se situe le pôle secondaire PI afin d'accroître la bande passante du régulateur de tension en boucle ouverte. Un étage de Miller introduit notamment un pôle P2 et un zéro Zl dans la fonction de transfert du régulateur de tension en boucle ouverte.The present invention has been described in relation to an ideal inverter stage 6 which does not introduce any pole in the transfer function of the open loop voltage regulator. In practice, the inverter stage 6 is not an ideal amplifier stage, but is, for example, a so-called "Miller" amplifier stage. The function of such an amplifier stage is notably to increase the frequency at which the secondary pole PI is located in order to increase the bandwidth of the open loop voltage regulator. A Miller stage notably introduces a pole P2 and a zero Zl into the transfer function of the open loop voltage regulator.
La figure 5 représente schématiquement un mode de réa- lisation d'un régulateur de tension selon la présente invention, dans lequel l'étage inverseur 6 du circuit d'amplification 2' est un étage de Miller. L'étage inverseur 6 comporte un transistor T7, à canal N, dont la grille et le drain sont respectivement reliés à l'entrée et à la sortie de l'étage 6. La source du transistor T7 est reliée au potentiel GND. Une impédance 10 est disposée entre la sortie de l'étage 6 et le potentiel Vbat. Un condensateur Cl et une résistance R2 sont disposés en série entre l'entrée et la sortie de l'étage amplificateur. La valeur du condensateur Cl, de la résistance R2, et le gain du transistor T7 permettent notamment d'ajuster les fréquences auxquelles se situent les pôles PI, P2. La chute de tension aux bornes du transistor 8 connecté en diode est dans ce cas choisie égale à la tension grille/source du transistor T7. La réduction de l'impédance de sortie reliée en entrée de l'étage inverseur 6 a également pour effet d'accroître la fréquence à laquelle se situe le pôle P2 introduit par l'étage 6, ce qui représente un avantage supplémentaire de la présente invention.FIG. 5 schematically represents an embodiment of a voltage regulator according to the present invention, in which the inverter stage 6 of the amplification circuit 2 'is a Miller stage. The inverter stage 6 comprises a transistor T7, with an N channel, the gate and the drain of which are respectively connected to the input and to the output of the stage 6. The source of the transistor T7 is connected to the potential GND. An impedance 10 is disposed between the output of stage 6 and the potential Vbat. A capacitor C1 and a resistor R2 are arranged in series between the input and the output of the amplifier stage. The value of the capacitor Cl, of the resistor R2, and the gain of the transistor T7 make it possible in particular to adjust the frequencies at which the poles PI, P2 are located. The voltage drop across the terminals of the transistor 8 connected as a diode is in this case chosen to be equal to the gate / source voltage of the transistor T7. The reduction in the output impedance connected at the input of the inverter stage 6 also has the effect of increasing the frequency at which the pole P2 introduced by stage 6 is situated, which represents an additional advantage of the present invention. .
Bien entendu, la présente invention est susceptible de diverses variantes et modifications qui apparaîtront à 1 'homme de l'art. A titre d'exemple, la présente invention a été décrite en relation avec un amplificateur opérationnel particulier, mais l'homme du métier adaptera sans difficultés la présente invention à un régulateur de tension utilisant d'autres types d'amplificateurs opérationnels.Of course, the present invention is susceptible of various variants and modifications which will appear to one skilled in the art. By way of example, the present invention has been described in relation to a particular operational amplifier, but a person skilled in the art will easily adapt the present invention to a voltage regulator using other types of operational amplifiers.
La présente invention a été décrite en relation avec un régulateur de tension utilisant un transistor de puissance Tl, mais l'homme du métier adaptera sans difficultés la présente invention à un régulateur de tension utilisant un autre type de commutateur de puissance à commande en tension.The present invention has been described in relation to a voltage regulator using a power transistor T1, but a person skilled in the art will easily adapt the present invention to a voltage regulator using another type of voltage-controlled power switch.
La présente invention a été décrite en relation avec des potentiels Vbat et Vref positifs, mais l'homme du métier adaptera sans difficultés la présente invention à des potentiels Vbat et Vref négatifs, en intervertissant les types des transis- tors MOS décrits et le branchement du transistor 8 connecté en diode.The present invention has been described in relation to positive Vbat and Vref potentials, but those skilled in the art will easily adapt the present invention to negative Vbat and Vref potentials, by reversing the types of transistors. tors MOS described and the connection of the transistor 8 connected in diode.
La présente invention a pour des raisons de simplicité été décrite en relation avec une charge résistive R, mais l'homme du métier adaptera sans difficultés la présente invention à une charge complexe.The present invention has been described for simplicity reasons in relation to a resistive load R, but a person skilled in the art will easily adapt the present invention to a complex load.
La présente invention a pour des raisons de simplicité été décrite en relation avec un régulateur de tension utilisant une boucle de contre-réaction non résistive et fournissant une tension égale à une tension de référence Vref reçue. Toutefois, l'homme du métier adaptera sans difficultés la présente invention à un régulateur de tension dont la boucle de contre- j réaction comprend un pont résistif, et qui fournit en sortie une tension différente de la tension Vref reçue. The present invention has for reasons of simplicity been described in relation to a voltage regulator using a non-resistive feedback loop and supplying a voltage equal to a reference voltage Vref received. However, a person skilled in the art will easily adapt the present invention to a voltage regulator, the feedback loop of which comprises a resistive bridge, and which supplies an output voltage different from the voltage Vref received.

Claims

REVENDICATIONS
1. Régulateur de tension ayant une borne de sortie (S) propre à être reliée à une charge (R) , comprenant : un amplificateur opérationnel (4) dont l'entrée non inverseuse est reliée à un premier potentiel de référence (Vref), et dont l'entrée inverseuse est reliée à la borne de sortie (S) , un étage inverseur (6) dont l'entrée est reliée à la sortie de l'amplificateur opérationnel, un commutateur de puissance (Tl) commandé par la sortie de l'étage inverseur (6), disposé entre la borne de sortie (S) et un potentiel d'alimentation (Vbat) , et un condensateur de charge (C) disposé entre la borne de sortie (S) et un potentiel de référence d'alimentation (GND), caractérisé en ce qu'il comprend un moyen (7) de réduction de l'impédance de sortie effective de l'amplificateur opérationnel (4) .1. Voltage regulator having an output terminal (S) capable of being connected to a load (R), comprising: an operational amplifier (4) whose non-inverting input is connected to a first reference potential (Vref), and whose inverting input is connected to the output terminal (S), an inverting stage (6) whose input is connected to the output of the operational amplifier, a power switch (Tl) controlled by the output of the inverter stage (6), disposed between the output terminal (S) and a supply potential (Vbat), and a charge capacitor (C) disposed between the output terminal (S) and a reference potential d power supply (GND), characterized in that it comprises means (7) for reducing the effective output impedance of the operational amplifier (4).
2. Régulateur de tension selon la revendication 1, caractérisé en ce que le moyen de réduction d'impédance comprend une première résistance (RI) dont une première borne est reliée à la sortie de l'amplificateur opérationnel (4), un transistor2. Voltage regulator according to claim 1, characterized in that the impedance reduction means comprises a first resistor (RI) of which a first terminal is connected to the output of the operational amplifier (4), a transistor
MOS connecté en diode (8) dont le drain est relié à une seconde borne de la première résistance (RI) et dont la source est reliée au deuxième potentiel de référence (GND) , et un moyenMOS connected as a diode (8) whose drain is connected to a second terminal of the first resistor (RI) and whose source is connected to the second reference potential (GND), and a means
(CS2) pour polariser le transistor connecté en diode (8) à l'état passant.(CS2) to bias the transistor connected as a diode (8) to the on state.
3. Régulateur de tension selon la revendication 2, dans lequel la première résistance (RI) a une valeur très inférieure à l'impédance de sortie de l'amplificateur opérationnel (4) . . Régulateur de tension selon la revendication 3 , dans lequel l'ampli icateur opérationnel (4) comporte : des premier (T2) et deuxième (T3) transistors MOS, d'un premier type, dont les sources sont reliées l'une à l'autre, et dont les grilles sont respectivement reliées aux entrées inverseuse et non inverseuse, une source de courant (CS1) disposée entre le potentiel d'alimentation (Vbat) et les sources des premier et deuxième transistors, des troisième 3. Voltage regulator according to claim 2, wherein the first resistance (RI) has a value much lower than the output impedance of the operational amplifier (4). . Voltage regulator according to claim 3, in which the operational amplifier (4) comprises: first (T2) and second (T3) MOS transistors, of a first type, the sources of which are connected one to the other, and whose gates are respectively connected to the inverting and non-inverting inputs, a current source (CS1) arranged between the supply potential (Vbat) and the sources of the first and second transistors, of the third
(T4) et quatrième (T5) transistors MOS, d'un second type, dont les sources sont reliées au premier potentiel de référence (GND) , dont les grilles sont reliées l'une à l'autre, et dont les drains sont respectivement connectés aux drains, des premier (T2) et deuxième (T3) transistors, le drain du premier transistor (T2) étant relié à la sortie de l'amplificateur opérationnel et le drain et la grille du quatrième transistor (T5) étant connectés l'un à 1 ' autre. (T4) and fourth (T5) MOS transistors, of a second type, whose sources are connected to the first reference potential (GND), whose gates are connected to each other, and whose drains are respectively connected to the drains, of the first (T2) and second (T3) transistors, the drain of the first transistor (T2) being connected to the output of the operational amplifier and the drain and the gate of the fourth transistor (T5) being connected to the one to another.
5. Régulateur de tension selon la revendication 4, dans lequel l'étage inverseur (6) comporte :5. Voltage regulator according to claim 4, in which the inverter stage (6) comprises:
.un cinquième transistor MOS (T7) , du type des troisième (T4) et quatrième (T5) transistors, dont la grille et le drain sont respectivement reliés à 1 'entrée et à la sortie de l'étage inverseur (6), et dont la source est reliée au premier potentiel de référence (GND) , une impédance (10) disposée entre la sortie de l'étage inverseur (6) et le potentiel d'alimentation (Vbat), et un condensateur (Cl) et une seconde résistance (R2) disposés en série entre l'entrée et la sortie de l'étage inverseur (6) ..a fifth MOS transistor (T7), of the type of the third (T4) and fourth (T5) transistors, the gate and the drain of which are respectively connected to the input and the output of the inverter stage (6), and whose source is connected to the first reference potential (GND), an impedance (10) arranged between the output of the inverter stage (6) and the supply potential (Vbat), and a capacitor (Cl) and a second resistance (R2) arranged in series between the input and output of the inverter stage (6).
6. Régulateur de tension selon la revendication 4, dans lequel le commutateur de puissance (Tl) est un sixième transistor MOS du type des premier (T2) et deuxième (T3) transistors .6. Voltage regulator according to claim 4, in which the power switch (Tl) is a sixth MOS transistor of the type of the first (T2) and second (T3) transistors.
7. Régulateur de tension selon la revendication 5, dans lequel les premier (T2) , deuxième (T3) et sixième (Tl) transistors sont des MOS à canal P, et dans lequel les troisième7. Voltage regulator according to claim 5, in which the first (T2), second (T3) and sixth (Tl) transistors are P-channel MOS, and in which the third
(T4) , quatrième (T5) et cinquième (T7) transistors sont des MOS à canal N. (T4), fourth (T5) and fifth (T7) transistors are N-channel MOS.
EP01995774A 2000-12-22 2001-12-21 Voltage regulator with static gain in reduced open loop Withdrawn EP1352302A1 (en)

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FR0016978 2000-12-22
FR0016978A FR2818762B1 (en) 2000-12-22 2000-12-22 REDUCED OPEN LOOP STATIC GAIN VOLTAGE REGULATOR
PCT/FR2001/004174 WO2002052364A1 (en) 2000-12-22 2001-12-21 Voltage regulator with static gain in reduced open loop

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FR2818762B1 (en) 2003-04-04
WO2002052364A1 (en) 2002-07-04
FR2818762A1 (en) 2002-06-28
US6933708B2 (en) 2005-08-23
US20040061485A1 (en) 2004-04-01

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