EP1246294A2 - Active broadband receiving antenna - Google Patents

Abstract

A frequency-dependent effective length has output connectors linked to input connectors on an amplifier circuit (21) that has a field effect transistor (2) and a low-loss filter circuit (3) with an input admittance (7). At its input the filter circuit connects to a source connection for the field effect transistor. At its output (4) an HF receiving signal (8) is decoupled and the low-loss filter circuit is charged with an effective resistance acting at its output.

Description

The invention relates to an active broadband reception antenna, consisting of a passive antenna part 1 with a frequency-dependent effective length l _e , the output connections of which are connected to the input connections of an amplifier circuit 21. Electrically long antennas or antennas that are in direct coupling with large electrical bodies have a frequency-dependent open circuit voltage when excited with an electrical field strength that is kept constant over frequency, which is expressed by the effective length l _e (f). In the frequency range above 30 MHz in particular, the antenna noise temperature T _A in terrestrial surroundings - coming from low frequencies - has dropped to such an extent that a source impedance near the optimum impedance Z _opt for the transistor must be required for the passive antenna part for bipolar transistors, in order not to suffer a significant loss of sensitivity due to the transistor noise. The basic form of an active antenna of this type is shown in Fig. 2a and is known for example from the DT-AS 23 10 616, the DT-AS xx xx 300. With active broadband antennas, which are not channel-selective, but on a frequency band, such as VHF radio frequency range are broadband tuned, it is necessary to transform the antenna impedance Z _s (f) of a short radiator in Z _A (f) in the vicinity of Z _opt . This leads to a frequency-dependent open circuit voltage at the transistor input, both in the case of electrically large and also electrically small antennas, which is expressed as a strongly frequency-dependent effective length l _e (f) of the passive antenna part, from which, in conjunction with the frequency dependence of the voltage division factor, between Z _opt and the Deviating input resistance of the transistor results in the need to smooth the resulting frequency response of the received signal at the load resistor Z _L with the help of an adaptation circuit at the output of the active circuit. This is also necessary to protect the subsequent receiving system against non-linear effects due to level overload.

The basic form of an active antenna of this type is shown in FIG. 2a and is known, for example, from DT-AS 23 10 616, DT-AS xx xx 300. Active antennas according to this prior art are used, for example, to a large extent above the high-frequency range Antenna arrangements are installed in a motor vehicle window pane together with a heating field for the pane heating, as described, for example, in EP 0 396 033, EP 0 346 591 and in EP 0 269 723. The structures of the heating fields used as passive antenna part 1 are vehicle parts which were not originally intended for use as an antenna and which can only be changed slightly due to their function for heating. If an active antenna according to the state of the art is implemented on such an antenna element as in FIG. 2a, the impedance present at the heating field is to be transformed into the vicinity of the impedance Z _opt for noise adaptation with the aid of a primary matching circuit and the frequency response of the active antenna is to be transformed Smooth with the help of an output-side adaptation network. This procedure necessitates the relatively cumbersome dimensioning of two filter circuits, which for an advantageous overall behavior of the active antenna cannot take place separately for each filter due to the mutual dependence on one another. In addition, the amplifier circuit for achieving sufficient linearity properties cannot be designed as a simple amplifying element as in FIG. 2a, as a result of which the creative freedom of the two matching networks is significantly restricted. In addition, the design of two filters involves increased effort. Another notable disadvantage of an active antenna of this type is the load on the matching circuit connected to the heating field with a downstream amplifier if several active antennas are designed from the same heating field to form an antenna diversity system or a group antenna with special directional properties or other purposes. This disadvantageous situation exists in all antenna arrangements, the passive antenna parts of which are in appreciable electromagnetic radiation coupling to one another. For example, according to the prior art, in a multi-antenna scanning diversity system formed from the heating field, switching diodes are attached to the connection points for the antenna amplifiers formed on the heating field, which in each case only switch on the matching circuit with amplifier whose signal is switched through to the receiver and which the other connection points unlock. In such systems, this leads to a considerable outlay and to the additional requirement of switching the diodes precisely in synchronism with the antenna selection.

The object of the invention is therefore to provide an active broadband receiving antenna To design the preamble of claim 1 so that under a given passive antenna part Ensuring high noise sensitivity is largely independent of the Frequency dependence of the effective length and the impedance of the passive antenna part freely selectable frequency dependence of the received power is achieved and that for Multi-antenna arrangements the multiple decoupling of received signals from a passive Antenna arrangement with several connection points, which in electromagnetic Radiation coupling are not noticeable due to the formation of the active antennas mutual influence of the received signals is given.

According to the invention, this object is achieved by the characterizing features of claim 1 solved.

The advantages that can be achieved with the invention consist in particular in the reduction of the Effort and simplicity to achieve a signal to noise ratio and optimal reception signal with regard to the risk of nonlinear effects. by virtue of the elimination of a primary matching network in connection with the input side High impedance of the amplifier circuit results in an extremely advantageous freedom in the Design of complex multi-antenna systems, the passive antenna parts in Radiation coupling to each other. Those related to the diversity arrangement above mentioned switching diodes for the activation of connection points, at each of which no signal for Switching to the receiver is therefore advantageously not necessary.

Fig. 1:

Aktive Breitbandempfangsantenne nach der Erfindung mit einer direkt an den passiven Antennenteil 1 angeschlossenen Verstärkerschaltung 21 mit einem Feldeffekttransistor 2, mit in der Sourceleitung befindlicher Eingangsadmittanz 7 der verlustarmen Filterschaltung 3 und ausgangsseitig wirksamem Wirkwiderstand 5.

Fig. 2:

a) Elektrisches Ersatzschaltbild einer aktiven Breitbandempfangsantenne nach der Erfindung mit Serienrauschspannungsquelle u_r und in ihrer Wirkung vernachlässigbarer Parallelrauschstromquelle i_r des Feldeffekttransistors 2 mit einer außerhalb des Übertragungsbereichs eingangsseitig hochohmigen verlustarmen Filterschaltung 3.

b) Elektrisches Ersatzschaltbild einer aktiven Breitbandempfangsantenne nach dem Stand der Technik mit Rauschanpassungsnetzwerk und frequenzabhängiger effektiver Länge des passiven Antennenteils 1 am Anschlusspunkt des Transistors und ausgangsseitigem Anpassungsnetzwerk zur Glättung des Frequenzgangs.

Fig. 3

a) Elektrisches Ersatzschaltbild wie in Fig. 2a, jedoch mit ausgangsseitigem Abschluß der verlustarmen Filterschaltung 3 mit einer Hochfrequenzleitung 10 und einer daran angeschlossenen Verstärkereinheit 11 mit Beitrag des Eigenrauschens der Verstärkereinheit 11 zum Gesamtrauschen.

b) Elektrisches Ersatzschaltbild wie in Fig. 3a mit einer Verstärkereinheit 11 am Ausgang der verlustarmen Filterschaltung 3 mit einer Hochfrequenzleitung 10 und einer weiterführenden Verstärkerschaltung.

Fig. 4
Gestaltung eines erweiterten Feldeffekttransistors 2 mit Hilfe eines Eingangs-Feldeffekttransistors 13 und eines durch die Source angesteuerten Bipolartransistors 14 in Emitterfolgerschaltung

Fig: 5
Beispiel einer aktiven Breitbandempfangsantenne nach der Erfindung mit einem miniaturisiert ausgeführten Frontend der aktiven Antenne, einer Hochfrequenzleitung 10 und einer ergänzenden Filterschaltung 3 zur Anbringung auf der Heckfensterscheibe eines Kraftfahrzeugs.

Fig. 6

a) Verlauf der seriellen Blindwiderstände X₁ und X₃ sowie des parallelen Blindleitwerts B₂ der erfindungsgemäßen T-Filteranordnung in Fig. 6b über der Frequenz am Beispiel der breitbandigen Abdeckung der Rundfunkbereiche UKW-Hörrundfunk sowie VHF- und UHF-Fernsehrundfunk.

b) Elektrisches Ersatzschaltbild einer Antenne nach der Erfindung für die unter a) genannten Frequenzbereiche.

Fig. 7
Aktive Antenne nach der Erfindung mit zwei Übertragungswegen für unterschiedliche Übertragungsfrequenzbereiche und Aufspaltung der Signalwege am Ausgang des Eingangs-Feldeffekttransistors 13 mit jeweils einem Bipolartransistor 14 und nachgeschalteter verlustarmer Filterschaltung 3 für den betreffenden Übertragungsfrequenzbereich und Zusammenschaltung der Ausgangssignale am gemeinsam wirksamen Wirkwiderstand 5 .

Fig. 8
Aktive Antenne nach der Erfindung mit einem weiteren Feldeffekttransistor 17 zur Kompensation von Effekten der Nichtlinearität geradzahliger Ordnung und einem ausgangsseitigen Umsymmetrierglied 20.

Fig. 9
Aktive Antenne wie in Fig. 8, jedoch mit einer Signalverzweigung entsprechend der Antenne in Fig. 7 mit jeweils einem Umsymmetrierglied 20 am Eingang der verlustarmen Filterschaltung 3.

Fig. 10
Aktive Antenne nach der Erfindung mit einem Übertrager 24 zur Schaffung günstiger Übertragungsverhältnisse und einem Linearisierungswiderstand 30 zur weiteren Erhöhung der Linearität.

Fig. 11
Aktive Antenne nach der Erfindung, jedoch mit einem Übertrager 24 mit hinreichend hochohmiger Primärinduktivität und hinreichend großem Übersetzungsverhältnis zur breitbandigen Erhöhung der effektiven Länge l_e.

Fig. 12
Aktive Antenne nach der Erfindung mit einer in der verlustarmen Filterschaltung 3 gestalteten frequenzselektiven Signalverzweigung.

Fig. 13
Gruppenantenne zur Gestaltung von Richtwirkungen mit einer passiven Antennenanordnung 27 mit elektrischer Strahlungskopplung zwischen den Anschlussstellen 18, welche jeweils mit einer Verstärkerschaltung 21 und einer Hochfrequenzleitung 10 beschaltet sind und deren Signale im Antennencombiner 22 zusammengefasst sind.

Fig. 14
Scanningdiversity-Antennenanlage mit einer Anordnung wie in Fig. 13, jedoch mit elektronischen Umschalter 25 an Stelle des Antennencombiners 22 und jeweils einem Ersatzlastwiderstand 26 zur Belastung der nicht durchgeschalteten Antennenzweige.

Fig. 15

a) Scanningdiversity-Antennenanlage gebildet aus auf die Fensterscheibe gedruckten Heizfeldern mit diversitätsmäßig geeignet positionierten Anschlussstellen 18 zur Erreichung diversitätsmäßig unabhängiger Empfangssignale 8.

b) Scanningdiversity-Antennenanlage wie unter a), jedoch mit einer auf die Fensterscheibe angebrachten leitenden Fläche mit hinreichend kleinem Oberflächenwiderstand und Gestaltung von Anschlussstellen 18 mit Hilfe galvanisch oder kapazitiv angekoppelter Sammelelektroden.

Fig. 16
Scanningdiversity-Antennenanlage gebildet aus auf die Fensterscheibe gedruckten Heizfeldern mit diversitätsmäßig geeigneten Anschlussstellen 18 und gesondert ermittelten Blindleitwerten 23 zur Erhöhung der diversitätsmäßigen Unabhängigkeit der Empfangssignale.

Fig. 17
Passiver Antennenteil 1 mit einer Anschlussstelle 18, deren beide Anschlüsse gegenüber dem Masseanschluß hochliegen, mit einem Feldeffekttransistor 2 und einem weiteren Feldeffekttransistor 17 und einer massesymmetrisch ausgeführten verlustarmen Filterschaltung 3 und einem ausgangsseitigen Umsymmetrierglied 20 zur Erzeugung unsymmetrisch vorliegender Empfangssignale am wirksamen Wirkwiderstand 5.

Fig. 18

a) und b): Beispielhafte Antennenkonfigurationen möglicher passiver Antennenteile 1

c) Impedanzverläufe der Antennenstrukturen A1, A2 und A3 in der Imepanzebene im Frequenzbereich von 76 bis 108 MHz und schraffierte Bereiche für R_A < R_Amin und R_A > R_Amax

d) Realteile der Antennenimpedanzen nach c) mit zulässigem Wertebereich R_Amin < R_A < R_Amax

Exemplary embodiments of active broadband reception antennas and antenna systems according to the invention are shown in the drawings and are described in more detail below. In detail shows:

Fig. 1:

Active broadband receiving antenna according to the invention with an amplifier circuit 21 connected directly to the passive antenna part 1 with a field effect transistor 2, with input admittance 7 in the source line of the low-loss filter circuit 3 and effective resistor 5 on the output side.

Fig. 2:

a) Electrical equivalent circuit diagram of an active broadband receiving antenna according to the invention with series noise voltage source u _r and the effect of negligible parallel noise current source i _r of field effect transistor 2 with a low-loss filter circuit 3 on the input side outside the transmission range.

b) Electrical equivalent circuit diagram of an active broadband reception antenna according to the prior art with noise adaptation network and frequency-dependent effective length of the passive antenna part 1 at the connection point of the transistor and output-side adaptation network for smoothing the frequency response.

Fig. 3

a) Electrical equivalent circuit as in Fig. 2a, but with the output-side termination of the low-loss filter circuit 3 with a high-frequency line 10 and an amplifier unit 11 connected to it with the contribution of the inherent noise of the amplifier unit 11 to the total noise.

b) Electrical equivalent circuit diagram as in FIG. 3a with an amplifier unit 11 at the output of the low-loss filter circuit 3 with a high-frequency line 10 and a further amplifier circuit.

Fig. 4
Design of an extended field effect transistor 2 with the aid of an input field effect transistor 13 and a bipolar transistor 14 driven by the source in an emitter follower circuit

Fig: 5
Example of an active broadband reception antenna according to the invention with a miniaturized front end of the active antenna, a high-frequency line 10 and a supplementary filter circuit 3 for attachment to the rear window of a motor vehicle.

Fig. 6

a) Course of the serial reactances X ₁ and X ₃ and the parallel reactive conductance B _{2 of} the T-filter arrangement according to the invention in Fig. 6b over the frequency using the example of the broadband coverage of the radio areas FM radio broadcasting and VHF and UHF television broadcasting.

b) Electrical equivalent circuit diagram of an antenna according to the invention for the frequency ranges mentioned under a).

Fig. 7
Active antenna according to the invention with two transmission paths for different transmission frequency ranges and splitting of the signal paths at the output of the input field-effect transistor 13, each with a bipolar transistor 14 and downstream low-loss filter circuit 3 for the relevant transmission frequency range and interconnection of the output signals at the common effective resistor 5.

Fig. 8
Active antenna according to the invention with a further field effect transistor 17 for compensating effects of the non-linearity of even order and an output-side balancing element 20.

Fig. 9
Active antenna as in FIG. 8, but with a signal branching corresponding to the antenna in FIG. 7, each with a balancing element 20 at the input of the low-loss filter circuit 3.

Fig. 10
Active antenna according to the invention with a transmitter 24 to create favorable transmission conditions and a linearization resistor 30 to further increase the linearity.

Fig. 11
Active antenna according to the invention, but with a transformer 24 with a sufficiently high-impedance primary inductance and a sufficiently large transmission ratio for broadband increase in the effective length l _e .

Fig. 12
Active antenna according to the invention with a frequency-selective signal branching designed in the low-loss filter circuit 3.

Fig. 13
Group antenna for designing directional effects with a passive antenna arrangement 27 with electrical radiation coupling between the connection points 18, which are each connected to an amplifier circuit 21 and a high-frequency line 10 and whose signals are combined in the antenna combiner 22.

Fig. 14
Scanning diversity antenna system with an arrangement as in FIG. 13, but with an electronic switch 25 instead of the antenna combiner 22 and in each case an equivalent load resistor 26 for loading the antenna branches which are not switched through.

Fig. 15

a) Scanning diversity antenna system formed from heating fields printed on the window pane with connection points 18 suitably positioned in terms of diversity in order to achieve reception signals 8 that are independent of diversity.

b) Scanning diversity antenna system as under a), but with a conductive surface attached to the window pane with a sufficiently small surface resistance and design of connection points 18 with the aid of galvanically or capacitively coupled collecting electrodes.

Fig. 16
Scanning diversity antenna system formed from heating fields printed on the window pane with connection points 18 suitable for diversity and separately determined blind conductance values 23 for increasing the diversity-independent of the received signals.

Fig. 17
Passive antenna part 1 with a connection point 18, the two connections of which are high in relation to the ground connection, with a field effect transistor 2 and a further field effect transistor 17 and a low-loss filter circuit 3 of symmetrical design and an output-side resymmetry element 20 for generating asymmetrical reception signals at the effective resistance 5.

Fig. 18

a) and b): Exemplary antenna configurations of possible passive antenna parts 1

c) Impedance profiles of the antenna structures A1, A2 and A3 in the image plane in the frequency range from 76 to 108 MHz and hatched areas for R _A <R _Amin and R _A > R _Amax

d) Real parts of the antenna impedances according to c) with permissible value range R _Amin <R _A <R _Amax

In Fig. 1 an antenna according to the basic form of the invention is shown. Using the example of the heating field of a motor vehicle printed on a window pane, it can be seen that the passive antenna part 1 cannot be designed in such a way that it has particular desired properties with regard to its use as an antenna in the meter and decimeter wave range and thus has a shape corresponding to its geometric structure and the metallic border of the window has a random frequency dependence of both the effective length l _e and its impedance. The essence of the present invention now consists in realizing an active antenna which allows this randomness of the frequency dependency of the given passive antenna part 1 to be compensated for with the aid of an active antenna which is not complex, easy to determine and easy to implement, and with regard to self-noise, linearity and Freely design frequency response and to achieve a predetermined frequency response between the incident wave with the electric field strength E and the high-frequency received signal 8. According to the invention, the reception voltage present at a connection point 18 is fed to the amplifier circuit 21, which is made up of a field effect transistor 2, which is coupled in its source line to the input admittance 7 of a low-loss filter circuit 3, which is terminated at its output with an effective resistance 5. In the case of an antenna of this type, the input admittance 7 is to be designed according to the invention, for example, in such a way that the strong frequency dependence, which the received open circuit voltage, expressed by the effective length l _{e of} the passive antenna part 1 designed in this way, is largely balanced in the high-frequency received signal 8.

Die Eignung eines vorgegebenen passiven Antennenteils 1 für die Gestaltung einer hinreichend rauschempfindlichen aktiven Antenne kann anhand der im Übertragungsfrequenzbereich herrschenden Antennentemperatur abgeschätzt werden. Feldeffekttransistoren besitzen in der Regel eine extrem kleine Parallelrauschstromquelle i_r, so dass deren Beitrag i_r*Z_A bei vernachlassigbar kleinen Gate-Source- und Gate-Drain-Kapazitäten C₂ und C₁ und den in der Praxis auftretenden Antennenimpedanzen Z_A im Vergleich zur Serienrauschspannungsquelle u_r des Feldeffekttransistors, ausgedrückt durch seinen äquivalenten Rauschwiderstand R_äF , stets vernachlässigbar klein ist. Die Empfindlichkeitsforderung reduziert sich somit darauf, dass die Rauschspannungsquelle u_r ² = 4kT_oBR_äF im Verhältnis zur empfangenen Rauschspannungsquelle u_rA ² = 4kT_AB R_A, welche durch die Antennentemperatur T_A und dem Realteil R_A der Antennenimpedanz Z_A gegeben ist, kleiner oder höchstens gleich groß ist. Bei gleich großen Rauschbeiträgen ist somit als hinreichendes Empfindlichkeitskriterium bei vernachlassigbar kleinen Kapazitäten C₁, C₂ lediglich die einfach zu prüfende Forderung RA > RäF *T0/TA zu erfüllen. Moderne Gallium-Arsenid-Transistoren besitzen im Vergleich zur übrigen Beschaltung vernachlässigbare Kapazitäten C₁ und C₂ und eine im Hinblick auf die vorgesehene Anwendung vernachlässigbare Wirkung von i_r als Ursache für die bei Rauschanpassung solcher Transistoren extrem kleinen Rauschtemperatur T_N0. Der äquivalente Rauschwiderstand ist vom Ruhestrom abhängig und kann oberhalb 30 MHz breitbandig mit 30 Ohm und weniger angesetzt werden. Für das Beispiel einer Antenne für den UKW-Frequenzbereich und einer dort vorherrschende Antennentemperatur von ca. 1000 K ist somit im Hinblick auf die Rauschempfindlichkeit für den Realteil der komplexen Antennenimpedanz, welcher bei verlustarmem Feldeffekttransistor 2 den Strahlungswiderstand darstellt, innerhalb des Übertragungsfrequenzbereichs ausschließlich R_A(f) > ca. 10 Ohm als hinreichende Bedingung zu fordern.

The mode of operation and the design principle of an antenna according to the invention are explained using the electrical equivalent circuit diagrams of FIGS. 2a and 3a:

The suitability of a given passive antenna part 1 for the design of a sufficiently noise-sensitive active antenna can be estimated on the basis of the antenna temperature prevailing in the transmission frequency range. Field effect transistors have i _r usually an extremely small parallel noise current source so that their contribution i _r * Z _A at negligibly small gate-source and gate-drain capacitances C ₂ and C ₁ and the antenna impedances occurring in practice Z _A in comparison to the series noise _{voltage source} u _{r of} the field effect transistor, expressed by its equivalent noise _resistance R _äF , is always negligibly small. The sensitivity requirement is thus reduced to the fact that the noise voltage source u _r ² = 4kT _o BR _af in relation to the received noise voltage source u _rA ² = 4kT _A BR _A , which is given by the antenna temperature T _A and the real part R _{A of} the antenna impedance Z _A , is smaller or at most the same size. With noise contributions of the same size, the sufficient sensitivity criterion with negligibly small capacitances C ₁ , C ₂ is therefore simply the requirement that is easy to test R A > R AEF * T 0 / T A to fulfill. Modern gallium arsenide transistors have negligible capacitances C ₁ and C ₂ compared to the rest of the circuitry, and an effect of i _{r that is} negligible with regard to the intended application as the cause of the extremely low noise temperature T _N0 when such transistors are adapted to noise. The equivalent noise resistance depends on the quiescent current and can be applied over 30 MHz broadband with 30 ohms and less. For the example of an antenna for the FM frequency range and an prevailing antenna temperature of approx. 1000 K, with regard to the noise sensitivity for the real part of the complex antenna impedance, which represents the radiation resistance in the case of low-loss field-effect transistor 2, only R _A ( f)> about 10 ohms as a sufficient condition.

3a shows the noise contribution of an amplifier unit 11 at the end of the high-frequency line 10 connected to the low-loss filter circuit 3 on the output side. With sufficient amplification in the amplifier circuit 21, this contribution is kept correspondingly small. To protect the downstream amplifier unit 11 from non-linear effects, it is often necessary to make this amplification largely independent of frequency within the transmission frequency range. This is achieved by appropriate, preferably lossless, transformation of the effective resistance 5 at the output of the low-loss filter circuit 3 into a suitable frequency-dependent input admittance 7. If the frequency dependency required for the input admittance 7 due to the frequency dependency of the effective length l _e (f) is known, then a circuit of reactive resistors for the low-loss filter circuit 3 can be found which largely corresponds to this requirement.

The criterion according to the invention for the exemplary design of a necessary and frequency-independent reception power within the transmission frequency range is explained for terrestrial broadcast reception of an active vehicle antenna with regard to the reception power in the downstream reception arrangement with reference to FIG. 3a. The largely frequency-independent reception behavior must be demanded in order not to significantly reduce the sensitivity of the overall system due to the noise contribution of the reception system downstream of the active antenna and on the other hand to avoid non-linear effects due to amplification increases as a result of the frequency-dependent reception behavior within a transmission frequency range. The reception system connected downstream of the active antenna is represented by the amplifier unit 11 with the noise figure F _v . Its noise contribution to the total noise is shown in FIG. 3b as an equivalent noise resistance Räv at the input of the amplifier circuit 21, where: R AEV = ( F V - 1) 4 · G ( f )

Herein, G (f) denotes the frequency-dependent real part of the input admittance 7 of the low-loss filter circuit 3. This noise contribution is insignificant then applies against the inevitable received sound of the rushing with T _A R _A when: G ( f ) ≻ ( F V - 1)· T 0 4 · T A · 1 R A ( f )

In order to meet the sensitivity condition, in an advantageous embodiment of an active antenna according to the invention, the frequency dependence of the real part G (f) of the input admittance 7 of the low-loss filter circuit 3 is to be selected reciprocally to the frequency response of the real part R _A (f) of the complex antenna impedance. For the example of an FM radio receiver with F _v ∼ 4, approximately G (f) <1 / (3 * R _A (f)) would therefore have to be selected. To protect the receiver from reception levels that are too high, on the other hand, it is expedient not to select the power gain of the active antenna to be substantially greater than for optimum sensitivity of the overall system and thus to make G (f) approximately as large as indicated in the right part of equation (3).

The invention is associated with the great advantage that the frequency response for G (f) specified from R _A (f) can be easily fulfilled because neither the source impedance of the low-loss filter circuit 3 which drives the input side, which is given with 1 / g _{m of} the field effect transistor 2 is, the effective resistance 5 at the output of the low-loss filter circuit 3 still have unavoidable essential reactive components. This results in the advantageously free design of the frequency response of the active antenna according to the present invention. In contrast to this, in the case of an active antenna according to the prior art in FIG. 2b, the frequency-dependent radiator impedance Z _s (f) is inevitable and inseparable as the source impedance of the primary-side transformation network. Their frequency behavior limits the achievable bandwidth of the impedance transformed into the vicinity of Z _opt and thus the bandwidth of the signal-to-noise ratio at the output of the active circuit.

bzw. für die Referenzantenne bei der Bezugsfrequenz:

The exemplary design of the frequency response of G (f) of an active vehicle antenna according to the invention is described below if there is a requirement that the received power P _a at the input of the receiving system connected downstream of the active antenna is greater by a factor V than with a passive reference antenna , for example a passive rod antenna on the vehicle with its resonance length. Due to the necessarily different directional diagrams, this factor is based on the azimuthal mean values at a defined constant elevation angle  of the wave incidence. Comparative azimuthal directional factor measurements with the aid of an antenna measuring section with vehicle rotation at the passive antenna part 1 and on the comparison antenna result in N angular steps for a full revolution and with the directional factor D _a ( _n , ) of the predetermined passive antenna part 1 and corresponding to the directional factor D _p ( _n , ) of the passive reference antenna for the nth angular step the following azimuthal mean values for the guide factors:

or for the reference antenna at the reference frequency:

The receiving system downstream of the active antenna, which is represented in FIG. 3a by the amplifier unit 11, is generally related to the line impedance Z _{L of} the high-frequency line system. The mean azimuthal received power in the load resistor 9 is obtained with a sufficiently high slope g _{m of} the input characteristic of the field effect transistor 2: P at the = 1 2 · e 2 · l em 2 ( f ) · G ( f ) where l _em ² (f) is the azimuthal mean value of the square effective length of the passive antenna part 1 that occurs at each frequency, taking into account the difference with D _am (f). Equation (2) resulting effective area of the passive antenna part 1 as follows:

With D _pm from equation 5, the mean azimuthal reception power of the passive reference antenna is: P pm = λ 2 8 × π · e 2 Z 0 · D pm

Taking into account the gain requirement P _am / P _pm = V, the frequency curve for G (f) to be required according to the invention results in: G ( f ) = 1 R A ( f ) · D pm D at the ( f ) · V

In the case of a lossy passive antenna part 1 with the efficiency η, the directional factor D _am (f) is to be replaced by D _am (f) * η in equation (8). This does not change the other dimensioning rules.

In the case of approximately equal azimuthal mean values D _pm and D _am (f), the frequency dependence of G (f) should be proportional to 1 / R _a (f). V is chosen so large that D pm D at the ( f ) · V □ ( F V - 1)· T 0 4 · T A applies, then the noise contribution of the receiving system downstream of the active antenna to the total noise is negligibly small. If the condition specified in equation (1) is also met, then the sensitivity is dependent solely on the directivity of the passive antenna part 1 and on the prevailing interference. The minimum necessary mean azimuthal radiation density S _am for a signal-interference ratio = 1 is then: S at the ( f ) = k · T A · B D at the ( f ) and increases with 1 / η if D _am (f) is to be replaced by D _am (f) * η.

Taking into account the interference radiation emanating from the vehicle itself, the selection of a passive antenna part 1 suitable for an antenna according to the invention as the structure located on the vehicle in connection with the condition for R _A (f) specified in equation (1) and discussed in more detail below can therefore be unerringly thereby take place that the ratio T _A / D _on (f) for the transmission frequency range is found to be sufficiently large.

18a and 18b exemplify antenna configurations of possible passive antenna parts 1 of active antennas according to the invention. At the connection points 18, the impedance profiles Z _A (f) shown in the complex impedance plane in FIG. 18c are present as a function of the frequency. The area marked by hatching in the left margin area of the diagram is one-sided by the value R _Amin = const. bounded. Impedance profiles that run outside the area identified in this way thus meet the condition of negligible noise of the field effect transistor 2 specified in accordance with equation (1) in the presence of a specific interference radiation according to T _A. The diagram shows convincingly the advantage of an active antenna according to the invention over an active antenna according to FIG. 2b according to the prior art, which lies in the fact that, without an input-side adaptation means, all antenna structures meet this condition without an input-side transformation means. In FIG. 18c, the real parts of the passive antenna parts 1 shown in FIGS. 18a and b are plotted against the frequency of 76 to 108 MHz. The frequency curve of the real part of the input admittance 7 to be designed according to the invention at the input of the low-loss filter circuit 3 is therefore inverted in each case to the curve curves shown in FIG. 18d according to aspects discussed in connection with equations (3) and (8).

In the amplifier circuit 21 according to the invention there is naturally an upper limit for the size of the tolerable voltage effective at the input, which results in the reception field over the effective length l _e due to possible nonlinear effects such as intermodulation. The maximum tolerable voltage can be increased by selecting a suitable field effect transistor 2 and by selecting a suitable operating point and by other circuit measures known per se. According to the invention, equation (6) can be assigned a maximum tolerable azimuthal mean value l _em with a known azimuthal guide factor D _am (f), a maximum tolerable active component R _Amax . The range of values with R _A > R _{Amax which} is not permissible for dimensioning is also hatched in FIGS. 18c and 18d. The radiation resistances R _{A of} the impedance values of particularly favorable structures for use as a passive antenna part 1 are therefore outside the hatched value range with R _amin <R _A <R _Amax .

In a further advantageous embodiment of the invention, a predetermined antenna structure is supplemented by using a low-loss transformer with the transmission ratio ü, as indicated in FIG. 11, which together with the antenna structure - for example a heating field on the window pane - forms the passive antenna part 1. The broadband transmission ratio is advantageously chosen such that the impedance that can be measured at the output of the transmitter is placed with its real part in the range of values with R _Amin <R _A <R _Amax . It is advantageous to make the primary inductance sufficiently high-resistance.

The linearity requirement is determined by a sufficiently large negative feedback, through which in the Source line located admittance 7 met. This requires one in the transmission area comparatively low negative feedback, which according to the gain requirement e.g. is dimensioned according to equation (8), but so outside the transmission range is as large as possible. In an advantageous embodiment of the invention are used for implementation such low-loss filter circuits 3 preferably T-half filter or T-filter or

Chain circuits of such filters are used. Such filters are in their basic structure in the Figures shown. To correspond to a more complicated frequency response of G (f) can the individual elements are supplemented by further blind elements. In the interest of high impedance on the input side and the blocking effect in the blocking area, it is advisable To form a series or parallel branch from a combination of reactances in such a way that both the absolute value of a reactance in the series branch 28 and the absolute value of a reactive conductance in the parallel branch 29 in each case within a transmission frequency range is sufficiently small and is sufficiently large outside such (Fig. 6b).

In a further advantageous application of the invention, it is proposed to store corresponding basic structures for low-loss filter circuits 3 with initially unknown values for the dummy elements in a modern digital computer for different characteristic courses of G (f) and to measure both the impedance Z _{A of} the passive antenna part 1 and the measurement technology to determine the azimuthal mean value D _{am of} the guide factor by measurement or calculation and likewise to store it in the digital computer. The frequency curve of G (f) thus determined on the basis of equation (8) enables the subsequent concrete determination of the blind elements of the low-loss filter circuit 3 for a suitably selected basic filter structure with the aid of known strategies of variation calculation for the given gain V of the active antenna.

In particular in the case of antenna systems in which a plurality of antennas are formed, such as antenna diversity systems, group antenna systems or multi-range antenna systems, it is helpful in an advantageous development of the invention, as indicated in FIG. 3b, to design the amplifier unit 11 as the active output stage of the amplifier circuit 21. This can be provided with an output resistance equal to the characteristic impedance Z _{L of} conventional coaxial lines. The effective resistance 5 is formed by the input impedance of the amplifier unit 11. G (f) is to be designed in accordance with the above statements with the aid of a low-loss filter circuit 3 terminated with this impedance.

To increase the internal steepness and thus special linearity properties of the To achieve field effect transistor 2 can, in a further advantageous embodiment Invention, as shown in Fig. 4, an extended field effect transistor 2 using an input field effect transistor 13 and a bipolar transistor 14 driven by its source Emitter follower circuit can be designed.

When using an antenna according to the invention as an active window antenna it is advantageously possible, as shown in Fig. 5, the amplifier circuit 21 in the very to accommodate the narrow edge area of the vehicle window invisibly. That's why it is desirable to miniaturize the part to be attached to the connection point 18 and to attach only the parts of the amplifier circuit 21 that are functionally necessary there. The others Parts of the low-loss filter circuit 3 are placed separately and via the high-frequency line 10 turned on.

In a further advantageous embodiment of the invention, the active antenna is designed as a multi-range antenna for several frequency ranges. For this purpose, the basic frequency profiles of reactive resistors X ₁ , X ₃ and of the reactive conductance B _{2 of} a T-filter arrangement of the low-loss filter circuit 3 shown in FIG. 6 b are given as examples for the frequency ranges VHF radio broadcasting and VHF and UHF television broadcasting , The T-filter configuration ensures the high-impedance of the low-loss filter circuit 3 on the input side in order to achieve a sufficiently large negative feedback of the field effect transistor 2 in the blocking regions.

In a further advantageous embodiment, the creation of a plurality of frequency ranges Invention based on separate transmission paths for the concerned Transmission frequency bands take place. Here, as shown in Fig. 7, for designing multiple transmission frequency bands multiple bipolar transistors 14 to expand the Field-effect transistor 2 is used, the base electrodes of which are connected to the source electrode of the input field-effect transistor 13 are connected and which each have an emitter follower circuit the input of a separate low-loss filter circuit 3 to form separate Transmission paths for the relevant frequency bands are connected.

To compensate for the effects of the non-linearity of even order and the resulting ones resulting interband frequency conversions in the amplifier circuit 21 is in one Another advantageous embodiment of the invention according to FIG. 8 in addition to the field effect transistor 2 a further field effect transistor 17 with the same electrical properties is used. in this connection are the input terminals of the amplifier circuit 21 by the two control terminals of the Field effect transistors 15 and 16 formed and the input of the low-loss filter circuit 3 with connected to the source terminals 19a and 19b. A balun 20 in the low loss Filter circuit 3 is used to resymmetrize the high-frequency received signals 8. Such Circuit can, as in Fig. 17, advantageously also to a connection point 18 with two against Ground voltage-carrying connections can be connected.

In a further advantageous embodiment of the invention, the measures for Suppression of the non-linear effects according to the explanations given for FIG. 8 and the Creation of separate transmission paths according to the explanations given for FIG. 7 as in Fig. 9 combined. The design of separate frequency-selective transmission paths can advantageous, as indicated in FIG. 12, also on the basis of signal branches in the low-loss Filter circuit 3 for frequency-selective decoupling of high-frequency received signals 8 can be designed for different transmission frequency bands at several outputs.

In a particularly advantageous embodiment of the invention, the present active antenna is used several times in an antenna system, the passive antenna parts 1 of which have frequency-dependent and with respect to incident waves according to the amount and or only in phase, different directional diagrams of the effective lengths l _e , which, however, are electromagnetic Radiation coupling to each other and together form a passive antenna arrangement 27 with a plurality of connection points 18. According to the invention, each is connected to an amplifier circuit 21 according to the invention, so that the coupling-out of the high-frequency received signals 8 to the passive antenna parts 1 does not have any noticeable mutual influence on the received voltages. Such an antenna arrangement is shown in general in FIG. 13. The received signals 8 present at the output of the amplifier circuit 21 are superimposed, for example in order to design a group antenna arrangement with predetermined reception properties with regard to directivity and antenna gain, without retroactive effect on the high-frequency received signals 8 applied to the passive antenna parts 1 in an antenna combiner 22 provided for this purpose.

The efficiency of antenna diversity systems is shaped by the number of available, mutually independent antenna signals. This independence is expressed in the correlation factor between the received voltages occurring in a Rayleigh wave field while driving. In a particularly advantageous development of the invention, a plurality of active reception antennas according to the invention are used in an antenna diversity system for vehicles, the passive antenna parts 1 being selected such that their reception signals E * l _e present at the connection points 18 in an idle state in a Rayleigh reception field are diverse are as independent as possible. Systems of this type, in which the connection points 18 are selected from this point of view and taking into account vehicle-technical aspects, are shown by way of example in FIGS. 15a and 15b. Because of the electromagnetic radiation coupling existing between the connection points 18, this independence then only applies to the connection points 18 operated in idle mode. By connecting the connection points 18 to the amplifier circuits 21 according to the invention, due to their negligibly small capacitive input conductance, the high-frequency received signals 8 are tapped without feedback at the antenna outputs. The diversity-independent independence of the received signals at the connection points 18 is thus advantageously not influenced by this measure, and this independence consequently exists in the same way for the received signals 8 at the antenna outputs. Thus, independent reception signals 8 are available at the antenna outputs for selection in a scanning diversity system or for further processing in one of the other known diversity methods.

In contrast to this, the connection of the connection point 18 with a Transformation circuit according to the prior art according to Fig. 2b on the at Junction 18 flowing currents a dependence of the antenna signals on Cause antenna output. This connection is used below for a passive Antenna part 1 with two connection points 18 explained in more detail:

mit N = 1-Z11·Y1 - Z22·Y2 + Z11·Z22·Y1·Y2 - Z122·Y1·Y2 Are U01 and U02 the open circuit voltage amplitudes at the connection points 18 of a passive antenna arrangement 27 in the reception field and Z11, Z22 the antenna impedances measured there and Z12 is also the interaction impedance due to the coupling of the connection point 18 and Y1 and Y2 are the input admittances of the amplifiers with which the connection point 18 are loaded, the following relationship results for the voltage amplitudes occurring under this load at the connection points 18:

With N = 1- Z 11 · Y 1 - Z 22 · Y 2 + Z 11 · Z 22 · Y 1· Y 2 - Z 12 2 · Y 1· Y 2

The correlation factor between the voltage amplitudes U1 and U2 and thus also between the antenna output voltages results from the mean values of the voltages U1 and U2 over time: ρ = U 1· U 2 U 1 2 · U 2 2

For the case assumed here, no-load received voltage amplitudes U10 and U20 result from one another when traveling in the Rayleigh reception field. This is expressed by a vanishing correlation factor, ie: ρ = U 10 · U 20 U 10 2 · U 20 2 = 0

If the input admittances of the amplifiers with which the connection points 18 are loaded are negligibly small according to the invention, ie Y1 = 0 and Y2 = 0, then the voltages U1 and U2 result from equation (11) as follows:

The interactions with the number 0 in the unit matrix in equation (13) show that the vanishing decorrelation in voltages U1 and U2 described in equation (13) is retained in an amplifier circuit 21 according to the invention. The evaluation of equation (11), on the other hand, shows a link between the two open circuit voltages via the interaction parameters Z12 * Y2 and Z12 * Y1 with the respective voltages under load, because the following then applies: U 1 = (1 - Z 22 · Y 2) · U 10 + Z 12 · Y 2 · U 20 respectively. U 2 = (1 - Z 11 · Y 1)· U 20 + Z 12 · Y 1· U 10

It is obvious that if the connection points 18, i.e. not disappearing Z12, the correlation factor only disappears if Y1 = Y2 = 0 is.

On the other hand, the previous considerations show that if the open-circuit voltages U10 and U20 are mutually dependent, special values for Y1 and Y2 can be found which, via the transformation described in equation (15), reduce or disappear the mutual dependency in the amplifier input voltages U1 and U2 to let. In an advantageous development of the invention it is therefore; As indicated in FIG. 15, it is provided to connect the passive antenna arrangement 27 at its connection points 18 by means of suitable conductance values - preferably for reasons of sensitivity - so that the correlation between the voltages at the connection points 18 is smaller in the interest of greater diversity efficiency becomes. Active antennas according to the invention have the decisive advantage that the definition of such suitable dummy elements can be made largely independently of sensitivity considerations. Because the radiation resistances R _A (f) that arise at the various connection points 18 do not require any precise adjustment, but only require that they belong to the permissible value range described in FIG. 18.

List of names

Mass 0

Passive antenna part 1

Field effect transistor 2

Low loss filter circuit 3

Exit 4

effective resistance 5

Entrance 6

Entry admittance 7

High-frequency received signal 8

Load resistance 9

Radio frequency line 10

Amplifier unit 11

Emitter connection 12

Input field effect transistor 13

Bipolar transistor 14

Control connection 15, 16

Another field effect transistor 17

Junction 18

Source connection 19

Balancing element 20

Amplifier circuit 21

Antenna combiner 22

Reactive conductance 23

Transmitter 24

Electronic switch 25

Equivalent load resistance 26

passive antenna arrangement 27

Series branch 28

Parallel branch 29

Linearization resistor 30

Noise figure F _v

Active conductance G

effective length le

Wavelength λ

Boltzmann constant k

Wave resistance of free space Z ₀

Measuring bandwidth B

Claims (23)

Active broadband receiving antenna, consisting of a passive antenna part (1) with a frequency-dependent effective length l _e , the output connections of which are connected to the input connections of an amplifier circuit (21),
characterized in that
the amplifier circuit (21) consists of a field effect transistor (2) and a low-loss filter circuit (3) with an input admittance (7) and the low-loss filter circuit (3) is connected at its input (6) to the source terminal of the field effect transistor (2) and the high-frequency received signal (8) is coupled out at its output (4), and the low-loss filter circuit (3) is loaded with an active resistor (5) effective at its output (4) and the blind elements of the low-loss filter circuit (3) are selected in this way that the frequency dependency of the real part G of the input admittance (7) effective at the input (6) of the low-loss filter circuit (3) is set such that the frequency response in high frequency caused by the frequency-dependent effective length l _{e of} the passive antenna part (1) is required when the received power is required Received signal (8) is designed within a broad frequency band according to freely chosen criteria and the amount the input admittance (7) effective at the input (6) of the low-loss filter circuit (3) outside this frequency band is sufficiently small to avoid nonlinear effects in the blocking frequency range (Fig. 1). Active broadband receiving antenna for use above 30 MHz according to claim 1
characterized in that
the field effect transistor (2) has a negligible parallel noise _current source i _r , a very small gate-drain capacitance C ₁ and a very small gate-source capacitance C ₂ and negligible 1 / f noise and its minimum noise temperature T _N0 in the case of noise adaptation, it is significantly lower than the ambient temperature T ₀ (FIG. 2). Active broadband receiving antenna according to claim 2
characterized in that
the active conductance (5) effective at the output (4) of the low-loss filter circuit (3) is designed by the input resistance of a high-frequency line (10) loaded at its end with the load resistor (9) and the load resistor (9) by the input impedance of a further amplifier unit ( 11) is formed with the noise figure F _v and the real part G of the admittance (7) effective at the input (6) of the low-loss filter circuit (3) is chosen to be sufficiently large that the noise contribution of the amplifier unit (11) is smaller than the noise contribution of the field effect transistor (2) (Fig. 3a). Active broadband receiving antenna according to one of claims 1 to 3
characterized in that
the field effect transistor (2) is designed as an extended field effect transistor, consisting of an input field effect transistor (13), from the source of which the bipolar transistor (14) is driven in an emitter follower circuit and through whose emitter connection (12) the source electrode of the extended field effect transistor (2) is formed (Fig. 4). Active broadband reception antenna for FM radio reception in the car according to one of claims 1 to 4
characterized in that
the passive antenna part (1) is designed with a conductor structure printed on a dielectric carrier, such as a window pane or a plastic carrier, and the low-loss filter circuit (3) is designed as a bandpass filter with a pass in the VHF frequency range and high-impedance input impedance outside the VHF frequency range ( Fig. 1). Active broadband receiving antenna according to one of claims 1 to 5
characterized in that
for the purpose of spatial separation of a miniaturized front end of the active antenna, the low-loss filter circuit (3) contains a high-frequency line (10) as an element that transforms the effective admittance (7) in a frequency-dependent manner (FIG. 5). Active broadband receiving antenna according to one of claims 1 to 6
characterized in that
For the design of several transmission frequency bands, the frequency dependence of the active conductance G of the effective input admittance (7) of the low-loss filter circuit (3) is designed in such a way that the frequency response in the high-frequency received signal (8) is largely broadband compensated within each of the frequency bands and the amount of the input ( 6) the low-loss filter circuit (3) effective input admittance (7) outside these frequency bands is sufficiently small (Fig. 6a). Active broadband receiving antenna according to one of claims 1 to 6
characterized in that
for the design of several transmission frequency bands, there are several bipolar transistors (14) for expanding the field effect transistor (2), the base electrodes of which are connected to the source electrode of the input field effect transistor (13) and which are each in emitter follower circuit with the input of a low-loss filter circuit ( 3) are connected to form separate transmission paths for the relevant frequency bands (Fig. 7). Active broadband receiving antenna according to one of claims 1 to 8
characterized in that
to compensate for effects of the non-linearity of even order and the resulting interband frequency conversions in the amplifier circuit (21) in addition to the field effect transistor (2) there is another field effect transistor (17) with the same electrical properties and the input connections of the amplifier circuit (21) by the two Control connections of the field effect transistors (15, 16) are formed and the input of the low-loss filter circuit (3) is connected to the source connections (19a and 19b) and in the low-loss filter circuit (3) a resymmetrizing element (20) for resymmetrizing the high-frequency received signals ( 8) is present (Fig. (8). Active broadband receiving antenna according to claim 8 in combination with claim 9
characterized in that
for the amplifier circuit (21) in claim 9 for the design of a plurality of transmission frequency bands for the relevant frequency bands, separate transmission paths are designed according to claim 8 (FIG. 9). Active broadband receiving antenna according to one of Claims 7 to 10
characterized in that
the low-loss filter circuit (3) is designed as a T-half filter or T-filter or as a chain circuit of such filters, the series or parallel branch of which is formed in each case from a combination of reactances in such a way that both the absolute value of a reactance in the series branch (28) and the absolute value of a reactive conductance in the parallel branch (29) is in each case sufficiently small within a transmission frequency range and sufficiently large outside such (FIG. 6b). Active broadband receiving antenna according to one of claims 1 to 11
characterized in that
Between the source connection of the field effect transistor (2) and the input connection of the filter circuit (3) an ohmic linearization resistor (30), the resistance value of which is smaller than the equivalent noise resistance R _{a of} the field effect transistor (2), is connected to further increase the linearity (Fig 10). Active broadband receiving antenna according to one of claims 1 to 12
characterized in that
for the broadband creation of favorable transmission conditions in the filter circuit (3) there is a transformer (24) with a suitable transmission ratio ü (FIG. 10). Active broadband receiving antenna according to one of claims 1 to 3
characterized in that
To increase the effective length l _{e of} the passive antenna part (1) between its connection point (18) and the input of the amplifier circuit (21), a transformer (24) with a sufficiently high-impedance primary inductance and a suitably chosen transmission ratio is present (Fig. 11). Active broadband receiving antenna according to one of claims 1 to 14
characterized in that
frequency-selective transmission paths for frequency-selective decoupling of high-frequency received signals (8) for different transmission frequency bands at several outputs are designed in the low-loss filter circuit (3) on the basis of signal branches (FIG. 12). Active broadband receiving antenna according to one of claims 1 to 15
characterized in that
there are a number of passive antenna parts (1) with frequency-dependent directional diagrams of the effective lengths l _e which differ in terms of magnitude and phase with respect to incident waves, which are in electromagnetic radiation coupling to one another and together form a passive antenna arrangement (27) with a plurality of connection points (18), Each of which is connected to an amplifier circuit (21) according to claims 1 to 12, so that the coupling of the high-frequency received signals (8) to the passive antenna parts (1) means that there is no noticeable mutual influence on the received voltages (Fig. 13) , Active broadband receiving antenna according to one of claims 1 to 16
characterized in that
The reception signals (8) present at the output of the amplifier circuit (21) for designing a group antenna arrangement with predetermined reception properties with regard to directivity and antenna gain without having an effect on the high-frequency reception signals (8) applied to the passive antenna parts (1) in an antenna combiner (22) provided for this purpose Amount and phase are superimposed (Fig. 13). Active broadband receiving antenna according to claim 16
characterized in that
the active receiving antennas are used in an antenna diversity system for vehicles and the passive antenna parts (1) are selected in such a way that their received signals in a Rayleigh reception field are as independent of one another in terms of diversity as possible and the high-frequency received signals (8) are non-reactive, i.e. without the diversity of independence Influencing received signals are provided for selection in a scanning diversity system or for further processing in one of the other known diversity methods (FIG. 14). Active broadband receiving antenna according to claim 18
characterized in that
To improve the diversity-independent of the received signals of the passive antenna parts (1) whose connection points (18) are loaded with separately determined reactive conductance values (23) parallel to the input of the amplifier circuit (21) (Fig. 16). Active broadband receiving antenna according to one of Claims 17 and 18
characterized in that
the passive antenna arrangement (27) is present as conductor structures on a plastic carrier inserted into the recess of a conductive vehicle body or on the window pane of a vehicle, for example in the form of one or more heating fields and / or conductor structures separate from the heater, and a plurality of connection points (18 ) for the formation of passive antenna parts (1) for connecting amplifier circuits (21) are present (Fig. 15a, b). Active broadband receiving antenna according to one of Claims 17 to 20
characterized in that
the passive antenna arrangement (27) is designed as an essentially coherent, conductive surface for suppressing the radiation transmission in the infrared range with a sufficiently small surface resistance on the window pane of a car and suitably positioned for decoupling reception signals at the edge of the conductive surface that is not connected to the conductive body Connection points (18) with amplifier circuits (21) are formed, the high-frequency received signals (8) via high-frequency lines (10) for designing a directional antenna, an antenna combiner (22) or for designing a scanning diversity system, an electronic switch (25), or to design a diversity arrangement operating according to any other method. Active broadband receiving antenna according to one of claims 1 to 21
characterized in that
the passive antenna part is derived from a vehicle part that was not originally intended for use as an antenna and its design can be changed only slightly, and a connection point (18) for forming a passive antenna part (1) is formed on this element and for the polarization applicable in the useful frequency range and the elevation of an incident wave, a specific azimuthal mean value D _{m of} the directional factor is determined and the real part R _{A of} the impedance Z _{A of} the passive antenna part (1) is given in the transmission frequency range in the range between R _amine and a maximum value R _Amax (FIG. 18). Active broadband receiving antenna according to one of claims 1 to 22
characterized in that
A modern digital computer is available and both the impedance Z _{A of} the passive antenna part (1) is measured or calculated and the azimuthal average value D _{m of} the directional factor, determined by measurement or calculation, is stored in the digital computer and in which suitable for various characteristic possible frequency profiles of antenna impedances are stored Basic structures for low-loss filter circuits (3) are stored in the digital computer and with the aid of known strategies of variation calculation, the blind elements of the low-loss filter circuit (3) are determined for a predetermined average gain of the active antenna.

Images