EP1212806B1 - High-frequency band pass filter assembly, comprising attenuation poles - Google Patents

High-frequency band pass filter assembly, comprising attenuation poles Download PDF

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Publication number
EP1212806B1
EP1212806B1 EP00960529A EP00960529A EP1212806B1 EP 1212806 B1 EP1212806 B1 EP 1212806B1 EP 00960529 A EP00960529 A EP 00960529A EP 00960529 A EP00960529 A EP 00960529A EP 1212806 B1 EP1212806 B1 EP 1212806B1
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EP
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Prior art keywords
resonator
resonators
frequency
bandpass filter
bandstop
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EP00960529A
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German (de)
French (fr)
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EP1212806A1 (en
Inventor
Heinz Chaloupka
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Cryoelectra GmbH
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Cryoelectra GmbH
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    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P1/00—Auxiliary devices
    • H01P1/20—Frequency-selective devices, e.g. filters
    • H01P1/201—Filters for transverse electromagnetic waves
    • H01P1/203—Strip line filters
    • H01P1/20327—Electromagnetic interstage coupling
    • H01P1/20354—Non-comb or non-interdigital filters
    • H01P1/20381—Special shape resonators
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P1/00—Auxiliary devices
    • H01P1/20—Frequency-selective devices, e.g. filters
    • H01P1/201—Filters for transverse electromagnetic waves
    • H01P1/202—Coaxial filters
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P1/00—Auxiliary devices
    • H01P1/20—Frequency-selective devices, e.g. filters
    • H01P1/201—Filters for transverse electromagnetic waves
    • H01P1/203—Strip line filters
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P1/00—Auxiliary devices
    • H01P1/20—Frequency-selective devices, e.g. filters
    • H01P1/201—Filters for transverse electromagnetic waves
    • H01P1/203—Strip line filters
    • H01P1/20327—Electromagnetic interstage coupling
    • H01P1/20354—Non-comb or non-interdigital filters
    • H01P1/20363—Linear resonators
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P1/00—Auxiliary devices
    • H01P1/20—Frequency-selective devices, e.g. filters
    • H01P1/207—Hollow waveguide filters
    • H01P1/209—Hollow waveguide filters comprising one or more branching arms or cavities wholly outside the main waveguide

Definitions

  • the invention relates to a high-frequency bandpass filter arrangement, consisting of one Main resonator and at least one to the main resonator coupled blocking resonator, the main resonator by one, on both sides by discontinuities in Shape of a break or metal wall limited Line section is defined, and at a center frequency has an electromagnetic natural vibration.
  • the invention relates to the structure of bandpass filters from coupled resonators for highly selective filtering of high frequency electromagnetic signals in one Operating frequency range, which is above about 0.5 GHz and is below about 100 GHz.
  • High frequency bandpass filters are an important one Component in systems of communication technology, such as B. in terrestrial and satellite-based round, Directional and mobile radio as well as in radar and Navigation systems.
  • B. in Individual filters the function of the radio receiver Preselection, i.e. suppressing unwanted Interference signals and filter banks the function of the Frequency drains.
  • Individuals serve in radio transmitters
  • Bandpass filter u. a. to suppress out-of-band spectral components in the output signal of the amplifier and Filter banks are used in the form of output multiplexers Merging different carrier signals on one common antenna.
  • passive electromagnetic filter is based on the Storage of electrical and magnetic field energy.
  • the Storage of electrical and magnetic field energy separately from each other, in a finite number spatial separate discrete elements, namely in capacities and Inductors instead. Because the geometric dimensions of these discrete components much smaller than that Operating wavelength, typically less than one Tenths of the guided wavelength, must be and on the other hand, the idle quality of these components Reduction in dimensions decreases sharply for steep-sided filters preferred above about 1 GHz Coupled resonator structures instead of Interconnections of discrete capacities and Inductors used.
  • resonators which are the building blocks of the filter class considered here is one large number of different types to choose from.
  • Out coaxial TEM line pieces and waveguide pieces become coaxial resonators or cavity resonators formed where the electromagnetic field is completely enclosed by conductive surfaces.
  • These resonators can be used for volume reduction and Partial change in the spatial field profile or completely with low loss dielectric material be filled. This takes place in dielectric resonators Field confinement mainly through the interface between the dielectric material and the surrounding one Air and that from this interface to the outside space decaying field is possibly by metal casing shielded.
  • the selection of the design of the resonators is u. a. of that of the filter specification (see below) required idle quality of the resonators affected.
  • a high idle quality means in conventional Technology has a relatively large geometric dimension Resonators.
  • this is in the lower GHz range for the entirety of all resonators of a filter Available volume is limited.
  • a reduction in Volume requirements of around 50% can be obtained from Dual use of resonators via orthogonal modes (Dual-mode resonators).
  • An exception to the rule that high idling quality, large geometric dimensions mean, is achieved when using cooled planar High temperature superconductor resonators.
  • the electrical behavior of a bandpass filter becomes characterized by frequency bandwidth (Pass width) and position of the pass band the maximum insertion loss and minimum Reflection attenuation in the pass band, by the width the transition areas between passband and Exclusion area as well as the minimal blocking attenuation in the Stop band.
  • the filter's resonators become the frequency response of the filter degraded in such a way that the achievable steepness of the Filter edges is limited by rounding effects and the dissipative insertion loss in the pass band is increased. Since this degradation is a first approximation only from N and not from the number M of transmission zeros depends, so you can at given Idle quality of the resonators filters with higher Slope steepness and less dissipative Realize insertion loss by increasing M / N.
  • the one with filters from coupled resonators today predominantly followed path to the generation of Transmission zeros consist in the introduction of Couplings between not directly adjacent resonators ("Couplings"), in addition to the direct couplings neighboring resonators.
  • Couplings with a suitable strength and sign that is, couplings between non-adjacent ones Resonators lead to transmission zeros in the restricted areas, whereby pro Overcoupling, depending on the position of the coupling path, one or two Transmission zeros are produced.
  • Ratio M / N and the greatest freedom of choice the frequency position of the individual transmission zeros this leads to a coupling scheme, which as "canonical coupling structure" is called and at even number N using N-2 different overcouplings on N-2 freely placeable Transmission zeros leads.
  • M N-2 zeros, which are symmetrical to the pass band one has at least (N-2) / 2 couplings.
  • This band stop serves Interference frequencies outside the pass band to eliminate.
  • FIG. 6 of this publication shows the layout of a 5-pin bandpass using microstrip technology.
  • the 4 transmission zeros are realized by 4 additional resonators in the form of ⁇ / 4 stub lines.
  • Figure 8 of the publication shows a 3-pin filter made of 3 parallel-coupled microstrip line resonators.
  • the 2 transmission zeros are realized in that the middle resonator is provided with 2 blocking resonators in the form of stub lines.
  • a band-stop filter with two mutually separate blocking ranges can also be used instead of a bandpass filter.
  • the used pass band lies between the two band band stop band.
  • a filter of this type is known from US Pat. No. 5,291,161 A, which consists of a continuous main line and galvanically coupled stub lines and in which each stub line generates a transmission zero point. From IC Hunter and JR Rhodes "Electronically tunable microwave bandstop filters" in IEEE Transactions on Microwave Theory and Techniques, vol. MTT-30, No.
  • the second disadvantage is that the number N of attenuation zeros is less than the number of resonators in the pass band used between the two stop areas and therefore the maximum steepness of the filter edges between pass band and stop area that can be achieved for a given resonator number N R cannot be achieved.
  • the object is achieved by the in the Objects described claims solved.
  • the invention uses bandpass filter structures proposed in which blocking resonators such the structure is integrated that each blocking resonator both one of the desired transmission zeros in the Restricted area as well as together with the rest Filter structure an additional damping zero in the Passband realized.
  • impedance-symmetrical and impedance-unbalanced filter elements are to be understood that with an impedance-symmetrical Filter element when connecting the input and output gates with the same termination resistance, the maximum values of the Power transmission factor with negligible Losses reach one, while at transmission unbalanced filter element complete Power transmission only for highly asymmetrical gate resistors is achievable.
  • Figure 1e shows the principle in a schematic manner Structure of an impedance-symmetrical according to the invention
  • Figures 1a to 1d show schematic way structures, which the state of the Correspond to technology and therefore only for gradual Explanation of the basic principle of the invention Serve structure according to Figure 1e.
  • Fig. 1a symbolically shows a homogeneous high-frequency line 1, in which this line as a metallic TEM line z. B. as a coaxial line, as a planar line such. B. a microstrip line or strip line or coplanar line, or as a waveguide or as a dielectric line.
  • FIG. 1b schematically shows a structure modified compared to FIG. 1a, in which two discontinuities 3 are inserted symmetrically into the cable run.
  • These discontinuities define a line section of finite length a, on which electromagnetic natural vibrations occur at those frequencies at which the length a corresponds to an integer multiple of half the line wavelength, and these natural vibrations are characterized by standing waves with nodes and antinodes of the electrical and magnetic field strength along the line , with a node of the electric or magnetic field strength existing in the plane of symmetry 4 at the resonance frequency.
  • the discontinuities limiting the line piece can technically z. B. in the form of line interruptions or in the form of metallic diaphragms, and it is also well known in the art that about the strength of the coupling between the leads and the ends of the line section serving as a resonator, the frequency bandwidth ⁇ f of the transmission curve can be changed.
  • Figure 1c shows a structure modified from Figure 1a, in which a resonance circuit 6 ("blocking resonator”) is coupled to the line, so that the frequency response of the power transmission factor 7 has a transmission zero at the frequency f s .
  • This structure represents the construction of a unipolar bandstop ("notch filter”), which is well known in the prior art.
  • FIG. 1d shows a structure modified from FIG. 1c, in which instead of a blocking resonator two blocking resonators 8 with different resonance frequencies are coupled and lead to two transmission zeros at f s1 and f s2 .
  • An essential aspect of the invention consists in forming the structure according to FIG. 1e from a combination of the structure according to FIG. 1b and the blocking resonator pair from FIG. 1d.
  • the line section of finite length forms a resonator, here referred to as the main resonator, which has a node of the electric or magnetic field in the middle.
  • An essential aspect of the invention is the choice of the coupling between the blocking resonators and the main resonator in such a way that this coupling disappears at the frequency f 0 . B.
  • the two blocking resonators thus assume a double function in that on the one hand they realize two transmission zeros - as in the structure according to FIG.
  • the frequency response 10 of the structure according to FIG. 1e is thus characterized by a suitable choice of the resonance frequencies and coupling strengths by three transmission maxima (damping zeros) at f 1 , f 2 and f 3 and two transmission zeros at f s1 and f s2 .
  • the frequency position of the transmission zeros is determined by the resonance frequencies of the blocking resonators and the frequency position of the average transmission maximum by the length of the main resonator.
  • the position of the two outer transmission maxima can be changed by the coupling strength between the main resonator and blocking resonators, with an increase in the coupling, these frequencies shifting towards the middle frequency.
  • the electrical field at frequency f 0 has a node in the plane of symmetry and thus the two blocking resonators must be electrically coupled according to the above design rules, while in the case of magnetic field maxima at the ends, because of the node of the magnetic one Field, a magnetic coupling must be present.
  • the length of the line section must correspond to a full wavelength instead of half the center frequency line wavelength.
  • N N g xQ
  • M NQ transmission zeros
  • An impedance-unbalanced filter element becomes realized according to the invention in that an impedance-symmetrical Filter element with a pair of blocking resonators 1e is modified, one of the two Discontinuities are brought close to the site which is coupled to the blocking resonator pair.
  • an impedance-symmetrical Filter element with a pair of blocking resonators 1e is modified, one of the two Discontinuities are brought close to the site which is coupled to the blocking resonator pair.
  • the impedance-symmetrical Link 5 located at one end of the cascade or it can be inserted centrally (see Fig. 4c).
  • Fig. 5 shows an example of the implementation of a 7-pin Filters with 6 transmission zeros in the form a single filter element according to that in Fig. 2c principle shown in coaxial line technology.
  • the Main resonator 1 has a rectangular exterior and Inner conductor and a length equal to 1.5 times that Center frequency wavelength.
  • the the line piece limiting discontinuities are more capacitive in form Coupler trained.
  • the blocking resonators 2 are as on End of short-circuited coaxial line pieces of one length of a quarter of a line wavelength, which are capacitively coupled to the main resonator.
  • Fig. 6 shows a modification of the structure of Fig. 5, by now the blocking resonators 2 galvanically with the Inner conductors of the main resonator are connected, but on Are capacitively loaded at the end.
  • Fig. 7 shows a structure of two impedance-unbalanced Filter elements and an impedance-symmetrical Link with 9 poles and 8 Receives transmission zeros.
  • the main resonator 1 consists of a short-circuited at both ends Rectangular waveguide, which at the center frequency Has a length corresponding to a waveguide wavelength.
  • the 4 blocking resonators 2 are in the form of short-circuited 1/4 waveguide pieces realized.
  • the coupling to the Gates can e.g. B. via a coaxial transition 3.
  • FIG. 9 shows an example of an implementation with dielectric resonators in the case of a filter comprising two impedance-symmetrical filter elements, each filter element producing three poles and two transmission zeros and thus the bandpass filter having a total of 6 poles and 4 transmission zeros.
  • the dimension of the main resonator is chosen so that it has a natural resonance at f 0 with the field distribution shown in FIG. 9b, and the dimension of the blocking resonators are chosen so that they resonate at the 4 blocking frequencies f 1 to f 4 and thereby have a field distribution corresponding to FIG. 9c. Because of the spatial field distribution of the main resonator, it does not couple to the resonance fields of the blocking resonators at f 0 . For frequencies different from f 0 , however, a coupling is obtained between the main resonator and the blocking resonators, with the result that an additional 4 natural resonances arise.
  • the coupling to the gates can e.g. B. via conductor loops 4.
  • the main resonator 5 consists of a dielectric cuboid of length a, which corresponds approximately to a wavelength of the surface wave on the dielectric cuboid. A field distribution corresponding to FIG. 10b is thereby obtained on the main resonator.
  • the 4 blocking resonators 1 to 4 also consist of dielectric cuboids, the individual lengths b1 to b4 of which influence the frequency position of the 4 transmission zeros.
  • the entire structure of the main dielectric resonator and 4 dielectric blocking resonators realizes 5 natural vibrations.
  • the frequency position of the poles can be changed via the coupling strength between the main and blocking resonators.
  • the "gaps" between the resonators with the widths h 1 to h 4 filled with air or a dielectric material with a relatively low dielectric constant serve to change this coupling strength.
  • the principle according to the invention can also apply to planar ones Resonator structures, such as B. microstrip line structures are used, including microstrip line structures from high temperature superconductors from Are interested as this despite an enormous Degree of miniaturization over a high idling quality feature.
  • FIG. 11 illustrates the implementation of an impedance-asymmetrical filter element according to the invention using microstrip line technology.
  • Fig. 11a the principle of a microstrip line resonator, which is well known in the prior art, is first brought to mind.
  • FIG. 11a shows the well-known structure of a microstrip line resonator 3, which at its ends is capacitively connected to the leads 4, 5 is coupled.
  • the frequency response of the power transmission factor 6 shows a maximum at the frequency f 0 and the width of this maximum can be changed via the strength of the coupling at the line ends (discontinuities).
  • FIG. 11b shows how an impedance-asymmetrical filter element according to the invention can be realized in microstrip line technology.
  • a T-shaped conductor structure is used in which the length of the individual arms corresponds to approximately a quarter of the line wavelength at the center frequency, a well-defined asymmetry in the length or width of the side arms 3 being necessary for the function.
  • the side arms represent a simple implementation of the blocking resonators, the blocking frequencies being influenced over the length of the arms. Together with the third arm, the side arms form a structure which resonates at two different frequencies and thus the T-structure is a special form of a dual-mode resonator.
  • the output gate can be capacitively connected to the T in the manner shown in FIG. 11b Structure to be coupled.
  • the frequency response 6 of the two-port thus created is characterized by two transmission maxima and two transmission zeros, the absolute value of the transmission maximum being able to be far below one due to the asymmetry. For this reason, a single asymmetrical filter element - in contrast to the impedance-symmetrical filter element - is not yet a usable bandpass filter.
  • this microstrip line structure can also be modified in a variety of ways, e.g. B. by using inhomogeneous line pieces of variable width.
  • Fig. 12 shows an example of how 4 impedance-unbalanced Filter members 1 and one conventional half-wave resonator 2 a 9-pin Filters with 8 transmission zeros are formed can.
  • the resonator 2 takes over in the cascade Providing an additional pole that Transformation of the impedance at gate 2 (e.g. 50 ohms) the low impedance level at the coupling point to Branch point of the T-shaped resonators.
  • the Dimensioning the parameters of each Filter elements for example. B. done so that a Cauer characteristic for the frequency response is achieved.

Abstract

The invention relates to a high-frequency band pass filter assembly, comprising a master resonator and at least one stop-band resonator which is coupled thereto. The aim of the invention is to construct a filter structure in such a way, that with a given number of poles, the highest possible number of transmission zero positions occur in the stop bands, whereby in relation to known resonator configurations, no overcoupling is used between non-adjoining resonators. To this end, the stop-band resonator(s) is/are coupled to the master resonator in such a way that the stop-band resonator generates both transmission zero positions and transmission pole positions in tandem with the master resonator.

Description

Die Erfindung betrifft eine Hochfrequenz-Bandpassfilteranordnung, bestehend aus einem Hauptresonator und mindestens einem an den Hauptresonator angekoppelten Sperr-Resonator, wobei der Hauptresonator durch ein, an beiden Seiten durch Diskontinuitäten in Form einer Unterbrechung oder Metallwand begrenztes Leitungsstück definiert ist, und bei einer Mittenfrequenz eine elektromagnetische Eigenschwingung aufweist. Insbesondere bezieht sich die Erfindung auf den Aufbau von Bandpass-Filtern aus gekoppelten Resonatoren zur hochselektiven Filterung hochfrequenter elektromagnetischer Signale in einem Betriebsfrequenzbereich, welcher oberhalb von ca. 0,5 GHz und unterhalb von ca. 100 GHz liegt.The invention relates to a high-frequency bandpass filter arrangement, consisting of one Main resonator and at least one to the main resonator coupled blocking resonator, the main resonator by one, on both sides by discontinuities in Shape of a break or metal wall limited Line section is defined, and at a center frequency has an electromagnetic natural vibration. In particular, the invention relates to the structure of bandpass filters from coupled resonators for highly selective filtering of high frequency electromagnetic signals in one Operating frequency range, which is above about 0.5 GHz and is below about 100 GHz.

Hochfrequenz-Bandpassfilter bilden eine wichtige Komponente in Systemen der Kommunikationstechnik, wie z. B. im terrestrischen und satellitengestützten Rund-, Richt- und Mobilfunk als auch in Radar- und Navigationssystemen. Hierbei übernehmen z. B. in Funkempfangern einzelne Filter die Funktion der Vorselektion, also des Unterdrückens unerwünschter Interferenzsignale und Filterbänke die Funktion der Frequenzkanalisation. In Funksendern dienen einzelne Bandpassfilter u. a. zur Unterdrückung von Außerband-Spektralanteilen im Ausgangssignal der Verstärker und Filterbänke dienen in Form von Ausgangsmultiplexern zum Zusammenführen verschiedener Trägersignale auf eine gemeinsame Antenne.High frequency bandpass filters are an important one Component in systems of communication technology, such as B. in terrestrial and satellite-based round, Directional and mobile radio as well as in radar and Navigation systems. Here take over z. B. in Individual filters the function of the radio receiver Preselection, i.e. suppressing unwanted Interference signals and filter banks the function of the Frequency drains. Individuals serve in radio transmitters Bandpass filter u. a. to suppress out-of-band spectral components in the output signal of the amplifier and Filter banks are used in the form of output multiplexers Merging different carrier signals on one common antenna.

Bei Hochfrequenz-Bandpassfiltern kann zunächst eine Unterscheidung zwischen aktiven und passiven Ausführungen vorgenommen werden. Bei hohen Anforderungen an die Linearität und Rauscharmut kommen nur die hier weiter betrachteten passiven Filter in Frage. Die Funktion passiver elektromagnetischer Filter beruht auf der Speicherung elektrischer und magnetischer Feldenergie. Bei Filtern aus diskreten Bauelementen findet die Speicherung elektrischer und magnetischer Feldenergie separat voneinander, in einer endlichen Zahl räumlich getrennter diskreten Elemente, nämlich in Kapazitäten und Induktivitäten statt. Da die geometrischen Abmessungen dieser diskreten Bauelemente sehr viel kleiner als die Betriebswellenlänge, typischerweise kleiner als ein Zehntel der geführten Wellenlänge, sein müssen und andererseits die Leerlaufgüte dieser Bauelemente mit Verkleinerung der Abmessungen stark abnimmt, werden für steilflankige Filter oberhalb von ca. 1 GHz bevorzugt Strukturen aus gekoppelten Resonatoren anstelle von Zusammenschaltungen aus diskreten Kapazitäten und Induktivitäten benutzt.With high-frequency bandpass filters, one can initially Differentiation between active and passive executions be made. With high demands on the Linearity and low noise levels only get ahead here considered passive filters in question. The function passive electromagnetic filter is based on the Storage of electrical and magnetic field energy. For filters made of discrete components, the Storage of electrical and magnetic field energy separately from each other, in a finite number spatial separate discrete elements, namely in capacities and Inductors instead. Because the geometric dimensions of these discrete components much smaller than that Operating wavelength, typically less than one Tenths of the guided wavelength, must be and on the other hand, the idle quality of these components Reduction in dimensions decreases sharply for steep-sided filters preferred above about 1 GHz Coupled resonator structures instead of Interconnections of discrete capacities and Inductors used.

Für die Bauformen von Resonatoren, welche die Bausteine der hier betrachteten Filterklasse darstellen, steht eine große Zahl unterschiedlicher Typen zur Auswahl. Aus koaxialen TEM-Leitungsstücken und Hohlleiterstücken werden Koaxialresonatoren bzw. Hohlraumresonatoren gebildet, bei denen das elektromagnetische Feld vollständig durch leitende Flächen eingeschlossen wird. Diese Resonatoren können zur Volumensreduktion und zur Veränderung des räumlichen Feldverlaufs teilweise oder vollständig mit verlustarmen dielektrischen Material gefüllt werden. In dielektrischen Resonatoren erfolgt der Feldeinschluß hauptsächlich durch die Grenzfläche zwischen dem dielektrischen Material und der umgebenden Luft und das von dieser Grenzfläche nach außen räumlich abklingende Feld wird gegebenenfalls durch Metallgehäuse abgeschirmt. Planare Resonatoren, zu denen Mikrostreifenleitungs- , Streifenleitungs- und Koplanarresonatoren gehören, bestehen aus planaren Leiterbahnen auf einem dielektrischen Substrat.For the designs of resonators, which are the building blocks of the filter class considered here is one large number of different types to choose from. Out coaxial TEM line pieces and waveguide pieces become coaxial resonators or cavity resonators formed where the electromagnetic field is completely enclosed by conductive surfaces. These resonators can be used for volume reduction and Partial change in the spatial field profile or completely with low loss dielectric material be filled. This takes place in dielectric resonators Field confinement mainly through the interface between the dielectric material and the surrounding one Air and that from this interface to the outside space decaying field is possibly by metal casing shielded. Planar resonators to which Microstrip, stripline and Coplanar resonators are made up of planar ones Conductor tracks on a dielectric substrate.

Die Auswahl der Bauform der Resonatoren wird u. a. von der von der Filterspezifikation (siehe unten) erforderlichen Leerlaufgüte der Resonatoren beeinflußt. Eine hohe Leerlaufgüte bedeutet in konventioneller Technologie eine relativ große geometrische Abmessung der Resonatoren. Andererseits ist im unteren GHz-Bereich das für die Gesamtheit aller Resonatoren eines Filters zur Verfügung stehende Volumen begrenzt. Eine Reduktion des Volumenbedarfs um ca. 50 % erhält man durch Doppelausnutzung von Resonatoren über orthogonale Moden (Dual-Mode-Resonatoren). Eine Ausnahme von der Regel, daß hohe Leerlaufgüten große geometrische Abmessungen bedeuten, erreicht man bei Verwendung gekühlter planarer Resonatoren aus Hochtemperatur-Supraleitern. Eine weitere technologische Entwicklung in Richtung auf kompakte Hochgüteresonatoren ergibt sich aus den Fortschritten bei der Entwicklung extrem verlustarmer dielektrischer Materialien mit hoher Dielektrizitätszahl für dielektrische Resonatoren. Auf die Auswahl der Resonator-Bauform hat auch die geforderte Leistungsverträglichkeit (Erwärmung, Multipakting) einen Einfluß. The selection of the design of the resonators is u. a. of that of the filter specification (see below) required idle quality of the resonators affected. A high idle quality means in conventional Technology has a relatively large geometric dimension Resonators. On the other hand, this is in the lower GHz range for the entirety of all resonators of a filter Available volume is limited. A reduction in Volume requirements of around 50% can be obtained from Dual use of resonators via orthogonal modes (Dual-mode resonators). An exception to the rule that high idling quality, large geometric dimensions mean, is achieved when using cooled planar High temperature superconductor resonators. Another technological development towards compact High-quality resonators result from the progress made the development of extremely low loss dielectric Materials with high dielectric constant for dielectric resonators. On the selection of the resonator design also has the required performance compatibility (Warming, multipacting) an influence.

Das elektrische Verhalten eines Bandpassfilters wird charakterisiert durch Frequenz-Bandbreite (Durchlaßbreite) und Lage des Durchlassbereichs, durch die maximale Einfügungsdämpfung und minimale Reflexionsdämpfung im Durchlaßbereich, durch die Breite der Übergangsbereiche zwischen Durchlaßbereich und Sperrbereich sowie durch die minimale Sperrdämpfung im Sperrbereich.The electrical behavior of a bandpass filter becomes characterized by frequency bandwidth (Pass width) and position of the pass band the maximum insertion loss and minimum Reflection attenuation in the pass band, by the width the transition areas between passband and Exclusion area as well as the minimal blocking attenuation in the Stop band.

Zur weiteren quantitativen Charakterisierung der Eigenschaften einer Filterstruktur wird die Zahl N der Dämpfungs-Nullstellen (Reflexions-Nullstellen) im Durchlaßbereich und die Zahl M der Dämpfungspole (Transmissions-Nullstellen) bei endlichen Frequenzen im Sperrbereich herangezogen. Bei dieser Charakterisierung durch Reflexions-Nullstellen und Transmissions-Nullstellen wird das Verhalten im (fiktiven) verlustfreien Fall zugrunde gelegt und Nullstellen werden entsprechend ihrer Ordnung mehrfach gezählt.For further quantitative characterization of the The number N of the properties of a filter structure Damping zeros (reflection zeros) in the Passband and the number M of the damping poles (Transmission zeros) at finite frequencies in Exclusion area used. With this characterization through reflection zeros and transmission zeros the behavior in (fictitious) lossless case and zeroing counted several times according to their order.

Zur Realisierung eines Bandpassfilters können NR Resonatoren untereinander so verkoppelt werden, daß das Gesamtsystem aus gekoppelten Resonatoren insgesamt N =NR Dämpfungs-Nullstellen im Bereich des Durchlaßbereichs aufweist (N=2NR bei Doppelausnutzung von Resonatoren). Weiterhin kann durch geeignete Koppelmaßnahmen (siehe weiter unten) erreicht werden, daß in den Sperrbereichen insgesamt M<N Dämpfungspole (Transmissions-Nullstellen) bei endlichen Frequenzen auftreten.To implement a bandpass filter, N R resonators can be coupled to one another in such a way that the overall system of coupled resonators has a total of N = N R damping zeros in the region of the pass band (N = 2N R when resonators are used twice). Furthermore, by means of suitable coupling measures (see further below) it can be achieved that a total of M <N damping poles (transmission zeros) occur at finite frequencies in the restricted areas.

Aus dem Verhältnis der Übergangsbreite zur Durchlaßbreite ("relative Steilheit der Filterflanken") folgt die Zahl N der notwendigen Dämpfungs-Nullstellen und somit die Mindestzahl notwendiger Resonatoren. From the ratio of the transition width to the passage width ("Relative slope of the filter edges") follows the number N the necessary damping zeros and thus the Minimum number of resonators required.

Für die folgende Beschreibung der mit der Erfindung erzielten Vorteile ist von großer Wichtigkeit, daß bei gegebener relativer Steilheit der Filterflanken die notwendige Zahl N von Dämpfungs-Nullstellen im Durchlassbereich mit wachsendem M/N monoton abnimmt. Bei gegebener Durchlaßbreite kommt man für eine verlangte Flankensteilheit mit einer geringeren Zahl N, und damit einer geringeren Zahl NR von Resonatoren aus, wenn man anstelle eines Tschebyscheff-Filters mit M = 0, ein quasi-elliptisches Filter mit M>0 verwendet. Die erforderliche Zahl N wird weiter verringert, wenn man anstelle eine quasi-elliptischen Filters mit M < N-1 ein "echt elliptisches" Filter mit M= N-1 verwendet.For the following description of the advantages achieved with the invention, it is of great importance that, given the relative steepness of the filter edges, the necessary number N of attenuation zeros in the pass band decreases monotonically with increasing M / N. For a given pass width, a smaller number N, and thus a smaller number N R, of resonators are sufficient for a required edge steepness if a quasi-elliptical filter with M> 0 is used instead of a Chebyshev filter with M = 0. The required number N is further reduced if, instead of a quasi-elliptical filter with M <N-1, a "really elliptical" filter with M = N-1 is used.

Aufgrund der ohmschen und dielektrischen Verluste in den Resonatoren des Filters wird der Frequenzgang des Filters in der Weise degradiert, daß die erzielbare Steilheit der Filterflanken durch Abrundungseffekte begrenzt wird und die dissipative Einfügungsdämpfung im Durchlaßbereich erhöht wird. Da diese Degradation aber in erster Näherung nur von N und nicht von der Zahl M der Transmissions-Nullstellen abhängt, kann man also bei gegebener Leerlaufgüte der Resonatoren Filter mit höheren Flankensteilheiten und geringerer dissipativer Einfügungsdämpfung realisieren, wenn man M/N erhöht.Due to the ohmic and dielectric losses in the The filter's resonators become the frequency response of the filter degraded in such a way that the achievable steepness of the Filter edges is limited by rounding effects and the dissipative insertion loss in the pass band is increased. Since this degradation is a first approximation only from N and not from the number M of transmission zeros depends, so you can at given Idle quality of the resonators filters with higher Slope steepness and less dissipative Realize insertion loss by increasing M / N.

Der bei Filtern aus gekoppelten Resonatoren heute überwiegend beschrittene Weg zur Erzeugung von Transmissions-Nullstellen besteht in der Einführung von Kopplungen zwischen nicht direkt benachbarten Resonatoren ("Überkopplungen"), zusätzlich zu den direkten Kopplungen benachbarter Resonatoren. Der konventionelle Bandpaß besteht aus einer Kaskade von Resonatoren, wobei die inneren Resonatoren mindestens mit ihren beiden Nachbarn gekoppelt und die beiden äußeren Resonatoren mit den Filtertoren gekoppelt sind. Ohne zusätzliche Kopplung zwischen nicht-benachbarten Resonatoren, treten keine Transmissions-Nullstellen bei endlichen Frequenzen auf, d. h. es gilt M = 0. Überkopplungen mit geeigneter Stärke und Vorzeichen, also Kopplungen zwischen nicht-benachbarten Resonatoren, führen zu Transmissions-Nullstellen in den Sperrbereichen, wobei pro Überkopplung, je nach Lage des Koppelpfades, ein bis zwei Transmissions-Nullstellen produziert werden. Strebt man aus den oben erwähnten Gründen ein möglichst großes Verhältnis M/N sowie die höchste Freiheit bei der Wahl der Frequenzlage der einzelnen Transmissions-Nullstellen an, so führt dies zu einem Kopplungsschema, welches als "kanonische Kopplungsstruktur" bezeichnet wird und bei geradzahliger Zahl N unter Benutzung von N-2 verschiedenen Überkopplungen auf N-2 frei plazierbare Transmissions-Nullstellen führt. Für M=N-2 Nullstellen, welche symmetrisch zum Durchlaßbereich liegen, benötigt man wenigstens (N-2)/2 Überkopplungen. Die praktische Realisierung solcher Filter mit einer hohen Zahl von Überkopplungen führt in der Regel auf topologische Probleme bei der Wahl der räumlichen Anordnung der Resonatoren und Koppelelemente. Da bei der kanonischen Kopplungsstruktur erster und letzter Resonator gekoppelt und damit in unmittelbarer Nähe zueinander angeordnet werden müssen, ergibt sich bei Filtern hoher Ordnung N ein Problem bei der Realisierung genügend hoher Sperrdämpfungen.The one with filters from coupled resonators today predominantly followed path to the generation of Transmission zeros consist in the introduction of Couplings between not directly adjacent resonators ("Couplings"), in addition to the direct couplings neighboring resonators. The conventional bandpass consists of a cascade of resonators, the inner resonators at least with their two neighbors coupled and the two outer resonators with the Filter gates are coupled. Without additional coupling between non-adjacent resonators, none occur Transmission zeros at finite frequencies, d. H. M = 0 applies. Couplings with a suitable strength and sign, that is, couplings between non-adjacent ones Resonators lead to transmission zeros in the restricted areas, whereby pro Overcoupling, depending on the position of the coupling path, one or two Transmission zeros are produced. One strives as large as possible for the reasons mentioned above Ratio M / N and the greatest freedom of choice the frequency position of the individual transmission zeros , this leads to a coupling scheme, which as "canonical coupling structure" is called and at even number N using N-2 different overcouplings on N-2 freely placeable Transmission zeros leads. For M = N-2 zeros, which are symmetrical to the pass band one has at least (N-2) / 2 couplings. The practical one Realization of such filters with a high number of Couplings usually lead to topological Problems with the choice of the spatial arrangement of the Resonators and coupling elements. Because with the canonical Coupling structure coupled first and last resonator and thus arranged in close proximity to each other high-order filters result in N a problem in realizing sufficiently high Stop band attenuation.

Nach dem Stand der Technik wird zur Realisierung von Transmissions-Nullstellen alternativ zur Verwendung einer Resonatoranordnung mit Überkopplungen zwischen nicht-benachbarten Resonatoren, eine in der angelsächsischen Literatur als "Extracted-Pole-Structure,, bezeichnete Konfiguration verwendet, wobei an die Zuleitungen zum Eingangs- und/oder Ausgangstor eines Bandpassfilters ohne Transmissions-Nullstellen bei endlichen Frequenzen (M = 0), zusätzliche Resonatoren so angekoppelt werden, daß sie Transmissions-Nullstellen in den Sperrbereichen realisieren. Eine solche Anordnung ist aus DE 42 32 054 A1 bekannt, bei dem einem Mikrowellen-Keramikfilter ohne Transmissions-Nullstellen bei endlichen Frequenzen (M = 0) eine Bandsperre aus mindestens einem Koaxialresonator in Reihe geschaltet wird (Kaskade aus Bandpassfilter mit M = 0 und Bandsperre). Diese Bandsperre dient dazu, Störfrequenzen, die außerhalb des Durchlassbereichs liegen, zu eliminieren. In US 3,747.030 A wird ein etwa eine viertel Wellenlänge langer Leitungsresonator dem Ein- oder Ausgang eines Filters aus konzentrierten Elementen mit M = 0 parallelgeschaltet. Dadurch wird dem Filterzweitor eine Bandsperre in Reihe geschaltet. According to the state of the art, the realization of Transmission zeros as an alternative to using a Resonator arrangement with overcouplings between non-adjacent ones Resonators, one in the Anglo-Saxon Referred to as "Extracted Pole Structure" Configuration used, with the leads to Input and / or output gate of a bandpass filter without Transmission zeros at finite frequencies (M = 0), additional resonators are coupled so that the transmission zeros in the restricted areas realize. Such an arrangement is known from DE 42 32 054 A1 known in which a microwave ceramic filter without Transmission zeros at finite frequencies (M = 0) a bandstop from at least one coaxial resonator is connected in series (cascade of bandpass filter with M = 0 and band stop). This band stop serves Interference frequencies outside the pass band to eliminate. In US 3,747,030 A there is an approx a quarter wavelength long line resonator Input or output of a filter made of concentrated Elements with M = 0 connected in parallel. This will A filter is connected in series to a filter stop.

Aus H. Fechner "Cauerparameter-Bandpässe in Mikrostreifenleiter-Technik" Frequenz; vol. 34 (1980.03), Seiten 78-89, ist bekannt, daß man zur Erzielung der Transmissions-Nullstellen die Sperr-Resonatoren auch in das Innere der Banspassfilter-Struktur verlagern kann. So zeigt Bild 6 dieser Veröffentlichung das Layout eines 5-poligen Bandpasses in Mikrostreifenleitungs-Technik. Die 4 Transmissions-Nullstellen werden durch 4 zusätzliche Resonatoren in Form von λ/4-Stichleitungen realisiert. Bild 8 der Veröffentlichung zeigt einen 3-poligen Filter aus 3 parallel-gekoppelten Mikrostreifenleitungs-Resonatoren. Die 2 Transmissions-Nullstellen werden dadurch realisiert, daß der mittlere Resonator mit 2 Sperr-Resonatoren in Form von Stichleitungen versehen wird.
Diese bekannten Konzepte, durch Einfügung zusätzlicher Sperr-Resonatoren in die Bandpassfilter-Struktur, Transmissions-Nullstellen zu realisieren, haben gegenüber dem oben beschriebenen Konzept der "Überkopplung" den Nachteil, daß für einen Filter mit N Dämpfungsnullstellen im Durchlaßbereich mehr als N Resonatoren verwendet werden müssen.From H. Fechner "Cauerparameter bandpasses in microstrip technology"frequency; vol. 34 (1980.03), pages 78-89, it is known that the blocking resonators can also be shifted into the interior of the banspass filter structure in order to achieve the transmission zeros. Figure 6 of this publication shows the layout of a 5-pin bandpass using microstrip technology. The 4 transmission zeros are realized by 4 additional resonators in the form of λ / 4 stub lines. Figure 8 of the publication shows a 3-pin filter made of 3 parallel-coupled microstrip line resonators. The 2 transmission zeros are realized in that the middle resonator is provided with 2 blocking resonators in the form of stub lines.
These known concepts of realizing transmission zeros by inserting additional blocking resonators into the bandpass filter structure have the disadvantage over the concept of "coupling" described above that more than N resonators are used for a filter with N attenuation zeros in the pass band have to.

Für die Funktion, die Signale in einem zusammenhängenden Frequenzbereich durchzulassen und die Signale in angrenzenden Frequenzbereichen zu sperren, kann anstelle eines Bandpassfilters auch eine Bandsperre mit zwei voneinander separierten Sperrbereichen eingesetzt werden. Hierbei liegt der ausgenutzte Durchlaßbereich zwischen den beiden Sperrbereichen der Bandsperre. Aus US 5,291,161 A ist ein Filter dieser Art bekannt, welches aus einer durchgehenden Hauptleitung und an diese galvanisch angekoppelten Stichleitungen besteht und bei dem jede Stichleitung eine Transmissionsnullstelle erzeugt. Aus I. C. Hunter und J.R. Rhodes "Electronically tunable microwave bandstop filters" in IEEE Transactions on Microwave Theory and Techniques , vol. MTT-30, No. 9, September 1982, Seiten 1361 bis 1367, ist bekannt, daß zur Erzeugung von Transmissions-Nullstellen anstelle der galvanisch angekoppelten Stichleitungen auch kapazitiv angekoppelte Stichleitungen Verwendung finden können. Aus DE 24 42 618 C2 ist ebenfalls eine durchgehende Transmissionsleitung mit an diese angekoppelten Stichleitungen (Zweigleitungen) bekannt. Ein Nachteil der Verwendung solcher Filterstrukturen aus einer vom Filtereingang zum Filterausgang durchgehenden Hauptleitung mit NR angekoppelten Stichleitungen als Sperr-Resonatoren ist die Tatsache, daß die hohe Sperrdämpfung auf Frequenzbereiche endlicher Breite beschränkt bleibt und somit das Filter jenseits dieser Bereiche wieder durchläßt. Der zweite Nachteil ist, daß im ausgenutzten Durchlaßbereich zwischen den beiden Sperrbereichen die Zahl N der Dämpfungs-Nullstellen geringer als die Zahl der Resonatoren ist und damit nicht die zu einer gegebenen Resonatorzahl NR maximal erreichbare Steilheit der Filterflanken zwischen Durchlass- und Sperrbereich erzielbar ist.For the function of allowing the signals to pass through in a coherent frequency range and to block the signals in adjacent frequency ranges, a band-stop filter with two mutually separate blocking ranges can also be used instead of a bandpass filter. Here, the used pass band lies between the two band band stop band. A filter of this type is known from US Pat. No. 5,291,161 A, which consists of a continuous main line and galvanically coupled stub lines and in which each stub line generates a transmission zero point. From IC Hunter and JR Rhodes "Electronically tunable microwave bandstop filters" in IEEE Transactions on Microwave Theory and Techniques, vol. MTT-30, No. 9, September 1982, pages 1361 to 1367, it is known that capacitively coupled stub lines can also be used to generate transmission zeros instead of the galvanically coupled stub lines. From DE 24 42 618 C2 a continuous transmission line with stub lines (branch lines) coupled to it is known. A disadvantage of the use of such filter structures from a main line with N R coupled stub lines from the filter input to the filter output as blocking resonators is the fact that the high blocking attenuation remains limited to frequency ranges of finite width and thus allows the filter to pass through again beyond these ranges. The second disadvantage is that the number N of attenuation zeros is less than the number of resonators in the pass band used between the two stop areas and therefore the maximum steepness of the filter edges between pass band and stop area that can be achieved for a given resonator number N R cannot be achieved.

Mit der vorliegenden Erfindung soll dementsprechend ein Weg zur Realisierung von Bandpassfiltern aus gekoppelten Resonatoren mit bis zu M=N-1 beliebig im Sperrband plazierbaren Dämpfungspolen angegeben werden, wobei keine Überkopplungen und keine "Extracted-Pole"-Resonatoren und keine Bandsperr-Strukturen mit durchgehender Hauptleitung eingesetzt werden und damit die oben beschriebenen Nachteile dieser Konzepte vermieden werden. Accordingly, with the present invention Way to realize bandpass filters from coupled Resonators with up to M = N-1 in any stop band placeable damping poles can be specified, whereby none Couplings and no "Extracted Pole" resonators and no band-stop structures with continuous main line are used and thus the ones described above Disadvantages of these concepts can be avoided.

Die Aufgabe wird erfindungsgemäß durch die in den Patentansprüchen dargestellten Gegenstände gelöst. Gemäß der Erfindung werden Bandpassfilter-Strukturen vorgeschlagen, bei denen Sperr-Resonatoren derartig in die Struktur integriert sind, daß jeder Sperr-Resonator sowohl eine der erwünschten Transmissions-Nullstellen im Sperrbereich als auch zusammen mit der übrigen Filterstruktur eine zusätzliche Dämpfungs-Nullstelle im Durchlassbereich realisiert.The object is achieved by the in the Objects described claims solved. According to The invention uses bandpass filter structures proposed in which blocking resonators such the structure is integrated that each blocking resonator both one of the desired transmission zeros in the Restricted area as well as together with the rest Filter structure an additional damping zero in the Passband realized.

Durch diese Doppelfunktion der Sperr-Resonatoren werden im Gegensatz zu der bekannten Struktur (siehe Fechner, Frequenz 1980) für die Realisierung eines Filters mit N Polen und M=N-1 Transmissionsnullstellen nur N Resonatoren benötigt, während bei den von Fechner für Mikrostreifenleitungs-Filter vorgeschlagenen Strukturen N+M=2N-1 Resonatoren in Form von Stichleitungen benötigt werden.This double function of the blocking resonators in contrast to the known structure (see Fechner, Frequency 1980) for the implementation of a filter with N Poland and M = N-1 transmission zeros only N Resonators needed, while those from Fechner for Microstrip line filters proposed structures N + M = 2N-1 resonators in the form of stub lines are required become.

Diese erfindungsgemäßen Bandpassfilter-Strukturen sind durch folgende Merkmale gekennzeichnet:

  • (a) Die Bandpassfilter werden aus einem, unten näher beschriebenen impedanz-symmetrischen Filterglied mit N = 2m+1 (m = natürliche Zahl) Polen und N-1 Transmissions-Nullstellen, oder aus einer Kaskade solcher impedanz-symmetrischen Filterglieder, oder einer Kaskade aus weiter unter beschriebenen impedanz-unsymmetrischen Filtergliedern mit jeweils N= 2 Polen und M =2 Transmissions-Nullstellen gebildet.
  • (b) Ein impedanz-symmetrisches Filterglied mit N = 3 Polen und M= 2 Transmissions-Nullstellen besteht aus einem durch zwei Diskontinuitäten begrenztem Leitungsstück, als Hauptresonator bezeichnet, an das in der Mitte ein Paar von Sperr-Resonatoren angekoppelt ist, derart, daß aufgrund der longitudinalen Feldverteilung auf dem Hauptresonator die Kopplung zu den Sperr-Resonatoren bei der Resonanzfrequenz des Hauptresonators (Mittenfrequenz) verschwindet, jedoch bei davon abweichenden Frequenzen einen endlichen Wert annimmt. Die Länge des Hauptresonators wird so gewählt, daß sie etwa gleich der halben Leitungswellenlänge bei der Mittenfrequenz des Bandpassfilters entspricht. Die Sperrfrequenz des einen Sperr-Resonators wird kleiner und die des anderen Sperr-Resonators größer als die Mittenfrequenz gewählt und dadurch erzeugt jeder der beiden Sperr-Resonatoren eine Transmissions-Nullstelle und durch Zusammenwirken mit dem Hauptresonator einen zusätzlichen Pol. Dient ein einzelnes impedanz-symmetrisches Filterglied als Bandpassfilter, so wird ein Ende des Hauptresonators mit dem Filtereingang und das andere Ende mit dem Filterausgang elektrisch, galvanisch oder magnetisch verkoppelt. Wird das Bandpassfilter aus einer Kaskade mehrerer impedanz-symmetrischer Filterglieder aufgebaut, werden die nicht an das Eingangs- oder Ausgangstor angekoppelten benachbarten Hauptresonator-Enden elektrisch, galvanisch oder magnetisch verkoppelt.
  • (c) Ein impedanz-symmetrisches Filterglied mit N = 2m+1 Polen (m = natürliche Zahl größer 1) und M= N-1 = 2m Transmissions-Nullstellen besteht aus einem durch zwei Diskontinuitäten begrenztem Leitungsstück, als Hauptresonator bezeichnet, an den m Paare von Sperr-Resonatoren im gegenseitigen Abstand von ca. einer halben Mittenfrequenz-Wellenlänge angekoppelt sind, derart daß aufgrund der longitudinalen Feldverteilung auf dem Hauptresonator die Kopplung zu den Sperr-Resonatoren bei der Resonanzfrequenz des Hauptresonators (Mittenfrequenz) verschwindet, jedoch bei davon abweichenden Frequenzen einen endlichen Wert annimmt. Die Länge des Hauptresonators wird so gewählt, daß sie etwa gleich dem m-fachen der halben Leitungswellenlänge bei der Mittenfrequenz entspricht und der Abstand der äußeren Sperr-Resonator-Paare von den Enden des Hauptresonators beträgt ca. eine viertel Leitungswellenlänge. Die Sperrfrequenzen der beiden Sperr-Resonatoren eines jeden der m Sperr-Resonator-Paare werden so gewählt, daß eine kleiner und die andere größer als die Mittenfrequenz ist, und dadurch erzeugt jeder der beiden Sperr-Resonatoren eine Transmissions-Nullstelle und durch Zusammenwirken mit dem Hauptresonator einen zusätzlichen Pol. Dient ein einzelnes impedanz-symmetrisches Filterglied als Bandpassfilter, so wird ein Ende des Hauptresonators mit dem Filtereingang verkoppelt und das andere Ende mit dem Filterausgang elektrisch, galvanisch oder magnetisch verkoppelt. Wird das Bandpassfilter aus einer Kaskade mehrerer impedanz-symmetrischer Filterglieder aufgebaut, werden die nicht an das Eingangs- oder Ausgangstor angekoppelten benachbarten Hauptresonator-Enden elektrisch, galvanisch oder magnetisch verkoppelt.
  • (d) Zusätzlich zu den impedanz-symmetrischen Filtergliedern können zum Aufbau eines Bandpassfilters aus einer Kaskade mehrerer Filterglieder impedanz-unsymmetrische Filterglieder mit jeweils N= 2 Polen und M = 2 Transmissions-Nullstellen Verwendung finden. Die impedanz-unsymmetrischen Filterglieder bestehen aus einem Hauptresonator dessen Länge ca. einer viertel Leitungswellenlänge bei der Mittenfrequenz des Bandpassfilters entspricht, bei dem das eine Ende so an das benachbarte Filterglied angekoppelt ist, daß dieses Leitungsende hochohmig abgeschlossen ist (Maximum der elektrischen Feldstärke) und das andere Ende so an das benachbarte Filterglied oder das Einoder Ausgangstor des Bandpasses angekoppelt ist, daß das Leitungsende niederohmig abgeschlossen ist (Strommaximum am Leitungsende), und bei dem am niederohmigen Ende des Hauptresonators ein Paar von Sperr-Resonatoren elektrisch oder galvanisch angekoppelt ist.
    • These bandpass filter structures according to the invention are characterized by the following features:
  • (a) The bandpass filters are made from an impedance-symmetrical filter element with N = 2m + 1 (m = natural number) poles and N-1 transmission zeros, described below, or from a cascade of such impedance-symmetrical filter elements, or a cascade formed from filter elements described below under impedance-unsymmetrical, each with N = 2 poles and M = 2 transmission zeros.
  • (b) An impedance-symmetrical filter element with N = 3 poles and M = 2 transmission zeros consists of a line section, delimited by two discontinuities, called the main resonator, to which a pair of blocking resonators is coupled in the middle, in such a way that due to the longitudinal field distribution on the main resonator, the coupling to the blocking resonators at the resonance frequency of the main resonator (center frequency) disappears, but assumes a finite value at frequencies that deviate therefrom. The length of the main resonator is chosen so that it corresponds approximately to half the line wavelength at the center frequency of the bandpass filter. The blocking frequency of the one blocking resonator is chosen to be lower and that of the other blocking resonator is chosen to be larger than the center frequency and as a result each of the two blocking resonators generates a transmission zero and an additional pole by interaction with the main resonator. If a single impedance-symmetrical filter element serves as a bandpass filter, one end of the main resonator is coupled electrically, galvanically or magnetically to the filter input and the other end to the filter output. If the bandpass filter is constructed from a cascade of several impedance-symmetrical filter elements, the adjacent main resonator ends that are not coupled to the input or output port are electrically, galvanically or magnetically coupled.
  • (c) An impedance-symmetrical filter element with N = 2m + 1 poles (m = natural number greater than 1) and M = N-1 = 2m transmission zeros consists of a line section delimited by two discontinuities, referred to as the main resonator, at the m Pairs of blocking resonators are coupled at a mutual distance of approximately half a center frequency wavelength, such that due to the longitudinal field distribution on the main resonator, the coupling to the blocking resonators at the resonance frequency of the main resonator (center frequency) disappears, but at frequencies that deviate therefrom takes on a finite value. The length of the main resonator is chosen so that it corresponds approximately to m times half the line wavelength at the center frequency and the distance of the outer blocking resonator pairs from the ends of the main resonator is approximately a quarter line wavelength. The blocking frequencies of the two blocking resonators of each of the m blocking resonator pairs are chosen such that one is smaller and the other is larger than the center frequency, and thereby each of the two blocking resonators generates a transmission zero and by cooperation with the Main resonator an additional pole. If a single impedance-symmetrical filter element serves as a bandpass filter, one end of the main resonator is coupled to the filter input and the other end is electrically, galvanically or magnetically coupled to the filter output. If the bandpass filter is constructed from a cascade of several impedance-symmetrical filter elements, the adjacent main resonator ends that are not coupled to the input or output port are electrically, galvanically or magnetically coupled.
  • (d) In addition to the impedance-symmetrical filter elements, impedance-asymmetrical filter elements, each with N = 2 poles and M = 2 transmission zeros, can be used to construct a bandpass filter from a cascade of several filter elements. The impedance-unbalanced filter elements consist of a main resonator whose length corresponds to approximately a quarter of the line wavelength at the center frequency of the bandpass filter, in which one end is coupled to the adjacent filter element in such a way that this line end is terminated with high resistance (maximum of the electric field strength) and that the other end is coupled to the adjacent filter element or the input or output port of the bandpass in such a way that the line end is terminated with a low resistance (current maximum at the line end), and in which a pair of blocking resonators is electrically or galvanically coupled to the low-resistance end of the main resonator.

    • Die oben getroffene Unterscheidung zwischen impedanz-symmetrischen und impedanz-unsymmetrischen Filtergliedern ist so zu verstehen, daß bei einem impedanz-symmetrischen Filterglied bei Beschaltung von Ein- und Ausgangstor mit dem gleichen Abschlußwiderstand, die Maximalwerte des Leistungs-Übertragungsfaktors bei vernachlässigbaren Verluste den Wert Eins erreichen, während beim übertragungs-unsymmetrischen Filterglied vollständige Leistungsübertragung nur für stark unsymmetrische Tor-Widerstände erreichbar ist.The above distinction between impedance-symmetrical and impedance-unbalanced filter elements is to be understood that with an impedance-symmetrical Filter element when connecting the input and output gates with the same termination resistance, the maximum values of the Power transmission factor with negligible Losses reach one, while at transmission unbalanced filter element complete Power transmission only for highly asymmetrical gate resistors is achievable.

    Die weitere Erläuterung der Erfindung erfolgt anhand des in den Zeichnungen 1 bis 4 dargestellten Grundprinzips und der in den Zeichnungen 5 bis 12 dargestellten Ausführungsbeispiele.The further explanation of the invention is based on the basic principle shown in the drawings 1 to 4 and that shown in the drawings 5 to 12 Embodiments.

    Figur 1e zeigt auf schematische Weise den prinzipiellen Aufbau eines erfindungsgemäßen impedanz-symmetrischen Filterglieds mit N = 3 Polen und M =2 Transmissions-Nullstellen und die Figuren 1a bis 1d zeigen auf schematische Weise Strukturen, welche dem Stand der Technik entsprechen und daher nur zur schrittweisen Erläuterung des Grundprinzips der erfindungsgemäßen Struktur nach Figur 1e dienen. Figure 1e shows the principle in a schematic manner Structure of an impedance-symmetrical according to the invention Filter element with N = 3 poles and M = 2 transmission zeros and Figures 1a to 1d show schematic way structures, which the state of the Correspond to technology and therefore only for gradual Explanation of the basic principle of the invention Serve structure according to Figure 1e.

    Fig. 1a zeigt symbolhaft eine homogene Hochfrequenzleitung 1, bei der diese Leitung als metallische TEM-Leitung z. B. als eine Koaxialleitung, als eine planare Leitung wie z. B. eine Mikrostreifenleitung oder Streifenleitung oder Koplanarleitung, oder als Hohlleiter oder als dielektrische Leitung ausgeführt sein kann. Bei Vernachlässigung der Dissipation wird der Frequenzgang des Leistungsübertragungsfaktors 2, also die Frequenzabhängigkeit des Verhältnisses der am reflexionsfrei abgeschlossenem Tor 2 heraustretenden Leistung P2 zu der am Tor 1 einfallenden Leistung Peinf, im betrachteten Betriebsfrequenzbereich der Leitung unabhängig von der Frequenz gleich Eins.Fig. 1a symbolically shows a homogeneous high-frequency line 1, in which this line as a metallic TEM line z. B. as a coaxial line, as a planar line such. B. a microstrip line or strip line or coplanar line, or as a waveguide or as a dielectric line. Neglecting the dissipation of the frequency response of the power transfer factor 2, so the frequency dependence of the ratio of the reflection-free completed Tor 2 emergent power P 2 is simp to the incident at port 1 output P, in the considered operating frequency range of the line regardless of the frequency equal to one.

    Figur 1b zeigt schematisch eine gegenüber Figur 1a abgeänderte Struktur, bei der zwei Diskontinuitäten 3 symmetrisch in den Leitungszug eingeführt sind. Diese Diskontinuitäten definieren ein Leitungsstück endlicher Länge a, auf dem elektromagnetische Eigenschwingungen bei denjenigen Frequenzen auftreten, bei denen die Länge a einem ganzzahligen Vielfachen einer halben Leitungswellenlänge entspricht und diese Eigenschwingungen sind durch stehende Wellen mit Knoten und Bäuchen der elektrischen und magnetischen Feldstärke entlang der Leitung gekennzeichnet, wobei in der Symmetrieebene 4 bei der Resonanzfrequenz ein Knoten der elektrischen oder magnetischen Feldstärke existiert. Die so entstandene Struktur stellt einen, nach dem Stand der Technik wohlbekannten 1-poligen Bandpass dar, der durch einen Frequenzgang des Leistungsübertragungsfaktors 5 mit einem Maximum P2/Peinf= 1 (Dämpfungs-Nullstelle) bei einer Frequenz f0 gekennzeichnet ist. Die das Leitungsstück begrenzenden Diskontinuitäten können technisch z. B. in Form von Leitungsunterbrechungen oder in Form metallischer Blenden ausgebildet sein, und es ist nach dem Stand der Technik ebenfalls wohlbekannt, daß über die Stärke der Kopplung zwischen den Zuleitungen und den Enden des als Resonator dienenden Leitungsstücks, die Frequenz-Bandbreite Δf der Transmissionskurve verändert werden kann.FIG. 1b schematically shows a structure modified compared to FIG. 1a, in which two discontinuities 3 are inserted symmetrically into the cable run. These discontinuities define a line section of finite length a, on which electromagnetic natural vibrations occur at those frequencies at which the length a corresponds to an integer multiple of half the line wavelength, and these natural vibrations are characterized by standing waves with nodes and antinodes of the electrical and magnetic field strength along the line , with a node of the electric or magnetic field strength existing in the plane of symmetry 4 at the resonance frequency. The structure thus created represents a 1-pole bandpass, which is well known in the prior art and is characterized by a frequency response of the power transmission factor 5 with a maximum P 2 / P einf = 1 (damping zero) at a frequency f 0 . The discontinuities limiting the line piece can technically z. B. in the form of line interruptions or in the form of metallic diaphragms, and it is also well known in the art that about the strength of the coupling between the leads and the ends of the line section serving as a resonator, the frequency bandwidth Δf of the transmission curve can be changed.

    Figur 1c zeigt eine gegenüber Figur 1a abgeänderte Struktur, bei dem ein Resonanzkreis 6 ("Sperr-Resonator") an die Leitung angekoppelt ist, so daß der Frequenzgang des Leistungsübertragungsfaktors 7 eine Transmissions-Nullstelle bei der Frequenz fs aufweist. Diese Struktur stellt den nach dem Stand der Technik wohlbekannten Aufbau einer einpoligen Bandsperre ("Notch-Filter") dar.Figure 1c shows a structure modified from Figure 1a, in which a resonance circuit 6 ("blocking resonator") is coupled to the line, so that the frequency response of the power transmission factor 7 has a transmission zero at the frequency f s . This structure represents the construction of a unipolar bandstop ("notch filter"), which is well known in the prior art.

    Figur 1d zeigt eine gegenüber Figur 1c abgeänderte Struktur dar, bei der anstelle eines Sperr-Resonators zwei Sperr-Resonatoren 8 mit unterschiedlichen Resonanzfrequenzen angekoppelt sind und zu zwei Transmissions-Nullstellen bei fs1 und fs2 führen.FIG. 1d shows a structure modified from FIG. 1c, in which instead of a blocking resonator two blocking resonators 8 with different resonance frequencies are coupled and lead to two transmission zeros at f s1 and f s2 .

    Ein wesentlicher Aspekt der Erfindung besteht nun darin, aus einer Kombination der Struktur nach Figur 1b und des Sperr-Resonator-Paars von Figur 1d, die Struktur nach Figur 1e zu bilden. Das Leitungsstück endlicher Länge bildet einen Resonator, hier als Hauptresonator bezeichnet, welcher in der Mitte einen Knoten des elektrischen oder magnetischen Felds besitzt. Ein wesentlicher Aspekt der Erfindung ist die Wahl der Kopplung zwischen den Sperr-Resonatoren und dem Hauptresonator in der Weise, daß bei der Frequenz f0 diese Kopplung verschwindet, wobei dies z. B. dadurch erreicht wird, daß bei Vorliegen eines Knoten des elektrischen Felds eine elektrische Kopplung und bei Vorliegen eines Knotens des magnetischen Felds, eine magnetische Kopplung zwischen Hauptresonator und den Sperr-Resonatoren gewählt wird. Durch diese Maßnahme wird einerseits die Resonanz des Hauptresonators bei der Frequenz f0 nicht durch das Sperr-Resonator-Paar gestört und andererseits erhält man aufgrund der Kopplung zwischen Sperr-Resonator-Paar und Hauptresonator für Frequenzen verschieden von f0 zwei zusätzliche Eigenschwingungen. In dieser erfindungsgemäßen Struktur übernehmen die beiden Sperr-Resonatoren damit eine Doppelfunktion, indem sie einerseits - wie in der Struktur nach Figur 1d - zwei Transmissions-Nullstellen realisieren und andererseits zusammen mit dem Leitungsstück insgesamt 3 Eigenschwingungen (3 Pole) produzieren. Der Frequenzgang 10 der Struktur nach Fig. 1e ist also bei geeigneter Wahl der Resonanzfrequenzen und Koppelstärken durch drei Transmissionsmaxima (Dämpfungs-Nullstellen) bei f1, f2 und f3 sowie zwei Transmissions-Nullstellen bei fs1 und fs2 gekennzeichnet. Bei diesem Filterglied zur Realisierung von 3 Polen und zwei Transmissions-Nullstellen wird die Frequenzlage der Transmissions-Nullstellen durch die Resonanzfrequenzen der Sperr-Resonatoren bestimmt und die Frequenzlage des mittleren Transmissions-Maximums durch die Länge des Hauptresonators. Die Lage der beiden äußeren Transmissionsmaxima kann durch die Koppelstärke zwischen Hauptresonator und Sperr-Resonatoren verändert werden, wobei bei einer Vergrößerung der Kopplung, diese Frequenzen sich in Richtung auf die mittlere Frequenz verschieben.An essential aspect of the invention consists in forming the structure according to FIG. 1e from a combination of the structure according to FIG. 1b and the blocking resonator pair from FIG. 1d. The line section of finite length forms a resonator, here referred to as the main resonator, which has a node of the electric or magnetic field in the middle. An essential aspect of the invention is the choice of the coupling between the blocking resonators and the main resonator in such a way that this coupling disappears at the frequency f 0 . B. is achieved in that in the presence of a node of the electric field, an electrical coupling and in the presence of a node of the magnetic field, a magnetic coupling between the main resonator and the blocking resonators is selected. This measure on the one hand does not disturb the resonance of the main resonator at the frequency f 0 by the blocking-resonator pair, and on the other hand, due to the coupling between the blocking-resonator pair and the main resonator, two additional natural oscillations are obtained for frequencies different from f 0 . In this structure according to the invention, the two blocking resonators thus assume a double function in that on the one hand they realize two transmission zeros - as in the structure according to FIG. 1d - and on the other hand produce a total of 3 natural oscillations (3 poles) together with the line section. The frequency response 10 of the structure according to FIG. 1e is thus characterized by a suitable choice of the resonance frequencies and coupling strengths by three transmission maxima (damping zeros) at f 1 , f 2 and f 3 and two transmission zeros at f s1 and f s2 . In this filter element for realizing 3 poles and two transmission zeros, the frequency position of the transmission zeros is determined by the resonance frequencies of the blocking resonators and the frequency position of the average transmission maximum by the length of the main resonator. The position of the two outer transmission maxima can be changed by the coupling strength between the main resonator and blocking resonators, with an increase in the coupling, these frequencies shifting towards the middle frequency.

    Ein weiterer wesentlicher Aspekt der Erfindung ist die in Figur 2a bis 2c dargestellte Verallgemeinerung des Prinzips nach Figur 1e zur Realisierung von Filtergliedern mit M = 2m Transmissions-Nullstellen und N = M+1 =2m+1 Polen. In Figur 2a ist nochmals der Fall m=1 entsprechend Figur 1e dargestellt. Falls das Leitungsstück bei der Mittenfrequenz eine halbe Wellenlänge lang ist ("Mittenfrequenz-Leitungswellenlänge"), hängt die Art der Kopplung zwischen den Sperr-Resonatoren und dem Hauptresonator davon ab, ob sich an den Enden des Leitungsstücks die Betrags-Maxima des elektrischen oder magnetischen Felds befinden. Im Falle elektrischer Feldmaxima an den Enden besitzt das elektrische Feld bei der Frequenz f0 einen Knoten in der Symmetrieebene und damit müssen nach obigen Designregeln die beiden Sperr-Resonatoren elektrisch gekoppelt werden, während im Falle magnetischer Feldmaxima an den Enden, wegen des Knoten des magnetischen Felds, eine magnetische Kopplung vorliegen muß. Um im Falle magnetischer Feldmaxima an den Enden trotzdem eine magnetische Kopplung zwischen Sperr-Resonatoren und Haupresonator verwenden zu können, muß die Länge des Leitungsstücks anstelle einer halben Mittenfrequenz-Leitungswellenlänge gleich einer vollen Wellenlänge entsprechen.Another essential aspect of the invention is the generalization of the principle according to FIG. 1e shown in FIGS. 2a to 2c for implementing filter elements with M = 2m transmission zeros and N = M + 1 = 2m + 1 poles. In FIG. 2a the case m = 1 is shown again in accordance with FIG. 1e. If the line section at the center frequency is half a wavelength long ("center frequency line wavelength"), the type of coupling between the blocking resonators and the main resonator depends on whether the magnitude maxima of the electrical or magnetic are at the ends of the line section Field. In the case of electrical field maxima at the ends, the electrical field at frequency f 0 has a node in the plane of symmetry and thus the two blocking resonators must be electrically coupled according to the above design rules, while in the case of magnetic field maxima at the ends, because of the node of the magnetic one Field, a magnetic coupling must be present. In order to be able to use a magnetic coupling between the blocking resonators and the main resonator in the case of magnetic field maxima at the ends, the length of the line section must correspond to a full wavelength instead of half the center frequency line wavelength.

    Figur 2b zeigt die erfindungsgemäße Verallgemeinerung für m=2, also N=5 Pole und M=4 Transmissions-Nullstellen, wobei zwei Paare von Sperr-Resonatoren im gegenseitigen Abstand von etwa einer halben Leitungswellenlänge verwendet werden.FIG. 2b shows the generalization for m = 2, i.e. N = 5 poles and M = 4 transmission zeros, being two pairs of blocking resonators in mutual Distance of approximately half a line wavelength be used.

    Figur 2c zeigt die erfindungsgemäße Erweiterung auf ein Filterglied mit N =7 Polen und M =N -1=6 Transmissions-Nullsteilen.FIG. 2c shows the extension according to the invention Filter element with N = 7 poles and M = N -1 = 6 transmission zero parts.

    Die Vergrößerung der Polzahl N eines Filterglieds nach dem in den Abbildungen 2a bis 2c gezeigtem Prinzip wird durch die Frequenzlage höherer unerwünschter Eigenschwingungen des Hauptresonators begrenzt, wobei die Verlängerung des Hauptresonators zur Erhöhung der Polzahl, die Eigenresonanzen des Hauptresonators im Frequenzbereich immer weiter zusammenrückt. Um trotz dieser Begrenzung Filter höherer Polzahl realisieren zu können, werden in einer weiteren Ausgestaltung der Erfindung zwei alternative Wege beschritten, nämlich eine Kaskadierung von impedanz-symmetrischen Filtergliedern nach Fig. 2a bis 2c und die Einführung impedanzunsymmetrischer Filterglieder mit zwei Polen und zwei Transmissions-Nullstellen pro Filterglied.The increase in the number of poles N of a filter element after the principle shown in Figures 2a to 2c due to the frequency situation higher undesirable Limits natural vibrations of the main resonator, the Extension of the main resonator to increase the Number of poles, the natural resonances of the main resonator in the Frequency range keeps moving closer together. To despite this limitation to realize filters with a higher number of poles can be in a further embodiment of the Invention followed two alternative paths, namely one Cascading of impedance-symmetrical filter elements 2a to 2c and the introduction of impedance unbalanced Filter elements with two poles and two Transmission zeros per filter element.

    Figur 3 zeigt, wie aus einer Kaskade von Q Filtergliedern mit jeweils Ng Polen und Mg=N-1 Transmissions-Nullstellen ein Filter mit der Polzahl N=NgxQ und M=N-Q Transmissions-Nullstellen gebildet wird. Beispielhaft sind der Fall eines 9-poligen (9-kreisigen) Filters mit 6 Transmissions-Nullstellen aus 3 Filtergliedern mit Ng=3 sowie eines 10-poligen Filters mit 8 Transmissions-Nullstellen aus 3 Filtergliedern mit Ng=5 dargestellt.FIG. 3 shows how a filter with the number of poles N = N g xQ and M = NQ transmission zeros is formed from a cascade of Q filter elements, each with N g poles and M g = N-1 transmission zeros. The case of a 9-pole (9-circuit) filter with 6 transmission zeros from 3 filter elements with Ng = 3 and a 10-pole filter with 8 transmission zeros from 3 filter elements with Ng = 5 are shown as examples.

    Ein impedanz-unsymmetrisches Filterglied wird erfindungsgemäß dadurch realisiert, daß ein impedanzsymmetrisches Filterglied mit einem Sperr-Resonatoren-Paar nach Fig. 1e modifiziert wird, wobei eine der beiden Diskontinuitäten in die Nähe der Stelle gebracht wird, an der das Sperr-Resonator-Paar angekoppelt ist. Damit entsteht die in Fig. 4a gezeigte T-förmige Struktur mit einer von der Ankoppelstelle des Sperr-Resonator-Paars ca. um eine viertel Mittenfrequenz-Leitungswellenlänge entfernten Diskontinuität 2 ("hochohmiges Ende") und einer zweiten Diskontinuität ("niederohmiges Ende", 3), welche sich nahe an der Koppelstelle des Sperr-Resonator-Paars befindet.An impedance-unbalanced filter element becomes realized according to the invention in that an impedance-symmetrical Filter element with a pair of blocking resonators 1e is modified, one of the two Discontinuities are brought close to the site which is coupled to the blocking resonator pair. In order to the T-shaped structure shown in FIG one from the coupling point of the blocking resonator pair about a quarter of the center frequency line wavelength removed discontinuity 2 ("high impedance end") and a second discontinuity ("low-resistance end", 3), which is close to the coupling point of the blocking-resonator pair located.

    Um die Impedanz-Unsymmetrie zu kompensieren, wird in einer Kaskade aus impedanz-unsymmetrischen Filtergliedern mindestens ein impedanz-symmetrisches Glied hinzugefügt. Hierbei kann sich, wie in Fig. 4b gezeigt, das impedanzsymmetrische Glied 5 an einem Ende der Kaskade befinden, oder es kann zentral (siehe Fig. 4c) eingefügt werden.In order to compensate for the impedance asymmetry, in a cascade of impedance-unbalanced filter elements added at least one impedance-symmetrical element. Here, as shown in FIG. 4b, the impedance-symmetrical Link 5 located at one end of the cascade or it can be inserted centrally (see Fig. 4c).

    Für die in den Abbildungen 1e, 2a bis 2c, 3 und 4 prinzipiell schematisch dargestellten erfindungsgemäßen Filterstrukturen ergibt sich eine sehr große Zahl von technischen Ausgestaltungsmöglichkeiten, die sich u. a. unterscheiden hinsichtlich

  • a) des Leitungstyps aus dem der Hauptresonator aufgebaut ist,
  • b) der Bauform der Sperr-Resonatoren
  • c) der Kopplungsart zwischen Sperr-Resonator und Hauptresonator
  • d) der Gestaltung der Diskontinuitäten (Kopplung) zwischen Hauptresonatoren in Kaskade und dem Hauptresonatoren und Toren.
    • For the filter structures according to the invention, shown schematically in principle in FIGS. 1e, 2a to 2c, 3 and 4, there is a very large number of technical design options which differ, inter alia, with regard to
  • a) the type of line from which the main resonator is constructed,
  • b) the design of the blocking resonators
  • c) the type of coupling between the blocking resonator and the main resonator
  • d) the design of the discontinuities (coupling) between the main resonators in cascade and the main resonators and gates.

    • Fig. 5 zeigt exemplarisch die Realisierung eines 7-poligen Filters mit 6 Transmissions-Nullstellen in Form eines einzelnen Filterglieds nach dem in Fig. 2c gezeigten Prinzip in Koaxialleitungstechnik. Der Hauptresonator 1 hat einen rechteckförmigen Außen- und Innenleiter und eine Länge gleich dem 1,5-fachen der Mittenfrequenz-Wellenlänge. Die das Leitungsstück begrenzenden Diskontinuitäten sind in Form kapazitiver Koppler ausgebildet. Die Sperr-Resonatoren 2 sind als am Ende kurzgeschlossenen Koaxilleitungsstücke einer Länge von ca. einer viertel Leitungswellenlänge realisiert, welche kaspazitiv an den Hauptresonator gekoppelt sind.Fig. 5 shows an example of the implementation of a 7-pin Filters with 6 transmission zeros in the form a single filter element according to that in Fig. 2c principle shown in coaxial line technology. The Main resonator 1 has a rectangular exterior and Inner conductor and a length equal to 1.5 times that Center frequency wavelength. The the line piece limiting discontinuities are more capacitive in form Coupler trained. The blocking resonators 2 are as on End of short-circuited coaxial line pieces of one length of a quarter of a line wavelength, which are capacitively coupled to the main resonator.

    Fig. 6 zeigt eine Modifikation der Struktur nach Fig. 5, indem nun die Sperr-Resonatoren 2 galvanisch mit dem Innenleiter des Hauptresonators verbunden sind, aber am Ende kapazitiv belastet sind.Fig. 6 shows a modification of the structure of Fig. 5, by now the blocking resonators 2 galvanically with the Inner conductors of the main resonator are connected, but on Are capacitively loaded at the end.

    Fig. 7 zeigt eine Struktur aus zwei impedanz-unsymmetrischen Filtergliedern und einem impedanz-symmetrischen Glied , bei der man 9 Pole und 8 Transmissions-Nullstellen erhält.Fig. 7 shows a structure of two impedance-unbalanced Filter elements and an impedance-symmetrical Link with 9 poles and 8 Receives transmission zeros.

    Fig. 8 zeigt ein Filter aus einem impedanz-symmetrischen Filterglied mit 5 Polen und 4 Transmissions-Nullstellen, welches auf der Basis von Rechteckhohlleitungen für den H10-Wellentyp realisiert ist. Der Hauptresonator 1 besteht aus einer an beiden Enden kurzgeschlossenen Rechteckhohlleitung, welche bei der Mittenfrequenz eine Länge entsprechend einer Hohlleiter-Wellenlänge hat. Die 4 Sperr-Resonatoren 2 sind in Form von kurzgeschlossenen 1/4-Hohlleiterstücken realisiert. Die Ankopplung zu den Toren kann z. B. über einen Koaxial-Übergang 3 erfolgen.8 shows a filter made of an impedance-symmetrical Filter element with 5 poles and 4 transmission zeros, which is based on rectangular hollow pipes for the H10 shaft type is realized. The main resonator 1 consists of a short-circuited at both ends Rectangular waveguide, which at the center frequency Has a length corresponding to a waveguide wavelength. The 4 blocking resonators 2 are in the form of short-circuited 1/4 waveguide pieces realized. The coupling to the Gates can e.g. B. via a coaxial transition 3.

    Fig. 9 zeigt beispielhaft eine Realisierung mit dielektrischen Resonatoren im Fall eines Filters aus zwei impedanz-symmetrischen Filtergliedern, wobei jedes Filterglied drei Pole und zwei Transmissions-Nullstellen produziert und somit das Bandpassfilter insgesamt 6 Pole und 4 Transmissions-Nullstellen aufweist. Die aus geeignetem dielektrischen Material, also Material mit einer möglichst hohen Dielektrizitätszahl, einem niedrigen Verlustwinkel und einem geringen Temperaturkoeffizienten (z.B. Bariumtitanat Zirkonat) hergestellten Hauptresonatoren 1 und Sperr-Resonatoren 2 sind über Abstandshalter 3, z. B. aus Quarzmaterial; zur Vermeidung zu starker ohmscher Verluste in einer genügenden Entfernung vom Boden des metallischen Gehäuses 5 positioniert. Die Abmessung des Hauptresonators wird so gewählt, daß dieser bei f0 eine Eigenresonanz mit der in Fig. 9b gezeigten Feldverteilung aufweist, und die Abmessung der Sperr-Resonatoren werden so gewählt, daß diese bei den 4 Sperrfrequenzen f1 bis f4 resonieren und dabei eine Feldverteilung entsprechend Fig. 9c aufweisen. Aufgrund der räumlichen Feldverteilung des Hauptresonators koppelt dieser bei f0 nicht an die Resonanzfelder der Sperr-Resonatoren. Für von f0 verschiedene Frequenzen erhält man jedoch eine Kopplung zwischen dem Hauptresonator und den Sperr-Resonatoren mit dem Resultat, daß zusätzlich 4 Eigenresonanzen entstehen. Die Ankopplung an die Tore kann z. B. über Leiterschleifen 4 erfolgen.9 shows an example of an implementation with dielectric resonators in the case of a filter comprising two impedance-symmetrical filter elements, each filter element producing three poles and two transmission zeros and thus the bandpass filter having a total of 6 poles and 4 transmission zeros. The main resonators 1 and blocking resonators 2, which are made from a suitable dielectric material, that is to say material with the highest possible dielectric constant, a low loss angle and a low temperature coefficient (for example barium titanate zirconate), are connected via spacers 3, e.g. B. made of quartz material; positioned to avoid excessive ohmic losses at a sufficient distance from the bottom of the metallic housing 5. The dimension of the main resonator is chosen so that it has a natural resonance at f 0 with the field distribution shown in FIG. 9b, and the dimension of the blocking resonators are chosen so that they resonate at the 4 blocking frequencies f 1 to f 4 and thereby have a field distribution corresponding to FIG. 9c. Because of the spatial field distribution of the main resonator, it does not couple to the resonance fields of the blocking resonators at f 0 . For frequencies different from f 0 , however, a coupling is obtained between the main resonator and the blocking resonators, with the result that an additional 4 natural resonances arise. The coupling to the gates can e.g. B. via conductor loops 4.

    Fig. 10 zeigt beispielhaft eine weitere mögliche Bauform eines Filterglieds aus dielektrischem Material. Der Hauptresonator 5 besteht aus einem dielektrischen Quader der Länge a, welche etwa gleich einer Wellenlänge der Oberflächenwelle auf dem dielektrischen Quader entspricht. Dadurch erhält man auf dem Hauptresonator eine Feldverteilung entsprechend Fig. 10b. Die 4 Sperr-Resonatoren 1 bis 4 bestehen ebenfalls aus dielektrischen Quadern, deren individuelle Längen b1 bis b4 die Frequenzlage der 4 Transmissions-Nullstellen beeinflussen. Das gesamte Gebilde aus dielektrischem Hauptresonator und 4 dielektrischen Sperr-Resonatoren realisiert 5 Eigenschwingungen. Die Frequenzlage der Pole kann über die Koppelstärke zwischen Haupt- und Sperr-Resonatoren verändert werden. Zur Veränderung dieser Koppelstärke dienen die mit Luft oder einem dielektrischen Material relativ geringer Dielektrizitätszahl gefüllten "Lücken" zwischen den Resomatoren mit den Weiten h1 bis h4.10 shows an example of a further possible design of a filter element made of dielectric material. The main resonator 5 consists of a dielectric cuboid of length a, which corresponds approximately to a wavelength of the surface wave on the dielectric cuboid. A field distribution corresponding to FIG. 10b is thereby obtained on the main resonator. The 4 blocking resonators 1 to 4 also consist of dielectric cuboids, the individual lengths b1 to b4 of which influence the frequency position of the 4 transmission zeros. The entire structure of the main dielectric resonator and 4 dielectric blocking resonators realizes 5 natural vibrations. The frequency position of the poles can be changed via the coupling strength between the main and blocking resonators. The "gaps" between the resonators with the widths h 1 to h 4 filled with air or a dielectric material with a relatively low dielectric constant serve to change this coupling strength.

    Das erfindungsgemäße Prinzip kann auch auf planare Resonatorstrukturen, wie z. B. Mikrostreifenleitungsstrukturen angewendet werden, wobei auch Mikrostreifenleitungs-Strukturen aus Hochtemperatur-Supraleitern von Interesse sind, da diese trotz eines enormen Miniaturisierungsgrads über eine hohe Leerlaufgüte verfügen.The principle according to the invention can also apply to planar ones Resonator structures, such as B. microstrip line structures are used, including microstrip line structures from high temperature superconductors from Are interested as this despite an enormous Degree of miniaturization over a high idling quality feature.

    In Fig. 11 wird die Realisierung eines erfindungsgemäßen impedanz-unsymmetrischen Filterglieds in Mikrostreifenleitungs-Technologie erläutert. In Fig. 11a wird zunächst das nach dem Stand der Technik wohlbekannte Prinzip eines Mikrostreifenleitungs-Resonators in Erinnerung gebracht. Bei dieser Struktur befindet sich auf einem geeigneten dielektischem Substrat 1 eine durchgehende Leiterschicht 2 auf der einen und eine strukturierte Leiterschicht auf der anderen Seite, Fig. 11a zeigt die wohlbekannte Struktur eines Mikrostreifenleitungs-Resonators 3, welcher an seinen Enden kapazitiv mit den Zuleitungen 4,5 verkoppelt ist. Der Frequenzgang des Leistungsübertragungsfaktors 6 zeigt ein Maximum bei der Frequenz f0 und die Breite dieses Maximums läßt sich über die Stärke der Kopplung an den Leitungsenden (Diskontinuitäten) verändern. Fig. 11b zeigt, wie ein erfindungsgemäßes impedanz-unsymmetrisches Filterglied in Mikrostreifenleitungs-Technologie realisiert werden kann. Dazu wird eine T-förmige Leiterstruktur verwendet, bei der die Länge der einzelnen Arme etwa einer viertel Leitungswellenlänge bei der Mittenfrequenz entspricht, wobei eine wohldefinierte Unsymmetrie in der Länge oder Breite der Seitenarme 3 für die Funktion notwendig ist. Die Seitenarme stellen eine einfache Realisierung der Sperr-Resonatoren dar, wobei die Sperrfrequenzen über die Länge der Arme beeinflußt wird. Zusammen mit dem dritten Arm, bilden die Seitenarme ein Gebilde, welches bei zwei unterschiedlichen Frequenzen resoniert und damit stellt die T-Struktur eine Sonderform eines Dual-Mode-Resonators dar. Das Ausgangstor kann auf die in Fig. 11b gezeigte Weise kapazitiv an die T-Struktur angekoppelt werden. Der Frequenzgang 6 des so entstandenen Zweitors ist durch zwei Transmissionsmaxima und zwei Transmissions-Nullstellen gekennzeichnet, wobei aufgrund der Unsymmetrie der absolute Wert des Transmissionsmaximums weit unter Eins liegen kann. Aus diesem Grund stellt ein einzelnes unsymmetrisches Filterglied - im Gegensatz zum Impedanz-symmetrischen Filterglied- noch kein brauchbares Bandpassfilter dar. Wie bei allen oben gezeigten Realisierungsbeispielen, läßt sich auch diese Mikrostreifenleitungsstruktur in vielfältiger Weise abändern, z. B. durch Verwendung inhomogener Leitungsstücke veränderlicher Breite.11 illustrates the implementation of an impedance-asymmetrical filter element according to the invention using microstrip line technology. In Fig. 11a, the principle of a microstrip line resonator, which is well known in the prior art, is first brought to mind. In this structure there is a continuous conductor layer 2 on one side and a structured conductor layer on the other on a suitable dielectric substrate 1. FIG. 11a shows the well-known structure of a microstrip line resonator 3, which at its ends is capacitively connected to the leads 4, 5 is coupled. The frequency response of the power transmission factor 6 shows a maximum at the frequency f 0 and the width of this maximum can be changed via the strength of the coupling at the line ends (discontinuities). 11b shows how an impedance-asymmetrical filter element according to the invention can be realized in microstrip line technology. For this purpose, a T-shaped conductor structure is used in which the length of the individual arms corresponds to approximately a quarter of the line wavelength at the center frequency, a well-defined asymmetry in the length or width of the side arms 3 being necessary for the function. The side arms represent a simple implementation of the blocking resonators, the blocking frequencies being influenced over the length of the arms. Together with the third arm, the side arms form a structure which resonates at two different frequencies and thus the T-structure is a special form of a dual-mode resonator. The output gate can be capacitively connected to the T in the manner shown in FIG. 11b Structure to be coupled. The frequency response 6 of the two-port thus created is characterized by two transmission maxima and two transmission zeros, the absolute value of the transmission maximum being able to be far below one due to the asymmetry. For this reason, a single asymmetrical filter element - in contrast to the impedance-symmetrical filter element - is not yet a usable bandpass filter. As with all of the implementation examples shown above, this microstrip line structure can also be modified in a variety of ways, e.g. B. by using inhomogeneous line pieces of variable width.

    Fig. 12 zeigt exemplarisch, wie aus 4 impedanz-unsymmetrischen Filtergliedern 1 und einem konventionellen Halbwellen-Resonator 2 ein 9-poliges Filter mit 8 Transmissions-Nullstellen gebildet werden kann. Der Resonator 2 übernimmt in der Kaskade neben der Bereitstellung eines zusätzlichen Pols, die Transformation der Impedanz an Tor 2 (z. B. 50 Ohm) auf das niedrige Impedanzniveau an der Koppelstelle zum Verzweigungspunkt der T-förmigen Resonatoren. Die Dimensionierung der Parameter der einzelnen Filterglieder, kann hierbei z. B. so erfolgen, daß eine Cauer-Charakteristik für den Frequenzgang erzielt wird.Fig. 12 shows an example of how 4 impedance-unbalanced Filter members 1 and one conventional half-wave resonator 2 a 9-pin Filters with 8 transmission zeros are formed can. The resonator 2 takes over in the cascade Providing an additional pole that Transformation of the impedance at gate 2 (e.g. 50 ohms) the low impedance level at the coupling point to Branch point of the T-shaped resonators. The Dimensioning the parameters of each Filter elements, for example. B. done so that a Cauer characteristic for the frequency response is achieved.

    Claims (14)

    1. A high-frequency bandpass filter arrangement, comprising a main resonator (1) and at least one bandstop resonator (4, 6, 8), coupled to the main resonator (1), the main resonator (1) being defined by a line piece delimited on both sides by discontinuities (2 and 3 in Figures 2a to 2c) in the form of an interruption or metal wall, and having an electromagnetic natural resonance at a mid-band frequency (f0),
      characterized in that the bandstop resonator (4), which is coupled to the main resonator, implements a reflection factor of absolute value one at its stop frequency (fs) for a wave on the line piece of the main resonator (1), and the at least one bandstop resonator is coupled to the main resonator at those locations along the line piece at which, due to the spatial variation of the electrical and magnetic fields along the line, the frequency-dependent coupling between the bandstop resonator and the main resonator disappears at the mid-band frequency of the bandpass filter, the bandstop resonator implementing a transmission zero in the stopband and, together with the remaining filter structure, a damping zero in the pass band.
    2. The high-frequency bandpass filter arrangement according to Claim 1,
      characterized in that the at least one bandstop resonator is formed by a pair of bandstop resonators positioned symmetrically to one another.
    3. The high-frequency bandpass filter arrangement according to Claim 1,
      characterized in that the coupling of the at least one bandstop resonator to the main resonator is performed electrically.
    4. The high-frequency bandpass filter arrangement according to Claim 1,
      characterized in that the coupling of the at least one bandstop resonator to the main resonator is performed magnetically.
    5. The high-frequency bandpass filter arrangement according to Claim 1,
      characterized in that the coupling of the at least one bandstop resonator to the main resonator is performed galvanically.
    6. The bandpass filter according to Claim 1, having three natural frequencies (poles) and two transmission zeros,
      characterized in that the main resonator is formed by a line piece having a length which, at the mid-band frequency of the bandpass filter, corresponds to approximately one-half of the line wavelength, two bandstop resonators are coupled to the main resonator in the middle of the line piece in such a way that the frequency-dependent coupling disappears at the mid-band frequency,
      the stop frequency of one of the two bandstop resonators is less than the mid-band frequency of the bandpass filter and the stop frequency of the other bandstop resonator is greater than the mid-band frequency of the bandpass filter,
      those frequencies in the stopband at which transmission zeros of the bandpass filter are desired are selected for the stop frequencies of the two bandstop resonators,
      and, using the strength of the coupling between the bandstop resonators and the main resonators, the three transmission maxima are displaced within the pass band in such a way that the reflection damping in the pass band range lies above a predetermined minimum value.
    7. The bandpass filter according to Claim 1 having five natural frequencies (poles) and four transmission zeros,
      characterized in that the main resonator is formed by a line piece having a length which, at the mid-band frequency, corresponds to approximately one line wavelength, two pairs of bandstop resonators are coupled to the main resonator, at a mutual distance of approximately one-half of the mid-band frequency line wavelength along the line piece of the main resonator and a distance of approximately one-fourth of the line wavelength between the external bandstop resonator pairs and the ends of the line piece, in such a way that the frequency-dependent coupling disappears at the mid-band frequency of the bandpass filter.
    8. The bandpass filter according to Claim 1 having 2m+1 (with m as a natural number) natural frequencies (poles) and 2m transmission zeros,
      characterized in that the main resonator is formed by a line piece having a length of approximately m times one half of the mid-band frequency line wavelength, m pairs of bandstop resonators are coupled to the main resonator, at a mutual distance of approximately one-half of the mid-band frequency line wavelength along the line piece of the main resonator and a distance of approximately one-fourth of the line wavelength between the external bandstop resonator pairs and the ends of the line piece, in such a way that the frequency-dependent coupling disappears at the mid-band frequency of the bandpass filter.
    9. The bandpass filter according to Claim 1 having 2m+1 (with m as a natural number) natural frequencies (poles) and 2m transmission zeros,
      characterized in that the main resonator is formed by a line piece having a length of approximately (m+1) times one half of the mid-band frequency line wavelength, m pairs of bandstop resonators are coupled to the main resonator, at a mutual distance of approximately one-half of the mid-band frequency line wavelength along the line piece of the main resonator and a distance of approximately one-half of the line wavelength between the external bandstop resonator pairs and the ends of the line piece, in such a way that the frequency-dependent coupling disappears at the mid-band frequency of the bandpass filter.
    10. The bandpass filter according to Claim 1 having a cascade of filter elements (number Q), in which these filter elements are formed by bandpass filters according to one of Claims 2 to 5,
      characterized in that one end of the line piece of the filter element used as the main resonator is electrically or magnetically or galvanically coupled to the neighboring end of the line piece of the next filter element, and in which the two external ends of the line pieces of the external filter elements are coupled to the input and/or the output gate.
    11. The bandpass filter according to one of Claims 1 to 10,
      characterized in that the resonators are implemented as coaxial resonators.
    12. The bandpass filter according to one of Claims 1 to 10,
      characterized in that the resonators are implemented as cavity resonators.
    13. The bandpass filter according to one of Claims 1 to 10,
      characterized in that the resonators are implemented as dielectric resonators.
    14. The bandpass filter according to one of Claims 1 to 10,
      having planar microstrip line resonators or coplanar resonators, including planar resonators made of high-temperature superconductors.
    EP00960529A 1999-08-31 2000-08-26 High-frequency band pass filter assembly, comprising attenuation poles Expired - Lifetime EP1212806B1 (en)

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    DE19941311 1999-08-31
    DE19941311A DE19941311C1 (en) 1999-08-31 1999-08-31 Band filter
    PCT/EP2000/008333 WO2001017057A1 (en) 1999-08-31 2000-08-26 High-frequency band pass filter assembly, comprising attenuation poles

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    EP1212806B1 true EP1212806B1 (en) 2003-03-05

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    CN1241289C (en) 2006-02-08
    WO2001017057A1 (en) 2001-03-08
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    EP1212806A1 (en) 2002-06-12

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