EP1185047A1 - Schätzung sowie Beseitigung von Intersymbolinterferenz in Mehrträgersignalen - Google Patents

Schätzung sowie Beseitigung von Intersymbolinterferenz in Mehrträgersignalen Download PDF

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Publication number
EP1185047A1
EP1185047A1 EP00118659A EP00118659A EP1185047A1 EP 1185047 A1 EP1185047 A1 EP 1185047A1 EP 00118659 A EP00118659 A EP 00118659A EP 00118659 A EP00118659 A EP 00118659A EP 1185047 A1 EP1185047 A1 EP 1185047A1
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EP
European Patent Office
Prior art keywords
symbol
filter
transient
filter function
received
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Withdrawn
Application number
EP00118659A
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English (en)
French (fr)
Inventor
Tore André
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Telefonaktiebolaget LM Ericsson AB
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Telefonaktiebolaget LM Ericsson AB
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Telefonaktiebolaget LM Ericsson AB filed Critical Telefonaktiebolaget LM Ericsson AB
Priority to EP00118659A priority Critical patent/EP1185047A1/de
Priority to EP01969582A priority patent/EP1314289A1/de
Priority to PCT/EP2001/009229 priority patent/WO2002019647A1/en
Priority to AU2001289795A priority patent/AU2001289795A1/en
Publication of EP1185047A1 publication Critical patent/EP1185047A1/de
Withdrawn legal-status Critical Current

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Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L25/03012—Arrangements for removing intersymbol interference operating in the time domain
    • H04L25/03019—Arrangements for removing intersymbol interference operating in the time domain adaptive, i.e. capable of adjustment during data reception
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L2025/0335—Arrangements for removing intersymbol interference characterised by the type of transmission
    • H04L2025/03375—Passband transmission
    • H04L2025/03414—Multicarrier
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L2025/03433—Arrangements for removing intersymbol interference characterised by equaliser structure
    • H04L2025/03439—Fixed structures
    • H04L2025/03445—Time domain
    • H04L2025/03471—Tapped delay lines
    • H04L2025/03484—Tapped delay lines time-recursive
    • H04L2025/03496—Tapped delay lines time-recursive as a prediction filter

Definitions

  • the invention relates to multi-carrier digital transmission systems and has particular relevance to discrete multi-tone or orthogonal frequency division multiplexed systems for use over digital subscriber lines or radio broadcast systems.
  • Digital subscriber line technologies commonly termed xDSL enable highspeed digital data to be transmitted down an ordinary phone line.
  • the modulation scheme standardised for asymmetric DSL (ADSL) and proposed for very high speed DSL (VDSL) is discrete multi-tone modulation DMT.
  • DSL asymmetric DSL
  • VDSL very high speed DSL
  • Modulation can be achieved by performing an inverse fast Fourier transform (IFFT), with fast Fourier transform (FFT) used for demodulation.
  • IFFT inverse fast Fourier transform
  • FFT fast Fourier transform
  • the output from one IFFT calculation is termed a discrete multi-tone symbol and is sent over the channel after conversion to an analogue signal.
  • the interference in one symbol is a combination of the interference due to a previously transmitted symbol, which is correctly termed the intersymbol interference ISI, and the interference due to the symbol itself, or the intercarrier interference.
  • ISI intersymbol interference
  • ISI can be viewed as a transient or decaying 'tail' generated at the discontinuity where consecutive symbols meet.
  • the guard time typically contains a cyclic extension of the symbol. Specifically, a copy of the end of each symbol is added to the beginning of the symbol in the form of a cyclic prefix.
  • the carriers are continuous from the beginning of the cyclic prefix to the end of the symbol. Thus any interference will be generated at the discontinuity between the start of the cyclic prefix and the end of the previous symbol.
  • the lengths of cyclic prefixes vary according to the application, but typically consist of no more than 10% of the symbol. Longer guard intervals are unfavourable because they introduce a bandwidth penalty.
  • the ISI transient generated at the boundary between symbols will terminate within the cyclic prefix, leaving the subsequent symbol intact.
  • the impulse response of the channel which includes the effects of filters in the transmitter and receiver, can be very long, and often exceed the guard interval. Residual intersymbol interference will then occur which can severely impair the quality of the received signals.
  • the estimated symbols are then converted back to the time domain using IFFT, and the ISI determined and removed using the algorithm.
  • the residual symbol is then reconverted to the frequency domain using FFT and the decisions made.
  • An iterative process then follows to remove the ICI. Since the decisions on the transmitted symbols may initially be erroneous, an iterative process is required to accurately determine the interference. This necessarily entails a very large number of calculations, so that the process as a whole demands very high processing power.
  • Time domain equalizers TEQ are also used in the art to mitigate the effects of ISI between symbols transmitted over a distorting channel.
  • a time domain equalizer is constituted by a filter, generally a finite impulse response (FIR) filter and has the effect of shortening the impulse response of the channel. This can be achieved, for example, by cancelling the poles in the channel transfer function. By using a suitable algorithm, the channel impulse response can be made shorter than the cyclic prefix utilised.
  • a drawback of TEQs is that both the noise and the signal are filtered. When a TEQ cancels the poles in the channel transfer function it will also attenuate some signal frequencies and amplify noise at other frequencies. The noise will leak into the side lobes of the fast Fourier transform in the receiver and degrade performance. Hence adapting the TEQ to minimise ISI will generally result in a sub-optimal signal to noise ratio.
  • intersymbol interference is compensated for by generating an estimate of the ISI tail and subtracting this from the received signals. This is achieved by subtracting the symbol prefix from the beginning of each received symbol to obtain a portion of a transient intersymbol interference signal generated during transmission and using the transient portion with a filter function adapted to generating an estimate of the full transient signal. This transient signal is then subtracted from the received symbol to correct the same.
  • the filter function may be preceded by a processing arrangement for generating the initial conditions of the filter function. The filter function is then used to generate the full transient signal when input with a predetermined value.
  • the processing arrangement may be a second filter or be adapted to perform calculations.
  • the filter function may alternatively be an adaptive filter function that is configured prior to use with a training sequence.
  • the adaptive filter function may also be adjusted using each received symbol to generate an error update signal.
  • the specific compensation for intersymbol interference in accordance with the invention means that the signal can be filtered separately to remove noise due to other sources in the normal way. This allows each process to be optimised without having a detrimental effect on the other. Moreover, the processing power required for this compensation is relatively small, since it requires the generation of a transient symbol using only the portion of the actual transient signal contained in the cyclic prefix and the cancellation of the interference by subtracting the generated signal from the beginning of the received symbol.
  • Fig. 1 shows a block diagram representing part of an end to end link of a multi-carrier transmission system that uses discrete multi-tone modulation DMT with a transmitter 1 and receiver 2.
  • the link includes at least part of a normal telephone line, which may include a twisted pair phone line.
  • this arrangement may be used for a variety of other transmission media, including broadcast radio, for example using orthogonal frequency division multiplexing (OFDM).
  • OFDM orthogonal frequency division multiplexing
  • the transmitter 1 and receiver 2 depicted in Fig. 1 are preferably part of modems that support Digital Subscriber Line (DSL) transmission.
  • DSL Digital Subscriber Line
  • these modems support either ADSL (asymmetric DSL) or VDSL (Very highspeed DSL) modulation schemes, or any of the other modulation schemes which fall under the collective term xDSL.
  • ADSL asymmetric DSL
  • VDSL Very highspeed DSL
  • the transmitter 1 incorporates several components including an encoder (not shown), a discrete multi-tone modulator 10, a cyclic prefix adder 11 a parallel to serial converter 12, a digital to analogue converter 13 and an analogue transmitter with high-pass filter 14.
  • the encoder is connected upstream of the discrete multi-tone modulator 10 and serves to translate incoming bit streams into in phase, in quadrature sequences for each of a plurality of sub-channels.
  • These encoded sequences are input to the multi-tone modulator 10, which is preferably an IFFT modulator 10 that computes the inverse fast Fourier transform by an appropriate algorithm.
  • the discrete multi-tone encoded symbols generated in the IFFT modulator 10 are then each cyclically extended by the addition of a cyclic prefix in CP circuitry 11. This is done by duplicating a number of samples at the end of a symbol and joining these to the beginning of the symbol. The number of samples included in the cyclic prefix depends on the application and acceptable bandwidth penalty. Typically a cyclic prefix does not exceed 10% of a symbol.
  • the parallel symbol sequences are then converted to a serial sample stream in a parallel to serial converter 12.
  • the serial sample stream is then converted to an analogue signal with digital to analogue converter 13, prior to transmission over the link by an analogue transmitter 14, which incorporates a high-pass filter for filtering out signals in the transmitter 1 that can interfere with the POTS band.
  • the signal is then sent over the transmission medium denoted by a channel 30 having a transfer function 'h' to a remote location.
  • an analogue receiver 20 which also incorporates a high-pass filter for filtering out noise from the POTS band, receives the signal and inputs this into an A/D converter 21 for digital conversion.
  • the digital bit stream is then sent to the ISI compensation block consisting of delay circuitry 22, a time domain equalizer (TEQ) 23, which may be a finite impulse response (FIR) filter, tail estimation circuitry 24 and an adder 25.
  • TEQ time domain equalizer
  • FIR finite impulse response
  • tail estimation circuitry 24 tail estimation circuitry 24
  • adder 25 an adder 25.
  • This block will be described in more detail below, however it should be noted that the time domain equalizer 23 is optional in this arrangement. It serves to equalise the received signal, but is not necessary for reducing the ISI.
  • the received symbols are then returned to parallel format in a serial to parallel converter 26.
  • the cyclic prefix is discarded in CP removal circuitry 27 and the symbols are then demodulated in FFT circuitry 28 which performs a fast Fourier transform, and decoded by a decoder (not shown).
  • Figs. 2 two consecutive DMT symbols transmitted over the distorting channel 30 are illustrated. For the purposes of illustration only one of the carriers of the DMT symbols is shown. Each symbol is made up of the DMT information-carrying portion 100, and a cyclic prefix CP 110.
  • the information carrying portion is correctly the DMT symbol, however in the present embodiment, the information is always transmitted with a cyclic prefix so it is considered more appropriate to term the whole signal a symbol.
  • the cyclic prefix 110 is constructed from a number of samples taken from the end of the information portion 100 and joined to the beginning of the information portion as a prefix. As is apparent from Fig.
  • the carrier signal is continuous from the beginning of the cyclic prefix CP 110 to the end of the information portion 100.
  • a discontinuity is present at the boundary between two DMT signals or rather between the end of the first information portion 100 and the beginning of the cyclic prefix 110 of the following symbol.
  • the resultant signal dispersion or spread in time causes interference between the symbols at this boundary.
  • Fig. 2b wherein the information portion 100 and cyclic prefix 110 of a DMT symbol are represented as blocks.
  • the interference between consecutive symbols manifests itself as a transient signal or 'tail' that originates at the discontinuity.
  • the transient signal is present in the cyclic prefix CP 110, but also extends to greater or lesser extent into the DMT information-carrying portion 100 depending on the impulse response of the channel 30.
  • an estimate of this transient interference signal is generated at the receiver for each received symbol and subsequently used to substantially cancel out the transient in the received symbol.
  • the corrected symbol can then be processed in the conventional manner to take account of noise that is present throughout the whole symbol. More specifically, the portion of the transient signal present in the cyclic prefix CP 110 is isolated and used to generate at least an estimate of the full transient signal.
  • the cyclic prefix is a known sequence and in the present embodiment is a copy of a number of samples taken from the end of the information portion 100 of the symbol.
  • the end portion of the received DMT symbol is in general not significantly affected by the intersymbol interference and can therefore be used to isolate the transient signal portion contained in the cyclic prefix 110 by subtraction from the latter.
  • the generation of a full transient signal from the isolated transient signal portion is achieved using a filter system.
  • Fig. 3 shows an arrangement for cancelling the intersymbol interference in accordance with a first embodiment of the present invention.
  • This arrangement shows the DMT symbol x(n) entering the transmission medium or channel 30 and exiting as the received signal y(n).
  • This received symbol is delayed in delay circuitry 101 by a period k to produce the symbol y(n-k).
  • the period k is equivalent to the delay required to generate an estimate of the complete transient signal.
  • the transient signal is then subtracted from the delayed received symbol y(n-k) in the adder 106 to obtain a DMT symbol with substantially no interference.
  • the tail estimation circuitry is provided by a first switch 107 that represents symbol synchronisation.
  • the switch 107 is initially closed to relay the beginning of the received symbol, that is the cyclic prefix CP 110 to the remaining tail estimation circuitry.
  • the switch then opens until the last few samples of the symbol from which the cyclic prefix were formed at the transmitter arrive at the receiver, at which point it closes again. These final samples are then also passed on to the remaining tail estimation circuitry.
  • This circuitry includes a delay element 102 which holds the extracted symbol bits CP bits for a time period that is equivalent to the delay required to transmit a full information portion 100, i.e. the time required to transmit a DMT symbol without the cyclic prefix 110.
  • the first bits of the final portion of the received symbol are conducted through the switch 107 the first bits of the cyclic prefix 110 will emerge from the delay element 102. These are then subtracted at the adder 103 to form a signal wherein the information content is substantially removed leaving only beginning of the transient signal as illustrated by the encircled signal curve.
  • This transient signal portion is fed to a filter arrangement that comprises two processing modules 104, 105.
  • the first module 104 uses the transient signal portion to calculate the initial conditions for the second module which is a filter 105.
  • the filter 105 receives an input of zeros and generates an estimate of the complete transient signal with the programmed initial conditions.
  • the filter 105 has an impulse response that is an estimate of the impulse response of the channel 30. This estimated impulse response is denoted by h.
  • the channel impulse response h is known at the receiver.
  • One such method starts by determining the channel frequency response H and. Specifically, a received signal Rx is divided by the known transmitted signal Tx at each frequency in the frequency domain to determine the attenuation and phase shift representing the channel response in the frequency domain Hest. Conversion to the time domain to obtain the estimated channel impulse response h and is accomplished by calculating the inverse fast Fourier transform of the determined frequency response H and.
  • the filter 105 If the filter 105 is provided with the same initial conditions as the channel in the transition from one symbol to the next and the input to the filter is zero, the output of the filter 105 will be the full transient signal or tail.
  • the initial condition for the filter 105 can be calculated in the first processing module 104 from the first samples of the transient signal or tail using reversed difference equations. If the channel can be modelled as a function h(z) of degree M, then M-1 samples are necessary to calculate the initial conditions. If M-1 is less than the length of the cyclic prefix it is possible to calculate the initial conditions. Naturally, the samples used should ideally be free from noise. This is usually not the case, however a good approximation of the transient signal is nevertheless obtainable.
  • the first processing module 104 is a filter that models the inverse response of the poles of the channel response to generate the initial conditions for the transient generating filter 105.
  • This is a finite impulse response (FIR) filter.
  • the output of this filter is used to initialise filter 105 as will be described below.
  • the transfer function of the tail filter 105 is a model of the poles in the impulse response of the channel 30.
  • This filter 105 is an infinite impulse response (IIR) filter of a transposed direct form II structure. The arrangement including the initial condition filter 104 and the transient generating filter 105 is illustrated in Fig. 4.
  • the filter 105 is depicted as a standard transposed direct form II filter with coefficient factors 115 and 116 for the zero and pole coefficients, respectively.
  • a feedback path is also included starting from the output of the last summing element 118 and passing through each of the pole coefficient factors into the summing elements 118.
  • Each summing element 118 is separated from the next by a delay element 117.
  • the output of the initial condition filter 104 illustrated schematically by dashed arrows is used to program the delay elements 117 and so initialise the filter 105.
  • the filter 105 is shown with coefficients for both the poles a and zeros b.
  • the coefficients b are not required and could be set at any random value or suppressed completely.
  • the zeros of the channel b(z) are effectively included in the programmed initial condition and do not have to be taken care of especially in the transient generating filter 105.
  • Fig. 4 provides an acceptable approximation of the transient signal with low complexity.
  • Other filter structures may be better in terms of sensitivity to small errors in the coefficient values or in the initial condition. It will be understood that when a different filter structure is used than the transposed direct form II structure illustrated in Fig. 4, then the initial condition must also be changed and adapted to the modified structure.
  • a finite impulse response (FIR) filter is used as the transient generating filter 105.
  • the circuit arrangement is similar to that of Fig. 3, with the sole exception that the initial condition of the filter 105 is obtained by solving a system of equations.
  • the processing module 104 is arranged to perform this calculation and does not have a filter structure.
  • the processing module may be any suitable processing means, including software controlled arrangements such as microprocessor, microcomputer, or custom built circuitry.
  • the equation system is described as follows.
  • the initial condition of filter 105 is denoted by d k and the transfer response of the FIR filter 105 denoted by g k .
  • the estimated tail or transient signal is denoted by y est .
  • d k d k can thus be calculated as gm/y est , where the operator '/' denotes matrix division.
  • the equation system can be either determined or over-determined.
  • the transient signal y est is the signal isolated from the cyclic prefix CP of the received symbol and will thus depend on the length of the cyclic prefix.
  • the inversion of gm need be calculated only once, while the matrix multiplication must be performed for each received symbol.
  • FIG. 5 An arrangement according to a further embodiment of the invention is schematically depicted in Fig. 5.
  • this circuit comprises the channel 30 with channel response h.
  • a delay element 101 for delaying the received DMT symbols.
  • a switch 107 for synchronising the tail estimation with the received symbols. As before, this switch feeds the cyclic prefix of each received symbol to the tail estimation circuitry and then opens until the final portion of the symbol, of the same length as the cyclic prefix, is received. This is then also fed to the tail estimation circuitry.
  • a delay circuit 102 delays the cyclic prefix until the final symbol portion is received whereupon the two are added in adder 103 to substantially remove the information content from the transient signal portion. This latter portion is then input into an adaptive filter 120, which, after suitable training, generates an estimate of the full transient signal. The estimated full transient signal is fed to adder circuit 106, which simultaneously receives first bits of the delayed DMT symbol from delay circuit 101.
  • the adaptive filter 120 may be trained initially using a special training sequence followed by silence.
  • the filter design can then be frozen and used as an unchanging transfer function.
  • the filter may be adapted using an error signal generated from each received symbol. Because the symbols will be received without intervals, an error signal present in each symbol must be generated. This is achieved by storing each received symbol that has been corrected by subtraction of the tail estimate in a delay buffer 109, while the symbol is simultaneously transformed back to the frequency domain using the fast Fourier transform demodulator 128, decoded using the frequency domain equalizer 107, and quantizer 106 and then subsequently encoded using the reverse function 108 of the frequency domain equalizer 107 and transformed back to the time domain using an inverse fast Fourier transform modulator 110. This decoded signal is then summed with the stored corrected received symbol at adder 111 to generate an error signal that can be used to update the adaptive filter 120.
  • filter and processing modules may be used to generate a full transient signal using the extracted transient portion.
  • filter and processing modules may be used to generate a full transient signal using the extracted transient portion.
  • an arrangement based on the embodiment depicted in Fig. 4 using two filter arrangements 104, 105 may be envisaged, wherein both filters are adaptive to provide enhanced performance.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Cable Transmission Systems, Equalization Of Radio And Reduction Of Echo (AREA)
EP00118659A 2000-08-29 2000-08-29 Schätzung sowie Beseitigung von Intersymbolinterferenz in Mehrträgersignalen Withdrawn EP1185047A1 (de)

Priority Applications (4)

Application Number Priority Date Filing Date Title
EP00118659A EP1185047A1 (de) 2000-08-29 2000-08-29 Schätzung sowie Beseitigung von Intersymbolinterferenz in Mehrträgersignalen
EP01969582A EP1314289A1 (de) 2000-08-29 2001-08-09 Schätzung sowie beseitigung von intersymbolinterferenz in mehrträgersignalen
PCT/EP2001/009229 WO2002019647A1 (en) 2000-08-29 2001-08-09 Estimation and removal of intersymbol interference in multicarrier signals
AU2001289795A AU2001289795A1 (en) 2000-08-29 2001-08-09 Estimation and removal of intersymbol interference in multicarrier signals

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Application Number Priority Date Filing Date Title
EP00118659A EP1185047A1 (de) 2000-08-29 2000-08-29 Schätzung sowie Beseitigung von Intersymbolinterferenz in Mehrträgersignalen

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EP01969582A Ceased EP1314289A1 (de) 2000-08-29 2001-08-09 Schätzung sowie beseitigung von intersymbolinterferenz in mehrträgersignalen

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Cited By (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP1525698A1 (de) * 2002-07-18 2005-04-27 Motorola, Inc. Verfahren und empfänger zur modulation eines trainingspräfixes
US7543009B2 (en) * 2002-12-24 2009-06-02 Stmicroelectronics Belgium Nv Fractional fourier transform convolver arrangement
WO2010101568A1 (en) * 2009-03-04 2010-09-10 Adaptive Spectrum And Signal Alignment, Inc. Dsl noise cancellation
CN102739598A (zh) * 2011-03-30 2012-10-17 承景科技股份有限公司 码元干扰移除方法与接收器

Families Citing this family (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2022539846A (ja) * 2019-07-11 2022-09-13 テレフオンアクチーボラゲット エルエム エリクソン（パブル） 協調ビームフォーミングのためのガードインターバル適応

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WO1999044326A2 (en) * 1998-02-26 1999-09-02 Wavesat Telecom Inc. Ofdm frame synchronisation and equalisation system
DE19858106A1 (de) * 1998-12-16 2000-06-29 Ericsson Telefon Ab L M Empfänger und Verfahren zum Verhindern einer Zwischensymbolstörung in einem Hochgeschwindigkeitsübertragungssystem

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WO1999044326A2 (en) * 1998-02-26 1999-09-02 Wavesat Telecom Inc. Ofdm frame synchronisation and equalisation system
DE19858106A1 (de) * 1998-12-16 2000-06-29 Ericsson Telefon Ab L M Empfänger und Verfahren zum Verhindern einer Zwischensymbolstörung in einem Hochgeschwindigkeitsübertragungssystem

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Cited By (8)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP1525698A1 (de) * 2002-07-18 2005-04-27 Motorola, Inc. Verfahren und empfänger zur modulation eines trainingspräfixes
EP1525698A4 (de) * 2002-07-18 2009-06-10 Motorola Inc Verfahren und empfänger zur modulation eines trainingspräfixes
US7543009B2 (en) * 2002-12-24 2009-06-02 Stmicroelectronics Belgium Nv Fractional fourier transform convolver arrangement
WO2010101568A1 (en) * 2009-03-04 2010-09-10 Adaptive Spectrum And Signal Alignment, Inc. Dsl noise cancellation
CN102379090A (zh) * 2009-03-04 2012-03-14 适应性频谱和信号校正股份有限公司 Dsl噪声消除
CN102379090B (zh) * 2009-03-04 2015-01-21 适应性频谱和信号校正股份有限公司 Dsl噪声消除
US9819388B2 (en) 2009-03-04 2017-11-14 Mark B. Flowers DSL noise cancellation
CN102739598A (zh) * 2011-03-30 2012-10-17 承景科技股份有限公司 码元干扰移除方法与接收器

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EP1314289A1 (de) 2003-05-28
WO2002019647A1 (en) 2002-03-07
AU2001289795A1 (en) 2002-03-13

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