EP1166435A1 - Non-linear distortion generator - Google Patents

Non-linear distortion generator

Info

Publication number
EP1166435A1
EP1166435A1 EP00919766A EP00919766A EP1166435A1 EP 1166435 A1 EP1166435 A1 EP 1166435A1 EP 00919766 A EP00919766 A EP 00919766A EP 00919766 A EP00919766 A EP 00919766A EP 1166435 A1 EP1166435 A1 EP 1166435A1
Authority
EP
European Patent Office
Prior art keywords
circuit
linear
signal
distortion
coupled
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Granted
Application number
EP00919766A
Other languages
German (de)
French (fr)
Other versions
EP1166435B1 (en
Inventor
Shutong Zhou
Timothy J. Brophy
Richard A. Meier
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Arris Technology Inc
Original Assignee
Arris Technology Inc
General Instrument Corp
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Arris Technology Inc, General Instrument Corp filed Critical Arris Technology Inc
Publication of EP1166435A1 publication Critical patent/EP1166435A1/en
Application granted granted Critical
Publication of EP1166435B1 publication Critical patent/EP1166435B1/en
Anticipated expiration legal-status Critical
Expired - Lifetime legal-status Critical Current

Links

Classifications

    • HELECTRICITY
    • H03ELECTRONIC CIRCUITRY
    • H03FAMPLIFIERS
    • H03F1/00Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/32Modifications of amplifiers to reduce non-linear distortion
    • H03F1/3241Modifications of amplifiers to reduce non-linear distortion using predistortion circuits
    • H03F1/3276Modifications of amplifiers to reduce non-linear distortion using predistortion circuits using the nonlinearity inherent to components, e.g. a diode
    • HELECTRICITY
    • H03ELECTRONIC CIRCUITRY
    • H03FAMPLIFIERS
    • H03F1/00Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/30Modifications of amplifiers to reduce influence of variations of temperature or supply voltage or other physical parameters
    • H03F1/303Modifications of amplifiers to reduce influence of variations of temperature or supply voltage or other physical parameters using a switching device

Definitions

  • the present invention relates generally to communication systems employing
  • the invention pertains to a non-linear
  • predistortion or postdistortion generator for coupling in-line with an amplifier
  • optical detector or laser to minimize the signal distortion caused by amplification.
  • Amplifiers are widely used in many types of communication applications.
  • Minimizing distortion is particularly important when a series of amplifiers is
  • a signal transmission path such as a series of RF amplifiers in a
  • CATV transmission system Disposed throughout a CATV transmission system are
  • RF amplifiers that periodically amplify the transmitted signals to counteract cable attenuation and attenuation caused by passive CATV components, such as, signal
  • the RF amplifiers are also employed to maintain the desired
  • each RF amplifier must provide minimum degradation
  • CTB composite triple beat
  • NLD reduce the signal power level
  • the first method reduces the
  • the second method is the feed forward technique. Using this technique, the
  • This distortion component is extracted. This distortion component is then amplified by
  • This circuitry is also complex and very temperature
  • the third method is the predistortion or postdistortion technique.
  • predistortion or postdistortion is used.
  • This circuit generates a distortion that is equal in amplitude but
  • circuitry disclosed therein is not matched to the NLD. Additionally, the '854
  • the present invention is an in-line predistortion or postdistortion generator
  • generator comprises an instant controlled non-linear attenuator which utilizes the non-linear current flowing through a pair of diodes to provide the proper amount of
  • circuitry is always matched to the NLD, thereby ensuring a frequency response that
  • the distortion generator also includes a temperature
  • NLD such as an RF amplifier, a laser diode
  • Figure 1 is a schematic diagram of a prior art distortion generator.
  • Figure 2 is a combination plot of the effect of using the outputs from the prior
  • Figure 3 is a combination plot of the effect of using the outputs from the prior
  • Figure 4 is schematic diagram of a ⁇ attenuator.
  • Figure 5 is a signal diagram of the diode non-linear current caused by the
  • Figure 6 is a schematic diagram of the preferred embodiment of the distortion
  • Figure 7 is a schematic diagram of the temperature compensation circuit.
  • the network 20 comprises a selected
  • the signal source is input at signal
  • Z j is the source of internal impedance which should be equal to the system
  • impedance Z 0 which is seen across the output 95.
  • the impedance values Z, and Z 0 are equal to
  • the values (Y) of resistors R 2 and R 3 are equal, and substantially larger
  • resistor R p is connected in parallel with resistor
  • the attenuator network 20 is matched at input and output, from D.C. to very high
  • Equation (8) shows that when R p (375 ohms) is in parallel with W x (7.5 ohms), the
  • the attenuator network 20 as shown is a linear
  • diodes are used to
  • Schottky diodes are utilized. At small current, diode current is exponentially proportional to the voltage across over the
  • diode can be used as a non-linear resistance.
  • the amount of attenuation can be calculated as:
  • I p is the current flow through R p , (the non-linear resistance).
  • I j is the
  • Equation 9 provides the relationship of the attenuation
  • Equation 9 provides a good estimation of how much non- linear current is required
  • I nonlinear Ii - 2 h + h Equation (10)
  • I non . l ⁇ near eff in Equation 12 is the effective non-linear current
  • Equation 13 shows that only a small
  • the ⁇ attenuator network 20 has low insertion loss and the voltage drop of
  • the input voltage on R x (shown in Figure 4) is proportional to the input voltage.
  • This voltage may be used to drive a pair of diodes to produce non-linear current.
  • the non-linear current flowing in the diodes will cause an attenuator to provide less
  • This may be used to compensate for the signal compression caused by amplification.
  • invention includes several additional components that modify a traditional ⁇ attenuator to achieve significantly better performance over a wide frequency and
  • the attenuator 100 has an input port 101, an output port 114 and
  • the attenuator 100 may be used in a predistortion
  • the output port 114 is connected to the input of an
  • the attenuator 100 is applied to the input port 101.
  • the attenuator 100 is applied to the input port 101.
  • resistors 105, 106, 107, 108, 112 includes resistors 105, 106, 107, 108, 112; capacitors 102, 103, 104, 111, 113, 115;
  • the inductor 117 is used in series with the resistor 108.
  • the function of the inductor 117 is used in series with the resistor 108.
  • inductor 117 is to make a parallel resonance circuit with the forward biased diode
  • 111, 113, and 115 are also used for D.C. blocking and AC coupling. From an AC
  • resistors 105 and 106 are equivalent to resistor R 2 of Figure 4.
  • resistors 105 and 106 are equivalent to resistor R 2 of Figure 4.
  • resistor 112 and capacitor 111 is functionally equivalent to resistor R 3 of Figure 4.
  • resistor 107 has no effect on RF signal attenuation.
  • resistors 105, 106, and 107 The other function for resistors 105, 106, and 107 is to supply a D.C. bias to
  • the diodes 109, 110 are first connected in series; and the series
  • resistor 107 has a low
  • the diodes 109, 110 will be primarily determined by the resistance of resistor 107.
  • resistor 107 in parallel with capacitors 103 and 104,
  • resistor 107 will
  • Diode 109 is connected to resistor 108 through capacitor 104 while diode 110
  • Diode 109 is responsible for the
  • capacitors 103 and 104 have the same value but different signs.
  • the present invention has several unique advantages over the prior art. Due to
  • the attenuator 100 produces only odd order distortion.
  • the attenuator 100 also uses two low series resistances 107, 108. From a D.C.
  • resistor 107 significantly improves the correction efficiency and reduces
  • resistor 108 provides for distortion correction with low insertion losses. Due to the
  • circuitry and delay lines This permits a circuit design which is much less complex
  • the present attenuator design uses low series resistance 108.
  • the third order correction circuit may work over a wide frequency range and
  • This correction circuit design is flexible and may be
  • This circuit is always matched to its input side and output side over wide frequency
  • Table 1 provides a listing of the components shown in Figure 6. However,
  • the value of resistor 108 may range from approximately 2 ⁇
  • resistor 107 may range from approximately 100 ⁇ to
  • the attenuator 100 uses the non-linear current
  • the attenuator 100 comprises capacitance, resistance and two
  • the diodes are the only components that are sensitive to temperature change
  • NLDs typically exhibit more distortion as the ambient temperature rises.
  • the temperature compensation circuit 200 is shown. The temperature compensation circuit 200
  • the temperature compensation circuit 200 As shown, the temperature compensation circuit 200
  • the negative temperature coefficient thermistor 211 is coupled in parallel with
  • resistor 210 to form a temperature linearized resistance, which is correlated to a
  • the PNP transistor 206 provides a constant current source
  • variable resistor 202 the amount of constant current through the PNP transistor
  • the voltage swing over temperature can be changed.
  • the constant current also passes through the variable resistor 209, thereby creating
  • the two diodes 205 and 208 are used to compensate for the junction voltage of the
  • Table 2 provides a listing of the components shown in Figure 7. However,
  • temperature compensation circuit 200 as disclosed herein is not utilized, the preferred

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Physics & Mathematics (AREA)
  • Nonlinear Science (AREA)
  • Amplifiers (AREA)
  • Cable Transmission Systems, Equalization Of Radio And Reduction Of Echo (AREA)
  • Semiconductor Lasers (AREA)

Abstract

An in-line distortion generator for coupling in-line with a non-linear device (NLD) produces an output signal of useful amplitude, but with low composite triple beat and cross modulation distortions. The distortion generator comprises an instant controlled non-linear attenuator which utilizes the non-linear current flowing through a pair of diodes to provide the proper amount of signal attenuation over the entire frequency bandwidth. The distortion generator circuitry is always matched to the NLD, thereby ensuring a frequency response that is predictable and predefined. The distortion generator may also include a temperature compensation circuit to ensure consistent operation throughout a wide temperature range.

Description

NON-LINEAR DISTORTION GENERATOR
BACKGROUND OF THE INVENTION
Field of the Invention
The present invention relates generally to communication systems employing
amplification devices. More particularly, the invention pertains to a non-linear
predistortion or postdistortion generator for coupling in-line with an amplifier,
optical detector or laser to minimize the signal distortion caused by amplification.
Description of the Related Art
Amplifiers are widely used in many types of communication applications.
Although it is preferable to keep amplifiers within their linear range of operation, it
has been increasingly necessary to extend the operation of amplifiers into high power
and high frequency regions of operation. Typically, the output power of an amplifier
is limited by the non-linearity of the active devices, including bipolar transistors and
FETs. These non-linearities result in distortions which are impressed upon the signal
being amplified. Reducing the non-linear distortions of an amplifier results in
increases of the output power, the system dynamic range and the carrier-to-noise
ratio. Accordingly, minimizing distortions and achieving linear frequency response
is paramount to efficient amplifier operation.
Minimizing distortion is particularly important when a series of amplifiers is
cascaded over a signal transmission path, such as a series of RF amplifiers in a
CATV transmission system. Disposed throughout a CATV transmission system are
RF amplifiers that periodically amplify the transmitted signals to counteract cable attenuation and attenuation caused by passive CATV components, such as, signal
splitters and equalizers. The RF amplifiers are also employed to maintain the desired
carrier-to-noise ratio. Due to the number of RF amplifiers employed in a given
CATV transmission system, each RF amplifier must provide minimum degradation
to the transmitted signal.
Many amplifiers are subject to a wide range of ambient operating
temperatures. These temperature changes may affect the operating characteristics of
certain electronic components within the amplifier, thereby inducing additional
distortions. A temperature range of -40°C to +85 °C is not uncommon for many
amplifier applications in a communication environment. To ensure consistent
performance over the operating bandwidth, and to minimize resulting distortions, an
amplifier must be designed for a broad range of ambient operating temperatures.
The distortions created by an amplifier which are of primary concern are
second (even) and third (odd) order harmonic intermodulation and distortions. Prior
art amplifier designs have attempted to ameliorate the effects of even order
distortions by employing push-pull amplifier topologies, since the maximum second
order cancellation occurs when equal amplitude and 180° phase relationship is
maintained over the entire bandwidth. This is achieved through equal gain in both
push-pull halves by matching the operating characteristics of the active devices.
However, odd-order distortion is difficult to remedy. Odd-order distortion
characteristics of an amplifier are manifest as cross modulation (X-mod) and
composite triple beat (CTB) distortions on the signal being amplified. X-mod occurs when the modulated contents of one channel being transmitted interferes with and
becomes part of an adjacent or non-adjacent channel. CTB results from the
combination of three frequencies of carriers occurring in the proximity of each
carrier since the carriers are typically equally spaced across the frequency bandwidth.
Of the two noted distortions, CTB becomes more problematic when increasing the
number of channels on a given CATV system. While X-mod distortion also
increases in proportion to the number of channels, the possibility of CTB is more
dramatic due to the increased number of available combinations from among the
total number of transmitted channels. As the number of channels transmitted by a
communication system increases, or the channels reside close together, the odd-order
distortion becomes a limiting factor of amplifier performance.
There are three basic ways of correcting distortion created by a non-linear
device (NLD): 1) reduce the signal power level; 2) use a feed forward technique;
and 3) use a predistortion or postdistortion technique. The first method reduces the
signal power level such that the NLD is operating in its linear region. However, in
the case of an RF amplifier this results in very high power consumption for low RF
output power.
The second method is the feed forward technique. Using this technique, the
input signal of the main amplification circuit is sampled and compared to the output
signal to determine the difference between the signals. From this difference, the
distortion component is extracted. This distortion component is then amplified by
an auxiliary amplification circuit and combined with the output of the main amplification circuit such that the two distortion components cancel each other.
Although this improves the distortion characteristics of the amplifier, the power
consumed by the auxiliary amplification circuit is comparable to that consumed by
the main amplification circuit. This circuitry is also complex and very temperature
sensitive.
The third method is the predistortion or postdistortion technique. Depending
upon whether the compensating distortion signal is generated before the non-linear
device or after, the respective term predistortion or postdistortion is used. In this
technique, a distortion signal equal in amplitude but opposite in phase to the
distortion component generated by the amplifier circuit is estimated and generated.
This is used to cancel the distortion at the input (for predistortion) or output (for
postdistortion) of the amplifier, thereby improving the operating characteristics of
the amplifier.
One such distortion design, as disclosed in U.S. Patent No. 5,703,530 and
shown in Figure 1, relies upon a traditional π-attenuation network and a delay line
for gain compensation; and a diode pair coupled with a delay line for distortion and
phase compensation. This circuit generates a distortion that is equal in amplitude but
opposite in phase to the distortion introduced by the amplifier. Plots of the
distortions contributed by the distortion generator and the distortions manifest by the
amplifier are shown in Figures 2 and 3. As shown, the distortion signal compensates
for the distortions generated by the amplifier. However, the use of delay lines in
such a manner is impractical since delay lines are physically large, are difficult to adjust and the results are inconsistent across a wide frequency range. Additionally,
both amplitude and phase information are required for correct compensation. The
'530 patent also states that the system disclosed therein is not ideal for certain
application, such as predistortion for CATV RF amplifiers, due to the excessive
losses introduced by the distortion circuit.
An inline predistortion design, as disclosed in U.S. Patent No. 5,798,854,
provides compensation for NLDs by applying a predistorted signal equal in
magnitude but opposite in phase to the distortion produced by the NLD. However,
the circuitry disclosed therein is not matched to the NLD. Additionally, the '854
patent presents a design that is typical of the prior art in the use of a high resistance
bias for the diodes. This will reduce the correction efficiency and increase the
effects of temperature upon the circuit.
Accordingly, there exists a need for a simple distortion generator which
counteracts the distortion created by an NLD. The circuit should not introduce
additional signal delay and should operate over a wide frequency bandwidth and
wide ambient temperature range.
SUMMARY OF THE INVENTION
The present invention is an in-line predistortion or postdistortion generator
for coupling in-line with an NLD to produce an output signal of useful amplitude,
but with low composite triple beat and cross modulation distortions. The distortion
generator comprises an instant controlled non-linear attenuator which utilizes the non-linear current flowing through a pair of diodes to provide the proper amount of
signal attenuation over the entire frequency bandwidth. The distortion generator
circuitry is always matched to the NLD, thereby ensuring a frequency response that
is predictable and predefined. The distortion generator also includes a temperature
compensation circuit to ensure consistent operation throughout a wide temperature
range.
Accordingly, it is an object of the present invention to provide a temperature
compensated distortion generator which minimizes cross modulation and composite
triple beat distortions manifested by an NLD such as an RF amplifier, a laser diode
or a photodetector.
Other objects and advantages of the of the present invention will become
apparent to those skilled in the art after reading a detailed description of the preferred
embodiment.
BRIEF DESCRIPTION OF THE DRAWINGS
Figure 1 is a schematic diagram of a prior art distortion generator.
Figure 2 is a combination plot of the effect of using the outputs from the prior
art distortion generator shown in Figure 1 with an RF amplifier.
Figure 3 is a combination plot of the effect of using the outputs from the prior
art distortion generator shown in Figure 1 with an RF amplifier.
Figure 4 is schematic diagram of a π attenuator. Figure 5 is a signal diagram of the diode non-linear current caused by the
input voltage.
Figure 6 is a schematic diagram of the preferred embodiment of the distortion
generator of the present invention.
Figure 7 is a schematic diagram of the temperature compensation circuit.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
The preferred embodiment of the present invention will be described with
reference to the drawing figures where like numerals represent like elements
throughout. Although the preferred embodiment of the present invention will be
described, for simplicity of explanation, as being coupled with an RF amplifier, those
skilled in the art would clearly recognize that such a distortion generator could also
be utilized to compensate for distortion in laser transmitters, optical detectors, and
other electronic components which operate over a wide range of frequencies. The
description herein is not intended to be limiting, rather it is intended to be
illustrative.
The present invention will be described with reference to Figure 4, whereby
a π attenuator network 20 is shown. The network 20 comprises a selected
configuration of resistors Zj, Rx, R2, R3, Z0, Rp. The signal source is input at signal
input 30 and the output of the attenuator network 20 is seen across the output 95.
Zj is the source of internal impedance which should be equal to the system
impedance Z0, which is seen across the output 95. In an embodiment of the invention for use with a CATV system, the impedance values Z, and Z0 are equal to
75 Ohms. Three of the resistors Rx, R2, R3 form a π attenuator configuration.
Preferably, the values (Y) of resistors R2 and R3 are equal, and substantially larger
than the value (X) of resistor R, . Resistor Rp is connected in parallel with resistor
As one skilled in the art would clearly recognize, when the following
condition is satisfied:
X = 2Z0 2 Y/ (Y2 - Z0 2) Equation (1)
the attenuator network 20 is matched at input and output, from D.C. to very high
frequencies. For one example of the attenuator when X = 7.5 and Y = 1.5K, the
power attenuation A for this attenuator network 20 is:
2 (YZo / (Y + Zo) + X )Y (YZo / (Y + Zo)) Equation (2)
(Y + X + Y Zo / (Y + Zo)) (X + (YZo / (Y + Zo)))
A = ( )2
(YZo / (Y + Zo) + X )Y
Y + X + YZo / (Y + Zo)
Under the condition when Z0 « Y, (as is the case when X = 7.5 and Y = 1.5K):
A * (2 Z0 / (2 Z0 + X))2 Equation (3)
A (dB) = 10 lg A Equation (4)
When X = 7.5 and Y = 1.5k, A (dB) = 0.42dB. This means the attenuator network
20 has very low insertion losses and a good frequency response. When X has a small
variation due to the parallel of R , shown in Figure 4, from Equation (3) Delta X
D elta A (dB ) = - 8.68 Equation (5) 2Zo + X
XR, X_
Delta X = X = Equation (6)
X + RP RP
From Equation (6):
X
D elta A (dB ) ≡ 8.68 Equation (7)
2Zo Rt
For example, If R = 375 ohms then:
7.5 7.5 D elta A (dB ) = 8.68 ———— = 0.00868dB Equation (8)
Equation (8) shows that when Rp (375 ohms) is in parallel with Wx (7.5 ohms), the
attenuation will be reduced by 0.00868dB. This amount of attenuation change is
needed for non-linear compensation for an amplifier. This example also shows that
when the value of Rp » R,, (i.e., when Rp is 50 times larger than R,), adding Rp
parallel with R, has almost no effect on the impedance match, and the voltage drop
over the Rp is mainly determined by the value of R,.
However, if a linear resistor Rp is used in the attenuator network 20, there will
be no distortion signal produced. The attenuator network 20 as shown is a linear
device. In order for a distortion circuit to operate effectively, diodes are used to
create a non-linear resistance. Preferably, Schottky diodes are utilized. At small current, diode current is exponentially proportional to the voltage across over the
diode. Thus diodes can be used as a non-linear resistance. For non-linear
applications, the amount of attenuation can be calculated as:
D elta A (dB ) = 8.68 Equation (9)
Where Ip is the current flow through Rp, (the non-linear resistance). Ij is the
current flow through R, . Equation 9 provides the relationship of the attenuation
change due to the current change in Ip. This equation is accurate over a broad
frequency range. The relationship between the delta attenuation and a change in
current is still valid when the resistance is a non-linear resistor. Accordingly,
Equation 9 provides a good estimation of how much non- linear current is required
for predistortion or postdistortion purposes.
Referring to Figure 5, when the input sinusoidal voltage wave changes from
V, to V2 to V3, the output current changes from I, to I2 to I3 respectively. The non¬
linear current used for third order correction is:
I nonlinear = Ii - 2 h + h Equation (10)
From Equation 9, the non-linear current needed is:
X 1 nonlinear
D elta A nonlinear correction (dB ) ≡ 8 .68 - Equation (11)
2Zθ I output
Only non-linear current will be useful for predistortion or postdistortion
purposes. Equation 11 can be rewritten in the form of : Inon near eff D elta A nonlinear correction (dB ) = 8.68 Equation (12) loutput
Inonhnear
In onlmear eff Equation (13)
Accordingly, Inon.lιnear eff in Equation 12 is the effective non-linear current
going to the output port 114 which is shown in Figure 6. Ioutput in Equation 12 is the
total current that goes to the output port 114. Equation 13 shows that only a small
part of the non-linear diode current is effectively being used for correction.
The π attenuator network 20 has low insertion loss and the voltage drop of
the input voltage on Rx (shown in Figure 4) is proportional to the input voltage.
This voltage may be used to drive a pair of diodes to produce non-linear current.
The non-linear current flowing in the diodes will cause an attenuator to provide less
attenuation at larger RF amplitudes, (i.e. when the input signal has a higher power).
This may be used to compensate for the signal compression caused by amplification.
Because of the relatively high value of the diode's non-linear resistance, the match
of the attenuator network is almost unchanged. This match will not be changed even
over temperature. Additionally, frequency response over multi-octave frequency
bands is favorable.
Referring to Figure 6, the preferred embodiment of the attenuator 100 for
predistortion and postdistortion is shown. The attenuator 100 of the present
invention includes several additional components that modify a traditional π attenuator to achieve significantly better performance over a wide frequency and
temperature range. The attenuator 100 has an input port 101, an output port 114 and
a bias control port 116. The attenuator 100 may be used in a predistortion
configuration with an amplifier or in a postdistortion configuration. For a
predistortion configuration, the output port 114 is connected to the input of an
amplifier. For the postdistortion configuration as shown in Figure 6, an output
signal generated by an amplifier, is applied to the input port 101. The attenuator 100
includes resistors 105, 106, 107, 108, 112; capacitors 102, 103, 104, 111, 113, 115;
diodes 109, 110, and an inductor 117.
The inductor 117 is used in series with the resistor 108. The function of the
inductor 117 is to make a parallel resonance circuit with the forward biased diode
capacitor. At the resonance frequency, the capacitance of the diode will be
compensated by the inductor 117 so that the impedance between points 118 and 119
will be pure resistive and can be calculated as follows:
R impedance between 118, 119 = L / (C * R ) ; Equation (14)
where L is the inductance of 117 in Henrys; C is the total forward biased capacitor
in Farads; and R is the resistance 108 in Ohms. By carefully controlling L and C,
one may get the following:
R impedance between 118. 119 = R E UaUOn (15)
This means the capacitive effect has been totally canceled and an ideal pure resistive
load over a very wide frequency range has been achieved. The function of the resistors 105, 106, 107, 108, 112 and the capacitors 102,
103, 104, 111, 113, 115 is to form a modified π attenuation network in comparison
to the π attenuation network 20 shown in Figure 4. The capacitors 102, 103, 104,
111, 113, and 115 are also used for D.C. blocking and AC coupling. From an AC
standpoint, the parallel combination of resistors 105 and 106 is functionally
equivalent to resistor R2 of Figure 4. Preferably, the values of resistors 105 and 106
should be chosen such that the parallel combination is equivalent to the value of
resistance of resistor 112, (i.e. ((R^RIOOVC IOS+RIOO)) = R-m)- Resistor 108 is
functionally equivalent to resistor Rα of Figure 4; and the in-series combination of
resistor 112 and capacitor 111 is functionally equivalent to resistor R3 of Figure 4.
The value of resistor 107 has no effect on RF signal attenuation.
The other function for resistors 105, 106, and 107 is to supply a D.C. bias to
the diodes 109, 110. The diodes 109, 110 are first connected in series; and the series
combination is connected to resistor 107 in parallel. Because resistor 107 has a low
resistance value and is in parallel with the diodes 109, 110, the voltage drop across
the diodes 109, 110 will be primarily determined by the resistance of resistor 107.
If the D.C. current flow in resistor 107 is much more than the current flow in the
diodes 109, 110, D.C. the voltage drop across the diode 109, 110, will be very stable
and will be insensitive to the presence or absence of a signal at the input port 101.
The integrated functions of signal attenuation and diode bias supply avoid any
parasitic effects due to the introduction of additional bias circuitry. This permits a
high frequency response and a favorable impedance match. From a D.C. perspective, resistor 107, in parallel with capacitors 103 and 104,
provides a dissipative circuit to the capacitors 103, 104. Therefore, resistor 107 will
discharge the accumulated electric charge of connected capacitors 103, 104 in every
AC cycle.
Diode 109 is connected to resistor 108 through capacitor 104 while diode 110
is connected to resistor 108 through capacitor 103. Diode 109 is responsible for the
RF distortion correction during the negative portion of the AC cycle, while the diode
110 has the same function during the positive half of the AC cycle. The non-linear
current of diode 109 charges capacitor 104, and the non-linear current of diode 110
charges capacitor 103. Due to the configuration of the circuit, the voltage produced
on capacitors 103 and 104 have the same value but different signs. The small
resistance from resistor 107 connected to the capacitors 103, 104 discharges the
accumulated electric charge during every AC cycle. As a result, there is no
additional D.C. voltage drop across the capacitors 103, 104 due to the input RF
signals. This permits the diode 109, 110 to provide the largest non-linear current for
the correction purpose.
The present invention has several unique advantages over the prior art. Due
to its symmetric structure, the attenuator 100 produces only odd order distortion.
Consequently, the circuit does not degrade the second order performance of an NLD.
The attenuator 100 also uses two low series resistances 107, 108. From a D.C.
perspective, resistor 107 significantly improves the correction efficiency and reduces
the susceptibility to ambient temperature effects. From an AC perspective, resistor 108 provides for distortion correction with low insertion losses. Due to the
attenuator 100 design, the voltage drop across resistor 108 fully loads the diodes 109,
110 even under non-linear operation of the diodes 109, 110. As a result, maximum
non-linear current is utilized for correction purposes. Finally, proper phasing of the
distortion signals is inherent in the design, thereby avoiding additional phase
circuitry and delay lines. This permits a circuit design which is much less complex
and results in a compact and robust design.
The present attenuator design uses low series resistance 108. The inductor
117 in series with the resistance 108 compensates for the capacitance of the diodes.
Thus, the third order correction circuit may work over a wide frequency range and
wide temperature range. This correction circuit design is flexible and may be
adjusted to different kinds of RF hybrids with different distortion characteristics.
This circuit is always matched to its input side and output side over wide frequency
range.
Table 1 provides a listing of the components shown in Figure 6. However,
one skilled in the art would clearly recognize that the values shown in Table 1 are
only for explanatory purposes, and should not be considered to be limiting to the
invention. For example, the value of resistor 108 may range from approximately 2Ω
to 30Ω. Likewise, the value of resistor 107 may range from approximately 100Ω to
3000Ω. TABLE 1
As previously described, the attenuator 100 uses the non-linear current
produced by the diodes 109, 110 to compensate for the voltage compression caused
by an NLD. As shown, the attenuator 100 comprises capacitance, resistance and two
diodes. The diodes are the only components that are sensitive to temperature change
and the only components that require correction during operation over a wide
temperature range. There are three factors which must be taken into consideration
when operating the attenuator 100 over a wide temperature range: 1) The diode operating current will change if the bias voltage remains
constant while the ambient temperature changes. Under the same input voltage swing
at the input port 101 and the same bias voltage, more non-linear diode current will
be created as the ambient temperature rises.
2) When the ambient temperature rises, the diode will produce less non¬
linear correction current for the same input signal voltage and the same diode bias
current.
3) NLDs typically exhibit more distortion as the ambient temperature rises.
Accordingly, a higher diode non-linear current is required for correction of the
greater distortion.
All of the temperature effects experienced by the attenuator 100 are related to
the bias voltage. Some of the effects are additive while others are subtractive.
However, the result is that for a given temperature, there will be an optimum bias
voltage to produce the proper correction output. Proper temperature correction will
be achieved when there is a predefined change of bias voltage verses temperature.
Referring to Figure 7, the preferred embodiment of the temperature
compensation circuit 200 is shown. The temperature compensation circuit 200
controls the bias of the diodes 109, 110 (shown in Figure 6) for optimum
compensation of the distortion. As shown, the temperature compensation circuit 200
comprises two transistors 206, 213; a capacitor 216; nine resistors 201, 202, 203, 204,
207, 209, 210, 214, 215; two diodes 205, 208; and a negative temperature coefficient
thermistor 211. The negative temperature coefficient thermistor 211 is coupled in parallel with
resistor 210 to form a temperature linearized resistance, which is correlated to a
change in temperature. The PNP transistor 206 provides a constant current source
through its collector to the linearized resistor combination 210, 211. The constant
current provided by the PNP transistor 206 induces a linearized voltage change across
the resistor combination 210, 211 as the temperature changes. By adjusting the value
of the variable resistor 202, the amount of constant current through the PNP transistor
206 can be changed. Therefore, the voltage swing over temperature can be changed.
The constant current also passes through the variable resistor 209, thereby creating
a constant voltage drop that is used as a starting bias point for bias voltage
adjustment. By selectively adjusting the resistance of resistors 202 and 209, any
combination of voltage swing and starting bias voltage can be obtained. A NPN
transistor 213, which is an emitter follower transistor, provides the control bias
voltage from line 217 through line 116 to the attenuator 100, as shown in Figure 7.
The two diodes 205 and 208 are used to compensate for the junction voltage of the
two transistors 206, 213 which change over temperature.
Table 2 provides a listing of the components shown in Figure 7. However,
one skilled in the art would clearly recognize that the values shown in Table 2 are
only for example, and should not be considered to be limiting to the invention. TABLE 2
It should be recognized that the present invention provides an instant voltage
controlled non-linear attenuator design combined with a bias supply for optimum
non-linear correction efficiency and bias temperature stability. Even if the
temperature compensation circuit 200 as disclosed herein is not utilized, the preferred
embodiment of the present invention provides adequate distortion correction over a
broad temperature range. When the temperature compensation circuit 200 is utilized, the distortion compensation results can be further improved. Accordingly, a trade off
between the performance of the compensating circuit and the complexity of the
circuit must be weighted.

Claims

What is claimed is:
1. An external distortion control circuit for selective attenuation of a
signal comprising:
a signal input port;
a non-linear circuit coupled to said input port and comprising:
a modified π attenuator network;
a pair of diodes coupled together in parallel and coupled to said
modified π attenuator network; and
a first resistor and a first inductor in series, said series coupling being
coupled in parallel to said coupled diodes; and
an output port for outputting said selectively attenuated signal from said non¬
linear circuit.
2. The distortion control circuit of claim 1 further including a low
resistance D.C. bias voltage circuit.
3. The distortion control circuit of claim 2 further including a temperature
compensation circuit coupled with said bias circuit, for selectively adjusting said
D.C. bias voltage in response to a change in ambient temperature.
4. The distortion control circuit of claim 1 wherein said first resistor
generates a voltage proportional to said input signal; whereby said proportional voltage creates a non-linear current through at least one of said diodes in said pair,
thereby creating a non-linear resistance to selectively attenuate said signal.
5. The distortion control circuit of claim 3 wherein said temperature
compensation circuit comprises:
a constant current source transistor;
a second transistor, coupled to the output of said current source transistor, for
outputting said D.C. bias voltage;
a linearized resistance circuit having a thermistor coupled in parallel to a
second resistor; and
a variable resistor that couples said current source transistor to said linearized
resistance circuit;
whereby the linearized resistance circuit is correlated to a change in ambient
temperature.
6. The distortion control circuit of claim 1 whereby said non-linear circuit
provides selective attenuation of the signal based upon the signal magnitude;
whereby less attenuation is provided for larger signal magnitudes and more
attenuation is provided for smaller signal magnitudes.
EP00919766A 1999-04-01 2000-03-29 Non-linear distortion generator Expired - Lifetime EP1166435B1 (en)

Applications Claiming Priority (3)

Application Number Priority Date Filing Date Title
US282958 1999-04-01
US09/282,958 US6577177B2 (en) 1999-04-01 1999-04-01 Non-linear distortion generator
PCT/US2000/008255 WO2000060734A1 (en) 1999-04-01 2000-03-29 Non-linear distortion generator

Publications (2)

Publication Number Publication Date
EP1166435A1 true EP1166435A1 (en) 2002-01-02
EP1166435B1 EP1166435B1 (en) 2003-09-10

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US (1) US6577177B2 (en)
EP (1) EP1166435B1 (en)
KR (1) KR20010111287A (en)
CN (1) CN1345478A (en)
AT (1) ATE249693T1 (en)
AU (1) AU761636B2 (en)
CA (1) CA2368578A1 (en)
DE (1) DE60005163T2 (en)
ES (1) ES2206218T3 (en)
MX (1) MXPA01009899A (en)
TW (1) TW498603B (en)
WO (1) WO2000060734A1 (en)

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Also Published As

Publication number Publication date
WO2000060734A1 (en) 2000-10-12
DE60005163D1 (en) 2003-10-16
KR20010111287A (en) 2001-12-17
US20010054927A1 (en) 2001-12-27
EP1166435B1 (en) 2003-09-10
MXPA01009899A (en) 2003-07-28
US6577177B2 (en) 2003-06-10
TW498603B (en) 2002-08-11
AU4039700A (en) 2000-10-23
ES2206218T3 (en) 2004-05-16
ATE249693T1 (en) 2003-09-15
DE60005163T2 (en) 2004-07-22
AU761636B2 (en) 2003-06-05
CA2368578A1 (en) 2000-10-12
CN1345478A (en) 2002-04-17

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