MXPA01009895A - Non-linear distortion generator - Google Patents

Abstract

An in-line distortion generator for coupling in-line with a non-linear device (NLD) produces an output signal of useful amplitude, but with low composite triple beat and cross modulation distortions. The distortion generator comprises an instant controlled non-linear attenuator which utilizes the non-linear current flowing through a pair of diodes, in parallel with a resistor and an inductor, to provide the proper amount of signal attenuation over the entire frequency bandwidth. The distorsion generator circuitry is always matched to the NLD, thereby ensuring a frequency response that is predictable and predefined. The distortion generator may also include a temperature compensation circuit to ensure consistent operation throughout a wide temperature range.

Description

NON-LINEAR DISTORTION GENERATOR BACKGROUND OF THE INVENTION FIELD OF THE INVENTION The present invention relates generally to communication systems employing amplification devices. More particularly, the invention relates to a non-linear pre-distortion or post-distortion generator for in-line coupling with an amplifier, optical detector or laser to minimize the third-order signal distortion caused by the processing of the signal. signal.

DESCRIPTION OF THE RELATED ART Amplifiers are widely used in many types of communication applications. Although it is preferable to keep the amplifiers within their linear range of operation, it has been necessary, increasingly, to extend the operation of the amplifiers in regions of high power and high frequency operation. Typically, the output power of an amplifier is limited by the non-linearity of the active devices, including bipolar transistors and FETs. These non-linearities result in distortions that have an impact on the signal that is amplified. The reduction of the non-linear distortions of an amplifier results in increases in the output power, the dynamic range of the system and the carrier-to-noise ratio. Therefore, the minimization of distortions and the achievement of a linear frequency response is greater than the efficient operation of the amplifier. Additionally, optical or laser detectors used within a circuit can also introduce distortions. It is preferable to minimize or eliminate all these types of distortions. The minimization of distortion is particularly important when a series of amplifiers is cascaded over a signal transmission path, such as a series of RF amplifiers in a CATV transmission system. Positioned throughout a CATV transmission system are RF amplifiers that periodically amplify the transmitted signals to counteract cable attenuation and attenuation caused by passive CATV components, such as signal splitters and compensators. RF amplifiers are also used to maintain the desired ratio of carrier to noise. Due to the number of RF amplifiers used in a given CATV transmission system, each RF amplifier must provide minimal degradation to the transmitted signal. Many amplifiers are subjected to a wide range of ambient operating temperatures. These temperatures change to affect the operating characteristics of certain electronic components within the amplifier, thereby inducing additional distortions. A temperature range of -40 ° C to + 85 ° C is not uncommon for many amplifier applications in a communication environment. To ensure consistent performance over the operating bandwidth, and to minimize the resulting distortions, an amplifier must be designed for a wide range of ambient operating temperatures. The distortions created by an amplifier that are of primary interest are the harmonic intermodulation of second (pair) and third (odd) order and distortions. The prior art amplifier designs have attempted to improve the effects of even-order distortions by employing double-effect amplifier topologies, since maximum second-order cancellation occurs when an equal amplitude and a phase-to-phase ratio are maintained. 180 ° over the full bandwidth. This is achieved through equitable gain in both halves of the double effect by matching the operating characteristics of the active devices. However, it is difficult to remedy the distortion of odd order. Odd-order distortion characteristics of the amplifier manifest as cross-modulation (X-od) and triple-pulse distortions, composite (CTB) in the signal that is amplified. The X-mod occurs when the modulated contents of a channel that are transmitted interfere with and become part of an adjacent or non-adjacent channel. The CTB results from the combination of other carrier frequencies that occur in the vicinity of each carrier since the carriers are typically spaced equally across the frequency bandwidth. Of the two distortions noted, the CTB becomes more problematic when the number of channels in a given CATV system is increased. As long as the distortion of X-mod also increases in proportion to the number of channels, the possibility of CATV is more dramatic due to the increased number of available combinations of the total number of channels transmitted. As the number of channels transmitted by a communication system increases, or the channels reside closely together, odd-order distortion becomes a limiting factor of the amplifier's performance. There are three basic ways to correct the distortion created by a non-linear device (NLD): 1) reduce the level of signal power; 2) use an early correction technique; and 3) use a pre-distortion or post-distortion technique. The first method uses the power level of the signal such as the NLD that is operating in its linear region. However, in the case of an RF amplifier, this results in a very high power consumption for a low RF output power. The second method is the error correction technique. Using this technique, the input signal from the main amplification circuit is sampled and compared to the output signal to determine the difference between the signals. From this difference, the distortion component is extracted. This distortion component is then amplified by an auxiliary amplification circuit and combined with the output of the main amplification circuit such that the two distortion components cancel each other out. Although this improves the distortion characteristics of the amplifier, the power consumed by the auxiliary amplification circuit is comparable to that consumed by the main amplification circuit. The circuitry is also complex and very sensitive to temperature. The third method is the pre-distortion or post-distortion technique. Depending on whether the compensation distortion signal is generated before the linear device or thereafter, the respective term, pre-distortion or post-distortion is used. In this technique, a distortion signal equal in magnitude is estimated and generated but phased in to the distortion component generated by the amplifier circuit. This is used to cancel the distortion in the input (for pre-distortion) or output (for post-distortion) of the amplifier, thereby improving the operating characteristics of the amplifier. This distortion design, as described in U.S. Patent 5,703,530 and as shown in Figure 1, depends on a traditional p-attenuation network, and a retrace line for gain compensation; and a pair of diodes coupled with a retrace line for phase compensation and distortion. This circuit generates a distortion that is equal in amplitude, but put in phase to the distortion introduced by the amplifier. The graphs of the distortions that have been contributed by the distortion generator and the distortions exhibited by the amplifier are shown in Figures 2 and 3. As shown, the distortion signal compensates for the distortions generated by the amplifier. However, the use of retrace lines in this manner is impractical since the retrace lines are physically large, difficult to adjust and the results are inconsistent across a wide range of frequencies. Additionally, both phase and amplitude information are required for correct compensation. The '530 patent also points out that the system described herein is not ideal for a certain application, such as pre-distortion for CATV RF amplifiers, due to excessive losses introduced by the distortion circuit. U.S. Patent No. 5,523,716 describes another example of a distortion compensation design; This design is aimed at satellite communication systems. Due to the high power operation range of the satellite system described in the '716 patent, the received RF signal drives the pair of diodes and therefore, a bias circuit is not required. Due to the extremely low signal level for CATV applications, and due to much lower operating frequencies, this design will not operate effectively in the CATV environment. An inline pre-distortion design, as described in U.S. Patent No. 5,798,854, provides the compensation for NLD by applying a predistorted signal that is equitable in magnitude but phased in to the distortion produced by the NLD. However, the circuitry described herein does not correspond to the NLD. Additionally, the '854 patent presents a design that is typical of the prior art in the use of a high strength polarization for the diodes. This will reduce the efficiency of the corrector and will increase the effects of the temperature in the circuit. Accordingly, there is a need for a simple distortion generator that counteracts the distortion caused by an NLD. The circuit must not introduce additional signal delay and must operate over a wide frequency bandwidth and a wide range of ambient temperatures.

BRIEF DESCRIPTION OF THE INVENTION The present invention is an in-line pre-distortion or post-distortion generator for in-line coupling with an NLD to produce a useful amplitude output signal, but with low cross-modulation and triple-pulse distortions. , composed. The distortion generator comprises a non-linear, controlled, instantaneous attenuator which uses the non-linear current flowing through a pair of diodes to provide the appropriate amount of signal attenuation over the complete frequency bandwidth. The distortion generator circuitry is always matched to the NLD, thus ensuring a frequency response that is predictable and predefined. The distortion generator also includes a temperature compensation circuit to ensure consistent operation throughout a wide range of temperatures. Accordingly, it is an object of the present invention to provide a distortion-compensated generator with respect to temperature that minimizes the composite triple-pulse, and cross-modulation distortions evident by an NLD such as an RF amplifier, a laser diode or a photodetector. Other objects and advantages of the present invention will become apparent to those skilled in the art upon reading a detailed description of the preferred embodiment.

BRIEF DESCRIPTION OF THE DRAWINGS Figure 1 is a schematic diagram of a distortion generator of the prior art. Figure 2 is a graph of the combination of the effect of using the outputs of the distortion generator of the prior art shown in Figure 1 with an RF amplifier. Figure 3 is a graph of the combination of the effect of using the outputs of the distortion generator of the prior art shown in Figure 1 with an RF amplifier. Figure 4 is a schematic diagram of an attenuator p. Figure 5 is a signal diagram of the non-linear diode current caused by the input voltage. Figure 6 is a schematic diagram of the preferred embodiment of the distortion generator of the present invention. Figure 7 is a schematic diagram of the temperature compensation circuit.

DESCRIPTION OF THE PREFERRED MODALITIES The preferred embodiment of the present invention will be described with reference to the Figures of the drawings where similar numbers represent similar elements throughout their length. Although the preferred embodiment of the present invention will be described, for simplicity of explanation, as being coupled with an RF amplifier, those skilled in the art will clearly recognize that this distortion generator could also be used to compensate for the distortion in laser transmitters, optical detectors and other electronic components that operate over a wide range of frequencies. The description herein is not intended to be limiting, rather it is proposed to be illustrative. The present invention will be described with reference to Figure 4, whereby an attenuator network 20 is shown p. The network 20 comprises a selected resistance configuration Zi, Ri, R2, R3, Z0, p. The signal source is input to the signal input 30 and the output of the attenuator network 20 is seen through the output 95. Zi is the internal impedance source which must be equal to the impedance Z0 of the system, which is see through output 95. In one embodiment of the invention for use with a CATV system, the impedance values Zi and Zrj are equal to 75 Ohms. Three of the resistors Ri, R2, R form an attenuator configuration p. Preferably, the values (Y) of the resistors P-2 and P-3 are equal, and substantially greater than the value (X) of the resistor Rx. The resistance Rp is connected in parallel with the resistance Ri. As will be clearly recognized by one skilled in the art, when the following condition is satisfied: X = 2Z02Y / (Y2 Zr Equation (1) the attenuator network 20 is matched at the input and output, of DC at very high frequencies. For an example of the attenuator, when X = 7.5 and Y = 1.5K, the attenuation A of power for this attenuator network 20 is: 2 (YZo / (Y-rZo) -X) Y (YZo / (Y + Zo) ) Equation (2) (Y + X + YZo / (Y + Zo)) (X + (YZo / (Y + Zo))) (YZo / (Y + Zo) + X) And Zo + Y + X + YZo / (Y + Zo) Under the condition when Z0 «Y, (as is the case when X = 7.5 and Y = 1.5K): A = (2 Z0 / (2 Z0 + X)) 2 Equation (3) A (dB) = 10 lg A Equation (4) When X = 7.5 and Y = 1.5 k, A (dB) = 0.42dB). This means that the attenuator network 20 has very low insertion losses and a good frequency response. When X has a small variation due to the parallel of Rp, shown in Figure 4, of Equation (3): Delta X Delta A (dB) = -8.68 2Zo + X Equation (5) 2 Delta X Equation (6) From Equation (6) X 'Delta A (dB) = 8.68 2Zo RP Equation (7) For example, if Rp = 375 ohms, then Delta A (dB) s 8.68 T ^^ r = 0.00868dB Equation (8) Equation (8) shows that when Rp (375 ohms) is in parallel with i (7.5 ohms), the attenuation will be reduced by 0.00868d B. This amount of attenuation change is needed for non-linear compensation for an amplifier. This example also shows that when the value of Rp is > > R ?, (that is, when Rp is 50 times greater than Ri), the addition of Rp parallel with R.? it does not always have the effect on the impedence correspondence, and the voltage drop with respect to Rp is mainly determined by the value of Ri. However, if a linear resistance Rp is used in the attenuator network 20, there will be no distortion signal produced. The attenuator network 20 as shown is a linear device. In order for an extrusion circuit to operate effectively, diodes are used to create a non-linear resistance. Preferably, Schottky diodes are used. At a small current, the diode current is exponentially proportional to the voltage across the diode. In this way, the diodes can be used as a non-linear resistor. For non-linear applications, the amount of attenuation can be calculated as: / Delta A (dB) = 8.68 Equation (9) Where Ip is the current flow through Rp, (the non-linear resistance), It is the current flow through Ri. Equation 9 gives the ratio of the change in attenuation due to the current change in Ip. This equation is accurate over a wide range of frequencies. The relationship between delta attenuation and a change in current is still valid when the resistance is a non-linear resistance. Accordingly, Equation 9 provides a good estimate of how much non-linear current is required for pre-distortion or post-distortion purposes. With reference to Figure 5, when the input sinusoidal voltage wave changes from Vi to V2, to V3, the output current changes from Ii to I / a I3, respectively. The non-linear current used for the third order correction is: I non-linear = II 212 13 Equation (10) From Equation 9, the necessary non-linear current is Non-linear correction Delta A (dB) = 8.68 X I non-linear Equation (11) 2Z01 output Only non-linear current will be useful for pre-distortion or post-distortion purposes. Equation 11 can be rewritten in the form of: Non-linear correction Delta A (dB) = 8.681 non-linear eff Equation (12) I output I non-linear eff = I non-linear R? / (2Z0) Ecuaaon (13) Therefore, I non-linear eff in the Equation 12 is the effective non-linear current going to the output port 114 shown in Figure 6. I output in Equation 12 is the total current going to the output port 114. Equation 13 shows that only a small part of the non-linear diode current is being used effectively for correction. The attenuator network p has low insertion loss and the voltage drop of the input voltage in Ri (shown in Figure 4) is proportional to the input voltage. This voltage can be used to drive or boost a pair of diodes to produce non-linear current. The non-linear current that influences the diodes will cause an attenuator to provide less attenuation at higher RF amplitudes, (that is, when the input signal has a greater power). This can be used to compensate for the signal compression caused by the amplification. Due to the relatively high value of the non-linear resistance of the diodes, this almost unchanged the matching of the attenuator network. This correspondence will not change even with respect to temperature. Additionally, the frequency response with respect to the multi-octave frequency bands is favorable. With reference to Figure 6, the preferred mode of the attenuator 100 for pre-distortion and post-distortion is shown. The attenuator 100 of the present invention includes several additional components that modify a traditional p attenuator to achieve significantly better performance over a wide range of frequencies and temperatures. The attenuator 100 has an input port 101, an output port 114, and a polarization control port 116. The attenuator 100 may be used in a pre-distortion configuration with an amplifier or in a post-distortion configuration. For a pre-distortion configuration, the output port 114 is connected to the input of an amplifier. For the post-distortion configuration, as shown in Figure 6, an output signal generated by an amplifier is applied to the input port 101. The attenuator 100 includes the resistors 105, 106, 107, 108, 112; the capacitors 102, 103, 104, 111, 113, 115; the diodes 109, 110, and an inductor 117. In most prior art applications, an inductor with a phase control element is used to change the correction signal phase. However, in the present invention the inductor 117 is used in series with the resistor 108 to make a parallel resonant circuit with the direct polarized diode capacitor. The inductive reactance cancels the specific capacitive reactance of the diodes. At the resonance frequency, the capacitance of the diodes 109, 110 will be compensated by the inductor 117 so that the impedance between points 118 and 119 will be purely resistive and can be calculated as follows: R impedance between 118, 119 = L / (C * R); Equation (14) where L is the inductance of 117 in Henrys; C is the direct polarized capacitor, total in Faradios; and R is the resistance 108 in Ohms. By carefully controlling L and C, you can obtain the following: R impedance between 118, 119 = R Equation (15) This means that the capacitive effect is completely canceled and an ideal pure resistive load over a very wide frequency range has been achieved. In the systems of the prior art, the capacitance associated with. the diodes has not been considered. In pre-distortion applications, Shottkey diodes are directly biased, resulting in higher capacitance. When an RF signal is introduced through the diodes, the average capacitance is increased. Even at a 0 volt polarization, the capacitance introduced by the capacitance of the diodes can not be ignored since the capacitance in parallel with the PN junction of the diodes will reduce the total voltage drop in the diodes, thus reducing the current nonlinear produced by the diodes and the total correction effect. The compensation of the capacitance associated with the diode 109, 110, the inductor 117 reacts with the capacitance of the diodes 109, 110 at higher RF frequencies, thereby extending the overall frequency response of the circuit. The function of the resistors 105, 106, 107, 108, 112 and of the capacitors 102, 103, 104, 111, 113, 115 and the inductance 117 is to form a modified attenuation network p compared to the attenuation network p shown in Figure 4. Capacitors 102, 103, 104, 111, 113 and 115 are also used for DC blocking and AC coupling. From an AC point of view, the parallel combination in the resistance 105 and 106 is functionally equivalent to the resistance R in Figure 4. Preferably, the values of the resistances 105 and 106 must be chosen such that the parallel combination is equivalent to the resistance value of resistance 112 (ie ((R? o5 * R? o6) / (E-105 + P-106)) = R? 2) - Resistance 108 is functionally equivalent to resistance Ri of Figure 4; and the series combination of the resistor 112 and the capacitor 111 is functionally equivalent to the resistor R3 of Figure 4. The value of the resistor 107 has no effect on the RF signal attenuation.

The other function for resistances 105, 106, and 107 is to provide a DC bias to the diodes 109, 110. The diodes 109, 110 are first connected in series; and the series combination is connected to resistor 107 in parallel. Because the resistor 107 has a low resistance value and is in parallel with the diodes 109, 110, the voltage drop across the diodes 109, 110 will be determined primarily by the resistance of the resistor 107. If the DC current at resistor 107 is much greater than the current flow at diodes 109, 110, the DC voltage drop across diodes 109, 110, will be very stable and will be insensitive to the presence or absence of a sign in port 101 entrance. The integrated functions of the signal attenuation and the diode polarization supply avoids any parasitic effect due to the interruption of additional biasing circuitry. This allows a high frequency response and a favorable impedance match. From a DC perspective, the resistance 107, in parallel with the capacitors 103 and 104, provides a dissipative circuit to the capacitors 103, 104. Therefore, the resistor 107 will discharge the accumulated electrical charge of the connected capacitors 103, 104 in each AC cycle. The diode 109 is connected to the resistor 108 through the capacitor 104 while the diode 110 is connected to resistor 108 through the capacitor 103. The diode 109 is responsible for the RF distortion correction during the negative portion of the cycle. AC, while diode 110 has the same function during the positive half of the AC cycle. The non-linear current of the diode 109 charges the capacitor 104, and the non-linear current of the diode 110 charges the capacitor 103. Due to the configuration of the circuit, the voltage produced in the capacitors 103 and 104 has the same value but different signals. The small resistance of resistor 107 connected to capacitors 103, 104, discharge the accumulated electrical charge during each AC cycle. As a result, there is no additional DC voltage drop across the capacitors 103, 104 due to the input RF signals. This allows diode 109, 110 to provide the largest non-linear current for the purpose of correction. The present invention has several unique advantages over the prior art. Due to its symmetrical structure, the attenuator 100 produces only odd-order distortion. Consequently, the circuit does not degrade the second-order performance of an NLD. The attenuator 100 also uses two resistors 107, 108 of low series. The resistance 107 significantly improves the correction efficiency. The resistor 108 provides correction and distortion with low insertion losses. Due to the design of the attenuator 100, the voltage drop across the resistor 108 fully charges the diodes 109, 110 to a low non-linear operation of the diodes 109, 110. As a result, a maximum non-linear current is used for the purposes of correction. The present attenuator design uses the low series resistor 108, in series with the inductor 117 to compensate for the capacitance of the resistors. diodes 109, 110. In this way, this circuit can work over a wide range of frequencies. This correction circuit design is flexible and can be adjusted to different kinds of RF hybrids with different distortion characteristics. This circuit is always matched to its input side and output side over a wide frequency range. Finally, the proper mapping of the distortion signals is inherent in the design, thus avoiding additional phase circuitry and retrace lines. This allows a circuit design 4 which is much less complex and which results in a compact and strong design. Table 1 provides a listing of the components shown in Figure 6. However, one skilled in the art will recognize that the values shown in Table 1 are for explanatory purposes only, and should not be construed as limiting the invention. For example, the value of the resistance 108 may vary from about 2O to 30O. Similarly, the value of the resistance 107 may vary from about 100O to 3000O.

TABLE 1 As previously described, the attenuator 100 uses the non-linear current produced by the diodes 109, 110 to compensate for voltage compression caused by an NLD. As shown, the attenuator 100 comprises capacitance, resistance and two diodes. Diodes are the only components that are sensitive to temperature change and the only components that require correction during operation over a wide range of temperatures. There are three factors that must be taken into consideration when operating the attenuator 100 over a wide temperature range: 1) The operating current of the diode will change if the bias voltage remains constant as the ambient temperature changes. Under the same input voltage that oscillates at input port 101 and the same bias voltage, more non-linear diode current will be created as the ambient temperature increases. 2) When the room temperature increases, the diode will produce less non-linear correction current for the same input signal voltage and the same diode bias current. 3) NLD typically exhibit more distortion as room temperature increases. Therefore, a higher non-linear diode current is required for the correction of the largest distortion. All temperature effects experienced by the attenuator 100 are related to the bias voltage. Some of the effects are additive while others are subtractive. Nevertheless, the result is that for a given temperature, there will be an optimal polarization voltage to produce the appropriate correction output. Proper temperature correction will be achieved when there is a predefined change in bias voltage against temperature. With reference to Figure 7, it is shown in the preferred embodiment of the temperature compensation circuit 200. The temperature compensation circuit 200 with full polarization of the diodes 109, 110 (shown in Figure 6) for optimal compensation of the distortion. As shown, the temperature compensation circuit 200 comprises two transistors 206, 213; a capacitor 216; nine resistors 201, 202, 203, 204, 207, 209, 210, 214, 215; two diodes 205, 208; and a thermistor 211 of negative temperature coefficient. The negative temperature coefficient thermistor 211 is coupled in parallel with the resistor 210 to form a linearized resistance at the temperature, which correlates to a change in temperature. The PNP transistor 206 provides a constant source of current through its collector to the combination 210, 211 of linearized resistance. The constant current provided by the PNP transistor 206 includes a linearized voltage change through the resistance combination 210, 211 as the temperature changes. By adjusting the value of the variable resistor 202, the amount of constant current can be changed through a PNP transistor 206. Therefore, the voltage that oscillates over the temperature can be changed. The constant current also passes through the variable resistor 209, thereby creating a constant voltage drop that is used as a starting bias point for the bias voltage setting. By selectively adjusting the resistance of the resistors 202 and 209, any combination of voltage that oscillates and initiates the bias voltage can be obtained. An NPN transistor 213 which is a transmitter of a follower emitter cover, provides the control bias voltage from line 217 through line 116 to alternator 100, as shown in Figure 7. The two diodes 205 and 208 they are used to compensate for the junction voltage of the two transistors 206, 213 that change with respect to the temperature. Table 2 provides a listing of the components shown in Figure 7. However, one skilled in the art will clearly recognize that the values shown in Table 2 are exemplary only, and should not be construed as limiting this invention. TABLE 2 It should be recognized that the present invention provides a voltage-controlled, instantaneous, non-linear attenuator design combined with a bias supply for optimal non-linear correction efficiency and bias temperature stability. Even if the temperature compensation circuit 200 as it is described herein is not used, the preferred embodiment of the present invention provides an adequate distortion correction with respect to a wide range of temperatures. When the temperature compensation circuit 200 is used, the results of the distortion compensation can be further improved. Therefore, an exchange between the performance of the compensation circuit and the complexity of the circuit can be weighted.

Claims (5)

1. CLAIMS 1. An external distortion control circuit for use in a CATV system for the selective attenuation of a CATV signal, comprising: a signal input port; a non-linear circuit coupled to the input port and comprising: a modified attenuator network p comprising first and second resistors coupled in parallel; the first and second resistors coupled in series with a third resistor, a first inductor and a fourth resistor; a first and a second diode coupled each in parallel with the third resistor and the first inductor; a fifth resistance coupled to both diodes; and an output port for transferring the selectively attenuated signal from the non-linear circuit; So the first, second and fifth resistors provide a polarization voltage of D.C. through the diodes.
2. 2. The distortion control circuit of claim 1, further including a temperature compensation circuit, for selectively adjusting the bias voltage of D.C. in response to a change in the ambient temperature.
3. The distortion control circuit according to claim 1, wherein the third resistor and the first inductor generate a voltage proportional to the input signal; whereby the proportional voltage creates a non-linear current through at least one of the diodes, thereby creating a non-linear resistance to selectively attenuate the signal.
4. The distortion control circuit according to claim 2, wherein the temperature compensation circuit comprises: a constant current source transistor; a second transistor, coupled to the output of the current bridge transistor, for transferring the DC bias voltage; a linearized resistance circuit having a thermistor coupled in parallel to a second resistor; and a variable resistance that couples the current source transistor to the linearized resistance circuit; so the linearized resistance circuit correlates to a change in the ambient temperature.
5. 5. The distortion control circuit according to claim 1, whereby the non-linear circuit provides selective attenuation of the signal based on the signal magnitude; so less attenuation is provided for large signal magnitudes and less attenuation is provided for lower signal magnitudes.