EP0255844A1 - Sources d'alimentation comportant un amplificateur magnétique de régulation de la tension - Google Patents

Sources d'alimentation comportant un amplificateur magnétique de régulation de la tension Download PDF

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Publication number
EP0255844A1
EP0255844A1 EP86110982A EP86110982A EP0255844A1 EP 0255844 A1 EP0255844 A1 EP 0255844A1 EP 86110982 A EP86110982 A EP 86110982A EP 86110982 A EP86110982 A EP 86110982A EP 0255844 A1 EP0255844 A1 EP 0255844A1
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EP
European Patent Office
Prior art keywords
winding
voltage
core
reactor
power supply
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Granted
Application number
EP86110982A
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German (de)
English (en)
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EP0255844B1 (fr
Inventor
Jerry Kyle Radcliffe
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International Business Machines Corp
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International Business Machines Corp
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Filing date
Publication date
Priority to US06/762,648 priority Critical patent/US4642743A/en
Application filed by International Business Machines Corp filed Critical International Business Machines Corp
Priority to DE8686110982T priority patent/DE3671553D1/de
Priority to EP19860110982 priority patent/EP0255844B1/fr
Publication of EP0255844A1 publication Critical patent/EP0255844A1/fr
Application granted granted Critical
Publication of EP0255844B1 publication Critical patent/EP0255844B1/fr
Expired legal-status Critical Current

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    • G—PHYSICS
    • G05—CONTROLLING; REGULATING
    • G05F—SYSTEMS FOR REGULATING ELECTRIC OR MAGNETIC VARIABLES
    • G05F1/00—Automatic systems in which deviations of an electric quantity from one or more predetermined values are detected at the output of the system and fed back to a device within the system to restore the detected quantity to its predetermined value or values, i.e. retroactive systems
    • G05F1/10—Regulating voltage or current
    • G05F1/12—Regulating voltage or current wherein the variable actually regulated by the final control device is ac
    • G05F1/32—Regulating voltage or current wherein the variable actually regulated by the final control device is ac using magnetic devices having a controllable degree of saturation as final control devices
    • G05F1/34—Regulating voltage or current wherein the variable actually regulated by the final control device is ac using magnetic devices having a controllable degree of saturation as final control devices combined with discharge tubes or semiconductor devices
    • G05F1/38—Regulating voltage or current wherein the variable actually regulated by the final control device is ac using magnetic devices having a controllable degree of saturation as final control devices combined with discharge tubes or semiconductor devices semiconductor devices only

Definitions

  • This invention relates to power supplies and, more particularly, to power supplies of the switching converter type using magnetic amplifier or saturable reactor voltage regulating means.
  • Switching power supplies are frequently used to provide a plurality of separate outputs, and independent control of the outputs is often required. This has usually been done in the past by packaging multiple power stages in one unit and using a separate control loop for each output. A separate switching power stage and transformer are needed for each controlled output. This solution works well but is expensive.
  • prior art converters of this type use a switch transistor to drive the primary of a transformer which has two output windings.
  • One of the output windings feeds a rectifier and filter to supply a first output voltage which is sensed by a control circuit which adjusts the operating duty cycle of the switch to maintain the output voltage at a preset value.
  • the second output winding is connected to a rectifier and filter through a saturable reactor having a core with a square B-H loop. When the core is not saturated, the reactor exhibits a high impedance and prevents the voltage in the second output winding from reaching the rectifier and filter.
  • the voltage will cause the core to saturate after a period of time determined by the starting point on the B-H loop and the applied voltage.
  • the reactor switches to a low impedance value permitting the voltage to be applied to the rectifier and filter.
  • a control circuit forces the reactor to reset with a current which is poled in a direction opposite to the direction of the current during the active conduction period.
  • the reset point is adjusted in response to the output voltage to maintain the output voltage at a preset value.
  • a magnetic amplifier voltage control system use relatively inexpensive substantially zero remanent, moderately low permeability cores with a B-H characteristic having poor squareness.
  • the reset point of the core is controlled by adjusting a current supplied during a control half-cycle in a control circuit including a variable resistance, a rectifier and a control winding for the saturable reactor.
  • the present invention contemplates the provision of a power supply in which the reset point of the saturable core is established by providing clamping means, including a transistor switch and diode in series, effectively to clamp a short circuit across the reactor winding when the core reaches a desired reset point on its B-H loop.
  • a control circuit includes a comparator for generating a control signal for actuating the switch whenever an error voltage derived from the output voltage of the power supply and a reference voltage exceeds the voltage level of a triangular wave developed by integrating a replica of the input pulsations of the power supply.
  • the power supply of the invention includes a source of alternating positive and negative voltage pulse- tions, such as square pulses of the type typically provided by a switching inverter, which are applied through a transformer to a secondary winding.
  • the saturable reactor winding which is connected between the secondary winding and rectifier and filter means providing the direct current output voltage, is driven to saturation at a time during the positive pul- sations related to the position of the reset point.
  • the control circuit controls the duty cycle of the voltage applied to the rectifier means. Since the operation of the control circuit depends on the magnitude of the error voltage, the circuit operates to maintain the output voltage at a preset value.
  • the diode in series with the transistor switch is poled to block current during positive pulsations.
  • the control circuit includes an integrator and a baseline clipper diode which prevents the generated triangular wave from becoming negative.
  • the error voltage is generated by an error circuit comprising a differential amplifier, the inputs of which receive the output voltage and a reference voltage.
  • the above-described technique for controlling voltage may be used in a switching converter having a plurality of secondary windings each feeding a rectifier-filter.
  • the output of one of the rectifier filters may be controlled as described above, while another output may be used to control a pulse width modulator controlling the duty cycle of the switching transistor driving the transformer primary winding.
  • the clamping switch may be connected directly across the reactor winding or may be effectively connected across the reactor winding by being connected across a secondary reactor winding coupled to the reactor winding.
  • a bias winding may also be provided to apply a bias signal for shifting the B-H characteristic of the core to the right to increase the range of adjustment.
  • a prior art converter circuit is shown in Fig. 1.
  • a switching inverter 10 includes a direct current supply 11, which drives a primary winding 12 of power transformer T1 through a transistor switch Q1.
  • the transformer has two secondary windings N s1 and N s2 .
  • the voltage appearing across winding N s1 is rectified and filtered in the conventional manner.
  • the rectifier 14, includes a series diode 16 and a shunt diode 18; and a low pass filter 20 includes a series inductor L1 and a shunt capacitor C1 to remove alternating current ripple components to provide a first output direct current voltage V o1 .
  • a control circuit 21 is a conventional pulse width modulator error circuit providing a control pulse on output 22.
  • the control pulse controls the switching of tran­sistor switch Q1 and thus adjusts the duty cycle of inverter 10. This adjustment maintains output voltage V o1 at a preset value, voltage V o1 being proportional to the duty cycle of inverter 10.
  • the second output winding N s2 of transformer T1 is connected to an identical rectifier 14' and low pass filter 20' through a saturable reactor L s1 which comprises a reactor winding 22 and a saturable core 24 formed of a material, such as tape wound permalloy, and an ungapped toroidal structure providing a highly square characteristic.
  • the square B-H hysteresis loop of core 24 is shown in Fig. 2 in which, in the usual manner, B re­presents magnetic flux density and H signifies magnetizing force.
  • the core is reset to a point a in the left hand plane, which consists of the upper left and lower left quadrants, of the characteristic.
  • a reset control circuit 28 senses output voltage V o2 and generates a control current which is a function of the difference between output voltage V o2 and a reference voltage.
  • the reset control current which is opposite the polarity to the current through reactor winding 22 during the period of conduction of transistor switch Q1, forces reactor core 24 to reset to point a of the characteristic so that the reactor will be ready for the next pulse.
  • the controlled range of adjustment of the flux density is shown as B in Fig. 2.
  • Saturable reactor L s1 acts to shrink the pulse from transistor Q 1 by an amount controlled by the location of the reset point a to maintain output voltage V o2 at a preset value.
  • Fig. 1 Although the prior art circuit of Fig. 1 is effective, it requires a saturable reactor core made of a square loop material having a high degree of squareness. Expensive metal tape-wound cores using permalloy or lossy square loop ferrites may be used, and ungapped toroidal structures which are expensive to wind and difficult to mount are needed.
  • an inverter 110 includes a direct current supply 111 which drives a primary winding 112 of a transformer T2 through a transistor switch Q1.
  • Transistor Q1 is turned on (become conductive) in response to a signal applied to its base electrode on lead 22 from pulse width control 21.
  • Inverter 110 thus operates as a switching inverter, generating a square wave, the pulse width of which is responsive to the pulse width control circuit 21.
  • a first output winding N1 developes a voltage whenever transistor Q1 is conducting.
  • This voltage is rectified in rectifier 14 which includes a series diode 16 and a shunt diode 18.
  • the rectified voltage is then passed through a low pass filter 20, which includes a series inductor L1 and a shunt capacitor C1, to remove the A>C> ripple component and apply a direct current output voltage V o1 across output terminals 23.
  • the output voltage V o1 is applied to pulse width control circuit 21 which compares it to a reference voltage to develop a pulse width control signal in a manner known in the art.
  • This pulse width control signal is connected to the base electrode of transistor Q1 to control the duty cycle of inverter 110 and maintain output voltage V o1 at a present value.
  • a second output winding N2 of transformed T2 develops a voltage V1 in response to current conducted through primary winding 112.
  • a saturable reactor L s2 include a reactor winding 122 and a saturable reactor core 124. Winding N2 is connected to a rectifier 114, again comprising a series diode 116 and a shunt diode 118, through reactor winding 122. The rectified voltage is then applied through low pass filter 120, which includes series inductor L2 and shunt capacitor C2, to provide a direct current output voltage V o2 across output terminals 126.
  • a clamping circuit 130 is connected across reactor winding 122 and includes a clamping transistor Q2 and a diode 134.
  • clamping transistor Q2 is actuated to clamp a short circuit across reactor winding 122 at a desired reset point on the B-H hysteresis characteristic of saturable core 124.
  • the control signal applied to the base electrode of transistor Q2 is obtained from a control circuit 136 and specifically from a comparator 138 is derived from an auxiliary winding N F of transformer T2.
  • the voltage appearing across winding N F includes information on the timing and voltage of the input pulse wave applied through the transformer. Winding N F typically may be the same winding used for feed-forward compensation (not shown).
  • the voltage provided by winding N F might instead be obtained from any other winding, such as windings N1 or N2 of the transformer.
  • the voltage from winding N F is applied to an integer 140 including a series resistor R and a shunt capacitor C.
  • the capacitor is shunted by a diode D1, which functions as a baseline clipper to keep the signal V3, which is applied to one input terminal of comparator 138, positive.
  • error circuit 150 develops an error signal V e from output voltage V o2 and applies it to the other input terminal of comparator 138.
  • error circuit 150 includes a differential amplifier 42.
  • a reference voltage V ref is applied to a first input terminal 43 of amplifier 42.
  • Output voltage V o2 is applied across input terminals 44 and 45, the latter of which is grounded.
  • Terminal 44 connects voltage V o2 through a series resistor R1 to a second input terminal 46 of differential amplifier 42.
  • Resistor R1 is shunter by a resistor R2 and a capacitor C3 in series, and a resistor R3 and capacitor C4 in series form a feedback circuit for amplifier 42.
  • Impedances R1, R2 and C3 and R3 and C4 are frequency shaping and compensation networks.
  • the reference voltage V ref is preferably selected to be of such magnitude that the error voltage V e wil always be of positive polarity.
  • the saturable core 124 may be formed of a wide variety of low cost, magnetically soft materials and may be formed in physical shapes which have small gaps.
  • Low remanenent core materials such as Stackpole 24B or Ferroxcube 3C8 )a ferrite material
  • Such materials provide a B-H hysteresis characteristic which is poor in squareness as illustrated by the B-H characteristic shown in Fig. 5.
  • Saturable reactor L s2 is thus much less expensive than saturable reactor L s1 of the prior art circuit of Fig. 1.
  • saturable core 124 will be at rest point e of the hysteresis loop when transistor Q1 switches on.
  • the core then travels the path e-f-g and saturates.
  • transistor Q1 turns off.
  • the voltage of secondary winding N2 then reverses during the reset period of transformer T2. This reverse voltage brings the core from point g back toward the remanent flux density B r along the upper branch of the loop.
  • transistor Q2 switches on to clamp a short circuit across reactor winding 122.
  • the current in winding 122 now circulates through transistor Q2 and core 124 stays at reset point e waiting for the next pulse.
  • the magnetic flux density falls an amount designated as B' in Fig. 5.
  • the available range of adjustment is designated by B A , the distance between the saturation point g and the remanent flux density B R . Since reset point e is in the same (upper right) quadrant of the hysteresist characteristic as the saturation point g , the core operates entirely within a single quadrant making it unnecessary to use a forcing current of reverse polarity to reset the core as is required in the prior art circuit of Fig. 1.
  • control circuit 136 will be understood from the voltage curves of Fig. 6.
  • the curve V1 represents the voltage V1 from secondary winding N2 as indicated on Fig. 3. From this curve, it is seen that V1 has a positive magnitude V F during the forward conduction period of winding N2 and a negative magnitude V R during the recovery period of transformer T2.
  • the voltage magnitudes V F and V R are usually, but not necessarily, equal.
  • V1 is held at magnitude V F for a time D1t cyc , where D1 represents the duty cycle of the main output voltage V o1 and t cyc represents the period of the switching inverter-regulator - ­that is, the time for one switch cycle of transistor Q1.
  • t cyc is equal to the inverse of the switching frequency f sw
  • Curve V2 represents the voltage V2 appearing at the output side of reactor winding 122 and is thus also the input voltage supplied to rectifier 116.
  • Voltage V2 has a magnitude V' F for a time D2t cyc , where D2 at a value of the control loop is to maintain duty cycle D2 at a value which will keep output voltage D o2 at its desired value.
  • the main control is effected by the delay td2, the delay between the onset of the positive pulses of voltage waves V1 and V2. This delay results from the operation of saturable core 124. At the time of the onset of positive pulse V F of input voltage wave V1, the core is at its reset point e and is not saturated.
  • Winding 122 therefore presents a high impedance to the applied voltage blocking the start of the corresponding positive pulse V' F of voltage V2 on the output side of the reactor winding.
  • the core reaches point g on its hysteresis loop, the core saturates and the impedance of reactor winding 122 becomes low permitting the reactor winding to apply the pulse V' F of voltage wave V2 to the output side of the reactor.
  • Delay td2 is a direct function of the clamp delay td1 as shown by the relationship:
  • control circuit 136 The desired relationship between delay td1 and V e is ob­tained by control circuit 136.
  • Resistor R and capacitor C form integrater 140 which provides voltage V3. If this circuit is treated as an ideal integrator, the slopes of the curve for V3 will be as seen in Fig. 6.
  • Diode D1 acts as a baseline clipper to keep the triangular wave signal positive.
  • the generated triangular wave is compared in comparator 138 with error voltage V e . Whenever the tri­angular wave voltage V3 is less than the error voltage V e , comparator 138 provides positive output signal on output lead 139. This output signal is applied to the base electrode of clamping transistor Q2 causing transistor Q2 to become conductive.
  • diode 134 is poled to block conduction through clamping circuit 130.
  • diode 134 no longer blocks conduction through circuit 130.
  • transistor Q2 becomes conductive and clamping curcuit 130 applies a short circuit clamp across reactor winding 122.
  • the short circuit current circulates in the loop formed by inductor winding 122, transistor Q2 and diode 134; and core 124 is held at reset point e .
  • the delay td1 is governed by the equation: where N2 and N F represent the number of turns of windings N2 and N F , respectively, and ⁇ is the constant of integrator 40, being equal to the product of the resistance of resistor R and the capacitance of capacitor C. Substituting the expression for td1 given in equation (3) in equation (2) and simplifying, we have:
  • the rising slope of the triangular wave of voltage V3 is defined by the expression and the declining slope by the expression
  • integrator 40 is treated as an ideal integrator. Some error is introduced by the approximate nature of the assumed integrator operation. This error may be held to an acceptable value by keeping ⁇ equal to or greater than t cyc .
  • Fig. 7 incorporates two modifications of the circuit of Fig. 3.
  • the clamping circuit is no longer connected directly across the reactor winding, but is, instead, connected across a secondary winding inductively coupled to the reactor winding.
  • the adjustment range of the circuit is increased by providing biasing means to shift the hysteresis characteristic of the core to the right.
  • a self-excited, inverter 200 As seen in Fig. 7, a self-excited, inverter 200, as shown, for example, in the aforementioned Hiramatsu et al article, generates a square wave to drive primary winding 202 of a transformer T3. It is to be understood, however, that a switching inverter as shown in the prior art circuit of Fig. 1 or the embodiment of Fig. 3 could be used to drive the transformer.
  • a square wave voltage is induced in secondary winding 204 of transformer T3.
  • a saturable reactor L s3 which is used to regulate the output voltage V o3 , includes a reactor winding 222, a reactor core 224, a secondary winding 226 and a bias winding 228.
  • Reactor winding 222 connects secondary winding 204 to a rectifier 214 and a low pass filter 220.
  • Rectifier 214 includes series and shunt diodes 216 and 218, and filter 220 includes series inductor L3 and shunt capacitor C5.
  • Output voltage V o3 appears across output terminals 221 on the output side of filter 220.
  • An error circuit 240 which may correspond to the circuit of Fig. 4, develops error voltage V e and applies it to one input of comparator 238.
  • the other input of comparator 238 is received from an integrator and clipper circuit 250, identical to the integrator 140 and diode clipper D1 of the embodiment of Fig. 3.
  • An auxiliary winding 206 on transformer T3 provides a sample of the input voltage from transformer T3 to integrator 250, but this sample could also be taken from across another winding, such as winding 204, of the transformer.
  • Comparator 238 provides an output signal on lead 239 whenever the magnitude of the triangular wave from integrator and clipper 250 is less than the error voltage V e . This output signal is applied on lead 239 to the base electrode of clamping transistor Q3 of clamping circuit 230 causing the transistor to become conductive. During the positive pulse in winding 204, diode 234 blocks the clamping circuit from applying a short circuit across a reactor secondary winding 226 inductively coupled to reactor winding 222.
  • Diode 234 is poled to permit conduction through transistor Q3 on the reverse wave appearing in winding 204; a short circuit is then clamped across secondary winding 226, effectively clamping a short circuit across reactor winding 222 as a current induced from reactor winding 222 circulates in the loop including winding 226, diode 234 and transistor Q3.
  • Core 224 of saturable reactor L s3 may be identical to the core 124 of the embodiment of Fig. 3. As explained above, the core may be made of magnetically soft material and be formed with small gaps. Such cores are relatively inexpensive and have hysteresis characteristics which are poor in squareness.
  • the effective B-H loop of core 224 is shifted to the right to increase the available flux swing. This is accomplished through the use of bias winding 228 connected across output voltage V o3 through inductor L4 and resistor R4. Because the reset point e may be adjusted as far as remanent flux density B' R over an available range of adjustment B' A which is much larger than the available range of adjustment B A for the embodiment of Fig. 3 (see Fig. 5), the use of bias winding 228 permits a wider range of voltage control.
  • the circuit of Fig. 7 otherwise operates in the same manner as the circuit as Fig. 3.
  • the output voltage V o3 is regulated by adjusting the position of reset point e of the hysteresis characteristic of core 224 in response to the magnitude of error voltage V e .
  • Core 224 is reset during the reverse wave when clamping transistor Q3 becomes conductive. The core is then clamped at its reset point e .
  • the reset point e determines the duty cycle of voltage V2 and thus the magnitude of output voltage V o3 .
EP19860110982 1985-08-05 1986-08-08 Sources d'alimentation comportant un amplificateur magnétique de régulation de la tension Expired EP0255844B1 (fr)

Priority Applications (3)

Application Number Priority Date Filing Date Title
US06/762,648 US4642743A (en) 1985-08-05 1985-08-05 Power supplies with magnetic amplifier voltage regulation
DE8686110982T DE3671553D1 (de) 1986-08-08 1986-08-08 Energieversorgungen mit magnetverstaerker zur spannungsregelung.
EP19860110982 EP0255844B1 (fr) 1986-08-08 1986-08-08 Sources d'alimentation comportant un amplificateur magnétique de régulation de la tension

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
EP19860110982 EP0255844B1 (fr) 1986-08-08 1986-08-08 Sources d'alimentation comportant un amplificateur magnétique de régulation de la tension

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EP0255844A1 true EP0255844A1 (fr) 1988-02-17
EP0255844B1 EP0255844B1 (fr) 1990-05-23

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Cited By (6)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP0357411A2 (fr) * 1988-08-31 1990-03-07 Zytec Corporation Régulateur à inductance commandée pour la commutation d'alimentation de puissance
EP0382307A2 (fr) * 1989-02-09 1990-08-16 Philips Patentverwaltung GmbH Dispositif d'alimentation de puissance à commutation
GB2233479A (en) * 1989-06-30 1991-01-09 Digital Equipment Int Regulated DC-DC power supply
DE3943027A1 (de) * 1989-12-27 1991-07-04 Ant Nachrichtentech Mit einem gleichrichter beschaltete saettigungssteuerbare drossel
EP0465903A1 (fr) * 1990-07-13 1992-01-15 André Bonnet Procédé magnétique de controle du transfert d'énergie dans un convertisseur statique
EP0745922A1 (fr) * 1995-05-30 1996-12-04 Motorola, Inc. Dispositif et procédé pour la régulation adaptative de puissance dans un circuit intégré

Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
FR2443763A1 (fr) * 1978-12-08 1980-07-04 Philips Nv Alimentation a decoupage comportant plusieurs sorties
EP0083216A2 (fr) * 1981-12-25 1983-07-06 Fanuc Ltd. Appareil pour stabiliser une source d'alimentation
DE3209975A1 (de) * 1982-03-18 1983-09-29 Nixdorf Computer Ag, 4790 Paderborn Schaltungsanordnung zur steuerung der hoehe einer abzugebenden pulsierenden spannung, insbesondere in einem gleichspannungswandler
EP0123098A2 (fr) * 1983-03-28 1984-10-31 Intronics, Inc. Réglage d'un dispositif d'alimentation à commutation
EP0150797A2 (fr) * 1984-01-23 1985-08-07 Hitachi, Ltd. Alimentation de puissance à découpage avec sortie à commande magnétique

Patent Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
FR2443763A1 (fr) * 1978-12-08 1980-07-04 Philips Nv Alimentation a decoupage comportant plusieurs sorties
EP0083216A2 (fr) * 1981-12-25 1983-07-06 Fanuc Ltd. Appareil pour stabiliser une source d'alimentation
DE3209975A1 (de) * 1982-03-18 1983-09-29 Nixdorf Computer Ag, 4790 Paderborn Schaltungsanordnung zur steuerung der hoehe einer abzugebenden pulsierenden spannung, insbesondere in einem gleichspannungswandler
EP0123098A2 (fr) * 1983-03-28 1984-10-31 Intronics, Inc. Réglage d'un dispositif d'alimentation à commutation
EP0150797A2 (fr) * 1984-01-23 1985-08-07 Hitachi, Ltd. Alimentation de puissance à découpage avec sortie à commande magnétique

Cited By (10)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP0357411A2 (fr) * 1988-08-31 1990-03-07 Zytec Corporation Régulateur à inductance commandée pour la commutation d'alimentation de puissance
EP0357411A3 (fr) * 1988-08-31 1990-10-31 Zytec Corporation Régulateur à inductance commandée pour la commutation d'alimentation de puissance
EP0382307A2 (fr) * 1989-02-09 1990-08-16 Philips Patentverwaltung GmbH Dispositif d'alimentation de puissance à commutation
EP0382307A3 (fr) * 1989-02-09 1990-11-14 Philips Patentverwaltung GmbH Dispositif d'alimentation de puissance à commutation
GB2233479A (en) * 1989-06-30 1991-01-09 Digital Equipment Int Regulated DC-DC power supply
GB2233479B (en) * 1989-06-30 1993-09-08 Digital Equipment Int Power supply
DE3943027A1 (de) * 1989-12-27 1991-07-04 Ant Nachrichtentech Mit einem gleichrichter beschaltete saettigungssteuerbare drossel
EP0465903A1 (fr) * 1990-07-13 1992-01-15 André Bonnet Procédé magnétique de controle du transfert d'énergie dans un convertisseur statique
US5184289A (en) * 1990-07-13 1993-02-02 Andre Bonnet Magnetic control process for transfering energy in a static converter
EP0745922A1 (fr) * 1995-05-30 1996-12-04 Motorola, Inc. Dispositif et procédé pour la régulation adaptative de puissance dans un circuit intégré

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DE3671553D1 (de) 1990-06-28
EP0255844B1 (fr) 1990-05-23

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