DD282788A5 - METHOD AND CIRCUIT FOR A POWER AMPLIFIER WITH LOW PHASE ROTATION, ESPECIALLY FOR A VARIABLE INDUCTIVE LOAD - Google Patents

Abstract

The invention relates to a method and a circuit for a Leistungsverstaerker with low phase rotation and high stability of the output voltage when switching on an inductive load. It is applicable in the field of NF power amplifier technology and in the measuring technique. The invention solves the problem of providing a method for a power amplifier, by which the load current generates only a very small voltage drop between the input and the output terminal. The power amplifier is intended as a current driver at the output of a bipolar operational amplifier allow a higher capacitive load as a complementary emitter follower. This object is achieved according to the invention by providing the load current of an amplifier with an NPN transistor through two controllable current sources, one device depending on whether the collector current of the NPN transistor exceeds or falls below a reference value, one or the other Current source and that the ratio between the maximum value and the minimum value of the collector current is kept small by a sufficient amount of the minimum value. The invention further solves the problem of providing a circuit suitable for carrying out the method with little effort. This object is achieved according to the invention by a circuit according to FIG. Fig. 4 {current driver; Load current; low voltage drop; Reference value; maximum collector current; minimum collector current; Meszeinrichtung; Control device; controllable power source}

Description

For this 3 pages drawings

Field of application of the invention

The invention is generally applicable in the field of NP power amplifier technology, and especially in metrology.

The inventive method allows the construction of a power amplifier, which is characterized by low phase rotation, especially in capacitive load, Jnd by a very small voltage drop between the input and output terminal with inductive load.

The low phase rotation is particularly advantageous when used as a current driver with capacitive load at the output of an operational amplifier.

The very small voltage drop with inductive load allows the generation of a high-constant AC voltage for a variable number of inductive Meßgrößenaufnehmer.

Characteristic of the known state of the art

The complementary emitter follower has gained great importance as a power amplifier due to its relatively high efficiency. A serious shortcoming is the insufficient linearity of its transfer characteristic. The transfer characteristic of the complementary emitter follower is linearized by the AB operation (Tietze / Schenk, Halbleiterschaltungstechnik, 5th edition, Springer-Verlag Berlin, Heidelberg, New York 1980, page 349.) In a complementary emitter follower in the AB mode, the quiescent current must be stabilized be used to suppress the effect of thermal Mitkopplung.

A very simple and reliable measure for this is in the current feedback lurch emitter resistors. At these emitter resistors, however, the load current causes an additional voltage drop, which can not be compensated by negative feedback (using a preamplifier), if it is an inductive load.

Without emitter resistance, this error reduces to the amount caused by the voltage drop across the differential resistance of the base-emitter diode of the respective conducting transistor.

In DE-OS 3727248 a power amplifier is shown, which reacts to inductive load virtually error-free, although it also contains emitter resistors to stabilize the quiescent current. Another problem, however, is the phase rotation of a power amplifier, especially at higher capacitive load, as occurs when the consumer has a longer connection cable. When the power amplifier is included in the negative feedback loop of an operational amplifier, the additional phase rotation of the power amplifier reduces the phase-locking efficiency of the operational amplifier circuit for a given compensation. The additional phase rotation can be compensated with the aid of a fast feedback loop, provided that the capacitive load is reasonably constant (Dost.11, operational amplifier, Verlag Technik Berlin 1986,

P. 340). However, this requirement is not met with a variable number of individual consumers with their own cable. Since the maximum capacitive load must be compensated, overcompensation always occurs when the load is reduced

At the typical transit frequency of a 3 MHz bipolar operational amplifier, the phase rotation of a capacitively loaded complementary emitter follower is not negligible, due to the unfavorable high frequency characteristics of a pnp transistor. This also applies to the power amplifier shown in DE-OS 3727248. In terms of phase rotation, the simple emitter follower with an npn transistor is superior to the complementary emitter follower. The simple emitter follower is characterized by a low efficiency; when using a constant current source as a working resistance, it is still less than 25%.

Solutions are known (DE OS 3416850, DE-PS 3120689, US-PS 4573021, US-PS 4509020), in which the input is connected to the output of the power amplifier as in the simple emitter follower through the base-emitter diode of a transistor, and with which a high efficiency is achieved. The load current causes but at a certain polarity still a Spannungsat case on the differential resistance of the base-emitter diode between the input and the output

of the amplifier, so that these solutions are not suitable for generating a highly constant output voltage for a variable inductive load. In addition, the non-linear current-voltage characteristic of the base-emitter diode affects the linearity of the transfer characteristic.

Object of the invention

The aim of the invention is a power amplifier with low phase rotation and high stability of the output voltage when switching on an inductive load.

Explanation of the essence of the invention

The invention solves the problem of providing a method for a power amplifier, by which the load current generates only a very small voltage drop between the input terminal and the output terminal. The power amplifier should be used as a current driver at the output of a bipolar operational amplifier with a typical transit frequency of 3MHz and allow under this condition, a higher capacitive load as a complementary Fmitterfolger with comparable performance dimensioning.

This means that the method must be able to be implemented in a circuit which firstly has a lower dynamic output resistance than the complementary emitter follower in the range of the mentioned transit frequency (3 MHz) and secondly does not preclude the higher capacitive load fundamentally by dynamic instability, in principle insofar as that too can be achieved by a frequency response correction under the condition of a sufficiently high power bandwidth of the amplifier (Slew rate at least 1 V / ps) no stability.

This object is achieved in that the load current is provided by two controllable current sources, in a known manner, the first current source with the collector cmitter current path an NPN transistor in collector circuit is connected in series, while according to the invention, the second current source to the collector emitter -Strompfad dos npn transistor is connected in parallel, that with the aid of a device, the first current source is turned on in a known manner, when the collector current of the NPN transistor falls below a reference value, while according to the invention, the second current source is turned on when the collector current exceeds this reference value, and the ratio between the maximum value and the minimum value of the collector current is kept small by a sufficient height of the minimum value.

The invention further solves the problem of providing a circuit suitable for carrying out the method with little effort. This object is achieved in that in a circuit of the type mentioned, the device for evaluating the collector current of the first transistor is a resistor that is connected to the second resistor, a fourth diode in series, that at the collector of the first transistor, the base of a fourth Transistor is connected by pnp line type, whose collector is connected to the output of the power amplifier, that further between the positive pole and the negative pole of the operating voltage source, a series circuit of a third resistor, a first, second and third diode and a fourth resistor in the is arranged, wherein all three diodes are poled in the flow direction, and that connected at the connection point between the first and the second diode, the emitter of the fourth transistor and at the connection point between the third diode and the fourth resistor, the base of the third transistor is.

AusfChrungsbeispiel

FIG. 1 shows the block diagram for the method according to claim 1.

FIG. 2 shows the currents ioi and Iq ₂ as a function of the collector current ic-

In practice, the lines \ q \ = f (i _c ) and Iq ₂ = f (ic) need not have exactly the same slope (apart from the sign).

FIG. 3 shows the course of the currents of FIG. 1 with a sinusoidal full scale control of the power amplifier. The inventive method works with two control devices that operate with the same reference variable, the reference value Ic.

If the collector current ic falls below the reference value l _c , the first control device is activated. It counteracts the further reduction of the collector current i _c by increasing the current of the first current source Q1. When the collector current i _c exceeds the reference value Ic, the second regulator is activated.

It counteracts the further enlargement of the collector current ic by increasing the current of the second current source Q 2.

Both control devices work with the measuring device 3 for the collector current i _c (Figure 1).

The required for the function of a control device for forming the difference between the collector current ic and the reference value l _c is part of the device 4. It can be designed for both control devices together or for each control device individually. It is important to note that the two straight lines Iqi = f (i'c) andio ₂ = f (ic) in Figure 2 coincide at ίο = 0 or intersect slightly above zero at a value. The stronger control of both power sources at the same time is reliably excluded.

The device 4 must generate a control signal with the required reference potential for each current source.

The gain required for the function of the control devices can be achieved in different ways. The device 4 may have an amplifier for both control devices in common, for. B. a differential amplifier with two outputs, or two individual amplifier included. The required gain can also be achieved by a sufficiently large

Ratio between the current of a power source and its control signal can be achieved. According to FIG. 1, the node equation applies

i _Q2

- ιοί - i _L - 0

The consequence of the nodal point calibration and the relationship i _E »i _c is that iT.rner has at least one control device for a change in the load current i |. reacts with the change of the current of its associated power source. As a result, the load current i _{L is} largely provided by the two current sources Q1 and Q2, and the collector current ic changes with the laser current only in the ratio \ _c : \ i = 1: v _o n ,, ic: ii. = 1: ν _ο κ ₂ or, if both control devices in the vicinity of iq = 0 are active, in the ratio IcMl = 1: (v _o in + V ₀ H ₂ ) (V ₀ H ₁ : open gain of the first control device with Q1, v _o n ₂ : open amplification of the second control device with Q2).

According to the invention, the ratio between the maximum value l _Cm , _x and the minimum value l _Cm in the collector current i _c (Figure 2) is kept small by a sufficiently large reference value Ic. From a small ratio results in a small change in the base-emitter voltage of the NPN transistor T and from a high constancy of the output voltage when switching on an inductive load.

Another advantage over an amplifier of the type mentioned is that the input resistance of the power amplifier is no longer dependent on the sign of the load current.

For It _- = 0, the current I _{R of} the power amplifier (see FIG. 3) becomes the quiescent current I _R of the current i _{Q1 of} the first current source Q1. The collisor current ic assumes the quiescent value I _C r.

As FIG. 3 shows, the first current source Q1 is charged somewhat stronger than the second current source Q 2, provided the load current κ is a pure alternating current. The quantitatively accurate representation of the curves of FIG. 3 is based on the assumption of the following relationships:

lc = 0.2 _{l Lmlx}

i _Q , = 10 χ (Ic - ic) I ic ^ Ic

i _Q2 = 10x (ic-Ic) | ic a Ic

FIG. 4 shows the circuit for a low-phase-rotation power amplifier, in particular for a variable inductive load, according to claim 2.

This power amplifier operates according to the method of claim 1.

The collector current i _{c of} the first transistor T1 generates a voltage drop across a measuring resistor R _M , with which the evaluation of the collector current takes place.

The resistor R4 generates a nearly constant current Ir ₄ at a constant operating voltage. When the fourth transistor is turned off, a voltage drop corresponding to the product Iq ₄ x R3 results at the third resistor R3.

The resistance R3 is very small, so that the voltage drop at R3 is very small. The voltage difference between the positive pole of the operating voltage source and the base of the third transistor is equal to the sum I _R4 χ R3 + 3 χ U _F (U _f :

Forward voltage of a diode).

The transition of the third transistor T3 from the blocked state to the conductive state and vice versa takes place practically at a base-emitter voltage Ube in the amount of the forward voltage Up of a diode. If the third diode D3 technologically corresponds exactly to the base-emitter diode of the third transistor T3, and as the switching point of the third transistor, the coincidence of its emitter current is defined with the current Ir ₄ , the third transistor T3 opens if and only if the collector current is the value ic = (Ir ₄ χ R3 + 2 χ U _f : Rm falls below.

It can be seen from Fig. 4 that the fourth transistor T4 opens when the collector current ic exceeds approximately the same value. It follows that the value Ic = (Ir ₄ χ R3 + 2 χ Uf: Rm corresponds to the reference value l _{c of} Figure 1, which shows the block diagram for the circuit of Figure 4 underlying method corresponds.

When the collector current i _c is less than the reference value Ic, the differential current I _c flows - ic as an emitter current \ ^ ₃ remains in the emitter of the third transistor T3, and the voltage drop across the measuring resistor R _M approximately constant (it reduces with increasing falls below the reference value l _c something, since with the emitter current of the third transistor T3, the base Fmitter voltage of T3 increases slightly). Advantageously, while the base-emitter diode of the fourth transistor T4 remains poled in the forward direction. The emitter current i _ET3 6. iugt on the second resistor R 2, the voltage drop (l _c - ic) x R2.Dievierte diode D4 is intended to compensate for the base-emitter voltage of the second transistor T2. As a result, the second transistor T2 opens precisely when the collector current i _{c falls} below the reference value l _c .

In addition, the compensation with the fourth diode allows the dimensioning R2 <R _M , and for the resistor R1, a very small value can be set, without requiring a very large ratio R 2: R 1 is required. The second transistor T2 forms, together with the first resistor R1, the first current source Q1 according to FIG. 1. A change in the collector current causes, under the condition i _c <Ic, a change in the current i _Q1 that is greater by R2: R1.

When the collector current ic exceeds the reference value I _c , the fourth transistor T4 is opened.

Advantageously, while the base-emitter diode of the third transistor T3 remains poled in the forward direction, because the potential at the base of the third transistor T3 changes by the same amount as the potential at the emitter of the fourth transistor T4. The fourth transistor, together with the third resistor R3, forms the second current source Q2 according to FIG.

A change in the collector current i _c causes, under the condition ic> Ic, a change of the current i _Q2 that is greater by the same amount Rm: R3.

So that the transmission ratio between the current Iq ₂ and the collector current ic under the condition ic> Ic matches the transmission ratio between the current io _t and the collector current ic under the condition ic> Ic, the dimensioning rule R _M : R3 = R2.R1 must be observed.

When the collector current ic exceeds the reference value Ic, the collector current of the third transistor T3 and thus the current through the fourth diode D4 goes to zero. This can be prevented by connecting the fourth resistor R4 not to the negative pole 2 of the operating voltage source but to the base of the second transistor T2, so that by mirroring the current I _R4 with the factor (R2: R1) a minimum value for the current i _{Q2 is} set.

When the fourth resistor R4 is connected to the negative pole of the operating voltage source, it can be replaced by a constant current source. As a result, in particular the dependence of the current I _{R4 is} eliminated from the operating voltage. When the fourth resistor R4 is connected to the dsr base of the second transistor T2, it may be replaced by a constant current dipole requiring no reference potential or by two constant current sources, of which d:

one connects the base of the third transistor T3 to the negative pole and the other connects the base of the second transistor T2 to the positive pole of the operating voltage source.

The reference value l _c was based on the switching point Λ of the third transistor T3. As a switching point of the third transistor, the match of its Emitturstroms i _{ET3 was defined} with the current I _R4 . Thus, at i _c = Ic, the current i _Q1 results from i _Q , = (R2: R1) x Ir ₄ . If at ic = l _c of the current i _Q2 as great as the current i _Q, seinsoll must the base-emitter voltage of the fourth transistor T4 in i _O 2 = (R2: R1) χ I _R4 to the base-emitter Voltage of the third transistor T3 at i _eT3 = I _R4 match.

This condition can be well maintained when integrating the circuit, when the area of the base-emitter junction of the fourth transistor is set in the ratio R2: R1 to the surface of the base-emitter junction of the third transistor.

At the same time, the higher thermal load of the fourth transistor, which is located in the load circuit, can be taken into account. In a discrete realization of the circuit, the match iai = Ic at ic = Ic can hardly be achieved.

The thermal load of the fourth transistor T4 is a particular disadvantage of the circuit according to FIG. 4 because it leads to a shift in the characteristic | q2 = Wc) of FIG. 2 via the dependence of the transistor's emitter-emitter voltage on the crystal temperature.

In addition, a PNP power transistor has particularly unfavorable high frequency characteristics.

FIG. 5 shows a circuit with a fifth transistor of the npn type according to claim 6, which avoids the mentioned disadvantages of the circuit of FIG.

The fifth transistor T5 relieves the fourth transistor T4, and the third resistor R3 is now to be dimensioned high-impedance. A change in the collector current ic causes a change in the current Iq _{2 which is} greater by (Rm: R3) x (R5: R6 ₎ under the condition ic> Ic.

The following dimensioning rule is particularly suitable:

R2 = R3 = R5 = R _M.

It is of course expedient to use for T3 and T4 two characteristic-like transistors, preferably integrated on a common chip.

Likewise, it is jwickmäßig, for the third, fourth and fifth diode each use a transistor, dr, r is integrated with the transistor, whose base-emitter voltage is werdan, is integrated on a common chip.

Claims (6)

1. Method for a power amplifier m ^! t low phase rotation, in particular for a variable inductive load, with an npn transistor whose collector is connected via a current path to the positive pole of an operating voltage source whose base is connected to the input and whose emitter is connected to the output of the power amplifier, with at least one component, which makes the collector current of the NPN transistor of a direct evaluation or via the base-emitter voltage of the NPN transistor accessible to an indirect evaluation, with a controllable current source which connects the emitter of the NPN transistor to the negative pole of the operating voltage source the current source can be realized by a further npn transistor and an emitter resistor, with a differential amplifier or an equivalent arrangement for forming and amplifying the difference between the collector current of the npn transistor or a signal derived therefrom and a reference value, wherein a pnp Transi can be used in base circuit, and with a connection between the output of the differential amplifier and the control input of the power source, so that the current source is turned on when the collector current falls below the reference value, characterized in that the above-mentioned differential amplifier having only one output , is replaced by a device (4) which has two outputs and on the first output the first, initially mentioned current source (Q 1) aufsteuert in a known manner, when the collector current (i _c ) of the npn transistor (T) has a reference value (l _c ), while via the second output a second, controllable current source (Q2), which is connected between the positive pole (1) of the operating voltage source and the output (A) of the power amplifier, is turned on, when the collector current (ic) exceeds the same reference value (Ic), wherein between the current O ₀₂ ) of the second current source and the amount d he Reference value exceeding the collector current ' _c ) the same or approximately the same, sufficiently large gear ratio as between dt_. .i current (i _Q1) of the first current source and the amount derReferenzwertunterschreitung of the collector current, and in that the ratio between the maximum value (Lcmax) ^lJ nd the minimum value DCmin) of the collector current (Ic) is kept small. Figure 1, Figure2 2. A circuit for a power amplifier with a low phase rotation, in particular for a variable inductive load, with a first transistor of the npn type conductivity, whose collector via a current path to the positive pole of an operating voltage source whose base with the input and its emitter with the output of Leistunpsverstärkers is connected to a second transistor of the npn type conductivity, whose collector is connected to the output of the power amplifier and its emitter via a first resistor to the negative pole of the operating voltage source, to a third transistor of the pnp-type conductivity, its emitter to the collector the first transistor is connected, and whose collector is connected to the base of the second transistor and via a second resistor to the negative pole of the operating voltage source, characterized in that the component for evaluating the collector current (i _c ) is a resistor (R _M ) that with the zw Resistor (R2) a fourth diode (D4) is connected in series, that the collector of the first transistor (T 1) is connected to the base of a fourth transistor (T4) of the pnp-type conductivity, whose collector to the output (A) of the Power amplifier is further connected that further between the positive pole (1) and the negative pole (2) of the operating voltage source, a series circuit of a third resistor (R3), a first, second and third diode (D 1, D2, D3) and a fourth Resistor (R4) is arranged in said order, wherein all three diodes (D 1, D2, D3) are poled in the flux direction, and that at the connection point between the first diode (D 1) and the second diode (D 2), the emitter the fourth 1 ransistors (T4) and at the connection point between the third diode (D3) and the fourth resistor (R4), the base of the third transistor (T3) is connected. Figur4 3. A circuit for a power amplifier according to claim 2, characterized in that the fourth resistor (R4) is not connected to the negative pole (2) of the operating voltage source, but with the base of the second transistor (T2). 4. A circuit for a power amplifier according to claim 2, characterized in that the fourth resistor (R4) is replaced by a constant current source. 5. A circuit for a power amplifier according to claim 3, characterized in that the fourth resistor (R4) is replaced by a Konstantstromzweipol, or that instead of the fourth resistor, two constant current sources are used, one of which is the base of the third transistor (T3) the negative pole (2) and the other connects the base of the second transistor (T2) to the positive pole (1) of the operating voltage source. 6. A circuit for a power amplifier according to claim 2,3,4 or 5, characterized in that the collector of the fourth transistor (T4) with the base of a fifth transistor (T5) of the npn type conductivity and via a fifth diode (D5) and a fifth resistor (R 5) is connected to the output (A) of the power amplifier, that the emitter of the fifth transistor (T5) is connected to the output (A) via a sixth resistor (R 6), and that the collector of the fifth Transistor (T5) is connected to the positive pole (1) of the operating voltage source.