CN1783692A - Speed controller of synchronous motor - Google Patents

Abstract

The invention provides a speed control device of a synchronous motor. Obtain the electrical angular frequency instruction ω1 ^* from the rotation instruction ωr ^* , obtain the correction amount Δω1 from the difference between Iq ^* and Iqc, add ω1 ^* and Δω1 to obtain ω1c, use the integrator (8) to integrate ω1c to obtain the AC phase θdc, The current 10 detected by the current detector (6) is sampled with the current sampler (91), and the alternating current is reproduced with the current reproducer (32). Current carries out coordinate transformation, seeks Iqc, seeks Iq ^* from Iqc with Iq ^* generator (10), according to Id ^* , Iq ^* , ω1 ^* , seeks applied voltage instruction Vdc ^* , Vqc ^* with voltage instruction arithmetic unit (12), Use the dq inverse converter (13) to find the three-phase AC voltage commands vu ^* ~vw ^* from these applied voltages, use the PWM generator (14) to generate PWM signals according to the three-phase AC voltage commands, and control the inverter according to the PWM signals ( 3).

Description

Synchronous motor speed control device

This application is a divisional application of the application dated July 3, 2003, the application number is 03146514.5, and the invention name is a speed control device for a synchronous motor.

technical field

The present invention relates to a speed control device for a synchronous motor, in particular to a speed control device for a synchronous motor suitable for controlling the speed of the synchronous motor without using a magnetic pole position sensor for detecting the magnetic pole position of the synchronous motor and a current sensor for detecting the current of the synchronous motor .

Background technique

As a control method for controlling the speed of a synchronous motor composed of a magnet motor, various methods have been proposed, such as a method not using a magnetic pole position sensor, a method not using a current sensor, and the like.

In the conventional control method, the control method that does not use the magnetic pole position sensor is to replace the magnetic pole position sensor and install the magnetic pole position estimator. The basic structure is composed of a speed controller and a current controller. The structure is the same, it is based on vector control.

The basic principle of magnetic pole position estimation is to perform estimation calculation of magnetic pole position based on electric constant of synchronous motor, motor voltage and motor current. As a device using induced voltage, for example, the device described in JP-A-2001-251889 is known. .

The principle of inferring the magnetic pole position is to infer the axis error Δθ between the rotating coordinate axis (d-q axis) based on the magnetic pole position of the synchronous motor and the rotating coordinate axis (dc-qc axis) assumed in the control, by correcting the synchronous motor The frequency command makes the axis error obtained by this operation 0, realizing position sensorless vector control.

In the case of position sensorless vector control, the size and phase of the driving current can be ideally controlled according to the load conditions, and the control of high torque and high performance synchronous motors can be realized.

On the other hand, as a control method that does not use a current sensor, a so-called current reproduction method has been proposed that detects the DC current of the inverter that drives the motor, and reproduces the AC current of the motor based on its instantaneous value and the gate pulse signal of the inverter. In this current reproduction method, for example, as described in JP-A-2-197295, a pulse signal for driving the inverter is used to sample and hold the motor current instantaneously expressed in the DC current of the inverter to indirectly detect the motor current.

In the conventional control method based on position sensorless vector control without magnetic pole position sensor, multiple speed controllers, current controllers, and magnetic pole position estimators must be installed to form a feedback loop controller, which makes the control structure complex. In particular, if the motor is to be rotated and driven at high speed, it is difficult to stabilize the control system as a whole. In order to make the overall stability of the control system, it is necessary to shorten the control operation cycle and set the control gain to be very high, but it is difficult to realize without using high-performance operation processors such as DSP (Digital Signal Processor).

In addition, using the current reproduction method has the following problems. That is, in the current reproduction method, since the motor current is reproduced based on the DC current of the inverter and the gate pulse signal of the inverter, when the command voltage is low and the pulse width of the gate pulse is extremely short at the time of starting, etc., It is difficult to capture the current component of the motor. In particular, when the speed of the motor is increased, the higher the average switching frequency (carrier frequency) of the inverter is set, the shorter the pulse width of the gate pulse is, and it becomes difficult to reproduce the current. As a countermeasure against this, the carrier frequency of the inverter is reduced only when the motor is started. Causes of harsh electromagnetic noise.

In this way, when the "magnetic pole position sensorless control method" and the "current reproduction method" are combined, when the motor is rotated at a high speed, for example, at a frequency of 400 Hz or more, the calculation cycle must be accelerated according to the carrier frequency. There is a limitation in the gate pulse width, so it is difficult to speed up the operation cycle alone. Therefore, it is difficult to realize a high-speed and high-performance synchronous motor control device without both the magnetic pole position sensor and the current sensing device.

Contents of the invention

An object of the present invention is to provide a speed control device for a synchronous motor that can rotate the motor at high speed in a state where the control system is stable without using a magnetic pole position sensor or a current sensor.

In order to solve the above-mentioned problems, the speed control device of the synchronous motor of the present invention includes:

Response to the pulse width control signal to change the output voltage of the DC power supply into a three-phase AC voltage of variable voltage and variable frequency, and apply it to the inverter on the synchronous motor;

an inverter current detector for detecting an inverter current supplied from the above-mentioned DC power source to the above-mentioned inverter;

a revolution command generator generating a revolution command with respect to the aforementioned synchronous motor; and

Generate the above-mentioned pulse width control signal according to the above-mentioned number of rotation instructions, and output it to the controller of the above-mentioned inverter,

Among them, the above-mentioned controller is composed of the following devices:

a sampling device for sequentially sampling the inverter current detected by the inverter current detector;

A current reproducing means for reproducing the alternating current flowing through the above-mentioned synchronous motor based on the sampled current value sampled by the above-mentioned sampling device;

dq coordinate transformation means for transforming the coordinates of the alternating current reproduced by the above-mentioned current reproducing means into the dc axis assuming the magnetic pole axis inside the above-mentioned synchronous motor and the current on the qc axis orthogonal to the above-mentioned dc axis;

According to the current component on the qc axis obtained by the coordinate transformation of the above-mentioned dq coordinate transformation device, a torque current command generation device for generating a torque current command about the above-mentioned synchronous motor;

a d-axis current command generating device for generating a d-axis current command;

An applied voltage command calculation device for calculating respective applied voltage commands on the dc axis and the qc axis based on the rotation speed command, the torque current command, and the d-axis current command;

A phase calculation device for calculating an AC phase related to the drive frequency of the above-mentioned synchronous motor according to the above-mentioned rotation number instruction;

According to the AC phase calculated by the above-mentioned phase calculation device, the coordinates of each applied voltage command are transformed into a dq inverse conversion device for a three-phase AC voltage command;

A pulse width control signal generating device for generating a pulse width control signal according to the above-mentioned three-phase AC voltage command;

a state quantity computing means for calculating a state quantity corresponding to an error angle between the dc-qc axis and the d-q axis which is the actual magnetic pole axis of the synchronous motor; and

A phase correction device for correcting the AC phase according to the state quantity.

In addition, as a device for driving a synchronous motor, the present invention does not use a complicated control system such as a rotation speed controller and a current controller, but constitutes a feed-forward control system based on a rotation speed command and a current command. At this time, the actual rotation speed is used. The torque current generates the torque current command, and reproduces the motor current based on the detection value of the DC current of the inverter in terms of current detection. By correcting the AC phase corresponding to the driving frequency of the motor based on the calculated value, the constant axis deviation is controlled to zero, the control system can be stabilized, and rotation at a high carrier frequency can be performed. Specifically, the present invention has an inverter that changes the output voltage of the DC power supply into a three-phase AC voltage of variable voltage and variable frequency in response to a pulse width control signal, and applies it to the synchronous motor; An inverter current detector for the inverter current in the above-mentioned inverter; a revolution command generator for generating a revolution command of the above-mentioned synchronous motor; generating the above-mentioned pulse width control signal according to the above-mentioned revolution command, and outputting it to the above-mentioned inverter A controller for an inverter, wherein the above-mentioned controller is composed of a sampling device that sequentially samples the inverter current detected by the above-mentioned inverter current detector; and reproduces the current flowing through the above-mentioned A current reproducing device for alternating current of a synchronous motor; transforming the coordinates of the alternating current reproduced by the above-mentioned current reproducing device into dq of the current on the dc axis assuming the magnetic pole axis inside the above-mentioned synchronous motor and the qc-axis orthogonal to the above-mentioned dc-axis Coordinate transformation means; According to the current component on the qc axis obtained by the coordinate transformation of the above-mentioned dq coordinate transformation means, a torque current command generating means for generating a torque current command about the above-mentioned synchronous motor; A current command, an applied voltage command calculation device for calculating each applied voltage command on the above-mentioned dc axis and the above-mentioned qc axis; a phase calculation device for calculating an AC phase related to the drive frequency of the above-mentioned synchronous motor according to the above-mentioned rotation number command; The AC phase calculated by the computing device converts the coordinates of the above-mentioned applied voltage commands into a dq inverse conversion device for a three-phase AC voltage command; a pulse width control signal generating device for generating a pulse width control signal according to the above-mentioned three-phase AC voltage command; the calculation and the above-mentioned The dc-qc axis and the d-q axis as the actual magnetic pole axis error angle of the above-mentioned synchronous motor are equivalent to a state quantity calculation device; a phase correction device for correcting the above-mentioned AC phase according to the above-mentioned state is constituted.

When configuring the speed control device for the synchronous motor described above, it is more preferable to add the following elements.

(1) The state quantity calculating means calculates the state quantity based on the difference between the current component on the qc axis obtained by the coordinate conversion by the dq coordinate transforming means and the torque current command generated by the torque current command generating means.

(2) If the q-axis inductance of the above-mentioned synchronous motor is denoted as Lq, the winding resistance is denoted as R, the current component on the qc-axis obtained through the coordinate transformation of the above-mentioned dq coordinate transformation device is denoted as Iqc, and the above-mentioned dq coordinates The current component on the dc axis obtained by the coordinate transformation of the conversion device is denoted as Idc, the electrical angular frequency command obtained from the above-mentioned rotation speed command is denoted as ω1 ^* , the applied voltage command on the above-mentioned dc axis is denoted as Vdc ^* , and the above The applied voltage command on the qc axis is denoted as Vqc ^* , then as the above-mentioned state quantity, the above-mentioned state quantity calculation device is based on the following formula (2):

Formula (2)

${\mathrm{<span\; class="notranslate">\; \&Delta;\&theta;</span>}}_{\mathrm{<span\; class="notranslate">\; c</span>}}<span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; the\; tan</span>}}^{<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}}\frac{{{\mathrm{<span\; class="notranslate">\; V</span>}}_{\mathrm{<span\; class="notranslate">\; dc</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; R</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; dc</span>}}<span\; class="notranslate">\; +</span>{{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}{\mathrm{<span\; class="notranslate">\; L</span>}}_{\mathrm{<span\; class="notranslate">\; q</span>}}<span\; class="notranslate">\; \&Center\; Dot;</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; qc</span>}}}{{{\mathrm{<span\; class="notranslate">\; V</span>}}_{\mathrm{<span\; class="notranslate">\; qc</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; R</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; qc</span>}}<span\; class="notranslate">\; -</span>{{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}{\mathrm{<span\; class="notranslate">\; L</span>}}_{\mathrm{<span\; class="notranslate">\; q</span>}}<span\; class="notranslate">\; \&Center\; Dot;</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; dc</span>}}}$

To calculate the axis error Δθc.

(3) Equipped with a voltage phase computing device for computing a voltage command phase based on the respective applied voltage commands on the dc axis and the above qc axis and the AC phase calculated by the phase computing device; the voltage command phase computed by the voltage phase computing device An interrupt signal generating means that outputs an interrupt signal for command sampling for each specific phase of the above-mentioned sampling means.

(4) Equipped with a polarity computing device for calculating the polarity of each phase of the three-phase AC voltage command output by the above-mentioned dq inverse conversion device, and outputting the polarity signal of each phase; in response to the polarity change of any of the above-mentioned phase polarity signals, for The above-mentioned sampling means outputs an interrupt signal generating means for an interrupt signal for instructing sampling.

(5) An absolute value calculation device is provided for calculating the absolute value of each phase of the three-phase AC voltage command output by the above-mentioned dq inverse conversion device; when the absolute values of two phases among the absolute values of the above-mentioned phases become similar values, for The above-mentioned sampling means outputs an interrupt signal generating means for an interrupt signal for instructing sampling.

(6) Subtracting means for calculating the difference between the current component on the dc axis obtained by the coordinate transformation of the dq coordinate transforming means and the d-axis current command; and correcting the applied voltage for calculating the qc axis based on the calculation result of the subtracting means Motor constant correction device for motor constant of voltage command.

(7) At least the above-mentioned inverter, the above-mentioned controller, and the above-mentioned inverter current detector are modularized.

In addition, the present invention provides an air conditioner including a synchronous motor and a speed control device for controlling the speed of the synchronous motor,

The speed control device of the synchronous motor is any one of the above-mentioned speed control devices of the synchronous motor.

According to the above-mentioned device, the state quantity caused by the shaft error Δθ of the d-q axis of the synchronous motor and the control axis dc-qc axis is calculated, and the state quantity is used as the correction amount to correct the electrical angular frequency command obtained from the rotation speed command. Drive frequency, calculate the AC phase from the drive frequency, and then reproduce the AC current of the synchronous motor according to the current obtained by sampling the inverter current, and perform dq coordinate transformation on the reproduced AC current according to the AC phase to obtain the torque current. Torque current generates the torque current command (q-axis current command), and calculates each applied voltage command on the dc-axis and qc-axis according to the electrical angular frequency command obtained from the torque current command and the number of revolutions command, and according to the AC phase The dq inverse transformation is performed on each voltage command to generate a three-phase AC voltage command, and a pulse width control signal is generated according to the three-phase AC voltage command, and the inverter is controlled according to the pulse width control signal, so even if there is no magnetic pole position sensor and no current sensor, it can Make the synchronous motor rotate stably and at high speed.

That is to say, in essence, feedback control is based on the state quantities corresponding to the error angles of the dc-qc axis and the d-q axis, only the control of correcting the axis deviation, and the correction loop gain used for the control of the axis deviation correction can be a response of about tens of milliseconds The processing time for reproducing the motor current from the inverter current and generating the actual torque current from the reproduced current is sufficient if it is performed in a processing cycle of about 1/5 of the correction loop gain. Therefore, even if the speed controller or the current controller is omitted, the synchronous motor can be rotated at a high speed in a stable state by delaying the time required for the process of detecting the torque current.

Description of drawings

Fig. 1 is a block diagram showing a system configuration of Embodiment 1 of a control device for a synchronous motor according to the present invention.

2 is a vector diagram showing the relationship between the d-q coordinate axis based on the magnetic pole axis of the synchronous motor and the dc-qc axis assumed for control.

Fig. 3 is a block diagram showing the internal structure of the ω1 corrector in Embodiment 1 of the present invention.

Fig. 4 is a block diagram showing the internal structure of a detection current processor in Embodiment 1 of the present invention.

Fig. 5 is a waveform diagram for explaining the operation of the detection current processor in Embodiment 1 of the present invention.

Fig. 6 is a block diagram showing the internal structure of a ω1 corrector in Embodiment 2 of the control device for a synchronous motor according to the present invention.

Fig. 7 is a block diagram showing an internal structure of a controller in Embodiment 3 of the speed control device for a synchronous motor according to the present invention.

Fig. 8 is a waveform diagram for explaining the operation of the controller in Embodiment 3 of the present invention.

9A to 9C are waveform diagrams for explaining the effect of the controller in Embodiment 3 of the present invention.

Fig. 10 is a block diagram showing the internal configuration of a controller in Embodiment 4 of the speed control device for a synchronous motor according to the present invention.

Fig. 11 is a waveform diagram for explaining the operation of the controller in Embodiment 4 of the present invention.

Fig. 12 is a block diagram showing the configurations of an absolute value calculator and an interrupt generator in Embodiment 5 of the speed control device for a synchronous motor according to the present invention.

Fig. 13 is a waveform diagram for explaining the operation of an absolute value calculator and an interrupt generator in Embodiment 5 of the present invention.

Fig. 14 is a block diagram showing the internal configuration of a controller in Embodiment 6 of the speed control device for a synchronous motor according to the present invention.

Fig. 15 is a perspective view showing the structure of Embodiment 7 of the speed control device for a synchronous motor according to the present invention.

Fig. 16 is a perspective view of applying the speed control device of the synchronous motor of the present invention to an outdoor unit of an air conditioner.

Detailed ways

Hereinafter, an embodiment of the present invention will be described with reference to the drawings.

Embodiment 1

Fig. 1 is a block diagram showing a system configuration of Embodiment 1 of a speed control device for a synchronous motor according to the present invention. In FIG. 1 , the speed control device of a synchronous motor is configured to include a revolution command generator 1 that generates a revolution command ωr ^* for providing a revolution command ωr ^* to a synchronous motor 5; the AC applied voltage of the synchronous motor 5 is calculated according to As a result of this operation, a pulse width modulation signal (PWM signal) is generated as a pulse width control signal, which is applied to the controller 2 on the inverter 3; the inverter 3 driven by the PWM signal; and supplied to the inverter 3 A DC power supply 4 for electric power; a current detector (inverter current detector) 6 that detects the inverter current I0 supplied to the inverter 3 from the DC power supply 4, and serves as a control object on the AC output side of the inverter 3 , for example, a synchronous motor 5 composed of a magnet motor is connected.

The controller 2 is configured to include a conversion gain unit 7; an integrator 8; a detection current processor 9; a torque current command (Iq ^* ) generator 10; an Id ^* generator 11; a voltage command calculator 12; ; PWM generator 14; ω1 modifier 15 and adder 16.

The conversion gain unit 7 uses the number of poles P of the synchronous motor 5 to transform the revolution command ωr ^* output by the revolution command generator 1 into the electrical angle frequency command (drive frequency command) ω1 ^* of the synchronous motor 5, and convert the converted electrical angle The frequency command ω1 ^* is output to the voltage command calculator 12 and the adder 16 . The adder 16 calculates the drive frequency ω1c by adding the electrical angular frequency command ω1 ^* to the correction amount Δω1 output by the ω1 corrector 15 , and outputs the calculation result to the integrator 8 . The integrator 8 is configured as a phase calculating device for calculating the AC phase θdc inside the control device to calculate the AC phase θdc related to the drive frequency of the synchronous motor 5 .

The detected current processor 9 is configured to calculate the current components of the synchronous motor 5 on the rotating coordinate axis (dc/qc axis) as Idc and Iqc based on the inverter current I0 detected by the current detector 6 . The torque current command generator 10 is configured to generate a torque current command by calculating the q-axis current command Iq as the torque current command based on the current component Iqc (actual torque current) on the qc axis output by the detection current processor 9. device. The Id ^* generator 11 is configured as a d-axis current command generator for generating a d-axis current command Id ^* . The voltage command calculator 12 is configured as an applied voltage command calculator for calculating voltage commands Vdc ^{*, Vqc* applied to the synchronous motor 5 on the dc-qc axis based on Id* , Iq *} , ω1 ^* . The dq inverse converter 13 is configured as a dq inverse conversion device for converting voltage commands Vdc ^* , Vqc ^* on the dc-qc axis into three-phase AC voltage commands vu ^* , vv ^* , vw ^* on the three-phase AC axis. The PWM generator 14 is configured to generate a PWM signal based on the three-phase AC voltage commands vu ^* , vv ^* , vw ^* , and output the generated PWM signal to the pulse width control signal generating device of the inverter 3 .

The ω1 corrector 15 is configured to calculate the state quantity generated by the axis error Δθ between the dq axis and the control axis dc-qc axis of the synchronous motor 5, and calculate the electrical angular frequency command (drive frequency command) ω1 for the synchronous motor 5 based on the calculation result. ^* The state quantity calculation device of the correction amount Δω1. The adder 16 is constituted as a phase correction means for calculating the drive frequency ω1c by adding the electrical angular frequency command ω1 ^* output from the conversion gain unit 7 and the correction amount Δω1 output from the ω1 corrector 15 . That is, the adder 16 adds the electrical angular frequency command ω1 ^* and the correction amount ω1 to correct the AC phase θdc based on the correction amount Δω1 as a state quantity, corrects the electrical angular frequency command ω1 ^* , and calculates the drive frequency ω1c.

The detected current processor 9 is configured to include a current sampler 91 ; a current reproducer 92 ; and a dq coordinate converter 93 . The current sampler 91 is configured to sequentially sample the instantaneous value of the inverter current I0 detected by the current detector 6 and output the sampled current to the sampling means of the current reproducer 92 . The current reproducer 92 is configured as a current reproducer for reproducing the AC currents Iuc, Ivc, and Iwc flowing into the synchronous motor 5 based on the sampled current values sampled by the current sampler 91 . The dq coordinate converter 93 is configured to convert the alternating current reproduced by the current reproducer 92 into current components on the dc axis assuming the magnetic pole axis inside the synchronous motor 5 and the qc axis orthogonal to the dc axis, that is, as a rotating coordinate axis. The dc coordinate transformation device of the current components Idc and Iqc on the dc-qc axis.

The inverter 3 is composed of a main circuit part 31 and a gate driver 32 that applies a gate pulse signal to each switching element of the main circuit part 31, wherein the main circuit part 31 is composed of switching elements Sup, Sun, Svp, Svn, Swp, Swn and diodes connected in antiparallel to each switching element.

The DC power supply 4 is configured to include a diode bridge 42 and a smoothing capacitor 43, rectifies the AC signal from the AC power supply 41, suppresses ripple components contained in the rectified signal by the smoothing capacitor 43, and applies a DC voltage V0 to the inverter. 3 on.

Next, the operating principle of Embodiment 1 will be described. The conversion gain unit 7 calculates the electrical angular frequency command ω1 ^* of the synchronous motor 5 based on the rotation command ωr ^* output from the rotation command generator, and outputs the calculation result to the voltage command calculator 12 and the adder 16 . In the voltage command calculator 12, the voltages Vdc ^* , Vqc ^* to be applied to the synchronous motor 5 are calculated by the following equation (3) from the electrical angular frequency ω1*, the current commands Id ^{* ,} Iq ^* .

[Formula 3]

${{\mathrm{<span\; class="notranslate">\; V</span>}}_{\mathrm{<span\; class="notranslate">\; dc</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; R</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>{{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; d</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; -</span>{{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}{\mathrm{<span\; class="notranslate">\; L</span>}}_{\mathrm{<span\; class="notranslate">\; q</span>}}<span\; class="notranslate">\; \&Center\; Dot;</span>{{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; q</span>}}}^{<span\; class="notranslate">\; *</span>}$

${{\mathrm{<span\; class="notranslate">\; V</span>}}_{\mathrm{<span\; class="notranslate">\; qc</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; =</span>{{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; \&CenterDot;</span>{\mathrm{<span\; class="notranslate">\; L</span>}}_{\mathrm{<span\; class="notranslate">\; d</span>}}<span\; class="notranslate">\; \&Center\; Dot;</span>{{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; d</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; R</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>{{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; q</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; K</span>}}_{\mathrm{<span\; class="notranslate">\; e</span>}}<span\; class="notranslate">\; \&CenterDot;</span>{\mathrm{<span\; class="notranslate">\; \&omega;</span>}\mathrm{<span\; class="notranslate">\; 1</span>}}^{<span\; class="notranslate">\; *</span>}$

In the formula, R: motor resistance, Ld: d-axis inductance, Lq: q-axis inductance, Ke: power generation constant of the motor.

Equation (3) is an arithmetic expression obtained from a general model of a synchronous motor, and the current commands Id ^* and Iq ^* supplied to the voltage command calculator 12 are generated by the Id ^* generator 11 and the Iq ^* generator 10, respectively. The d-axis current command Id ^* is usually given as Id ^* =0 when a non-saliency type motor is used as the synchronous motor 5 . On the other hand, when a salient-pole type motor is used as the synchronous motor 5 , a negative value is given in order to maximize the efficiency. Iq ^* , which is the torque current command, is obtained by calculation from the current detection value Iqc on the qc axis obtained by the current detection processor 9 .

Next, in the Iq ^* generator 10, Iq ^* is calculated, for example, according to the following equation (4).

[Formula 4]

${{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; q</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; T</span>}}_{\mathrm{<span\; class="notranslate">\; r</span>}}<span\; class="notranslate">\; \&CenterDot;</span>\mathrm{<span\; class="notranslate">\; the\; s</span>}}<span\; class="notranslate">\; \&Center\; Dot;</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; qc</span>}}$

In the case of vector control, Iq ^* is often provided as an output of the speed controller, but in the controller 2 of the present invention, Iq ^* is generated from the detected value Iqc.

That is, as is known from equation (4), in a constant state, since Iqc=Iq ^* , a voltage value necessary for the load condition is supplied from the control device to the synchronous motor 5, and vector control can be realized. As a result, compared with the conventional vector control, the control system can be greatly simplified, and the stability of the control system can be improved.

When the applied voltage Vdc ^* , Vqc ^* can be obtained according to the formula (3), in the dq inverse converter 13, the applied voltage Vdc ^* obtained in the formula (3), the coordinates of Vqc ^* are transformed into three-phase AC axis Three-phase AC voltage commands vu ^* , vv ^* , vw ^* . Next, in the PWM generator 14 , the AC voltage commands vu ^* , vv ^* , vw ^* are converted into PWM signals, and the converted PWM signals are output to the gate driver 32 . The gate driver 32 drives the switching elements Sup, Sun, Svp, Svn, Swp, and Swn based on the PWM signal (pulse signal), and applies voltages corresponding to Vdc ^* and Vqc ^* to the synchronous motor 5 .

On the other hand, in the ω1 corrector 15, as shown in FIG. 2 , the actual magnetic pole axis in the synchronous motor 5 is taken as the d-axis, and the axis perpendicular to the d-axis is taken as the q-axis, and further assumes The coordinate axis of is used as the dc/qc axis, and the state quantity corresponding to the axis error Δθ is used as the correction amount Δω1 for calculation.

Specifically, as shown in FIG. 3, the ω1 corrector 15 includes an adder 17 that is a subtractor that calculates (subtracts) the difference between Iq ^* and Iqc, and a proportional element that multiplies the output of the adder 17 by a gain K0. The correction gainer 18 constitutes. When Iq ^* and Iqc are in a constant state, they agree with each other, but when acceleration/deceleration or load disturbance occurs, a deviation occurs between the two. For example, if a load torque disturbance occurs, the dq axis lags behind the dc-qc axis, increasing the axis error Δθ. In this case, Iqc also increases. Conversely, when the load disturbance is reduced, its opposite phenomenon occurs. Thus, if the difference between Iq ^* and Iqc is observed, information on the axis error Δθ can be obtained. In addition, in the structure of FIG. 3, it is not necessarily limited to the fact that a correct Δθ value can be obtained. On the other hand, for the purpose of aligning the dc-qc axis with the dq axis, it is not necessary to calculate Δθ with high precision, but it is sufficient to know whether there is an axis deviation. The structure for making Δθ accurate is described in the second embodiment below.

Δω1 which is the output of the ω1 corrector 15 takes a "positive" value when the dc-qc axis lags behind the dq axis. If the electrical angular frequency command ω1 ^* is corrected according to the "positive" correction amount Δω1, the driving frequency ω1c of the synchronous motor 5 will increase, the dc-qc axis will return to the side of the dq axis, and the dc-qc axis will coincide with the dq axis, which can make The axis error Δθ becomes zero. Conversely, when the dc-qc axis is ahead of the dq axis, Δω1 becomes a "negative" value. If the electrical angular frequency command ω1 ^* is corrected according to the “negative” Δω1, the driving frequency ω1c of the synchronous motor 5 decreases, the AC phase θdc becomes negative sequentially, the dc-qc axis coincides with the dq axis, and the axis error Δθ can be made zero.

Next, a specific configuration of the detection current processor 9 will be described with reference to FIG. 4 . The detected current processor 9 is configured to include a current sampler 91 for sampling the inverter current I0 ; a current reproducer 92 ; and a dq coordinate converter 93 . The current sampler 91 is composed of a sampling time setter 911 that determines the timing for sampling the inverter current I0 according to the timing of the sequence command according to the three-phase AC voltage commands vu ^* , vv ^* , vw ^* ; according to the sampling time setting A timer 911, two timers 912a, 912b for setting the time of sampling/holding signals; receiving signals from each timer 912a, 912b, and sampling/holding the inverter current I0 2 sampling/holding devices ( S/H) 913a, 913b; a signal inverter 914 for inverting the sign of the signal.

The current reproducer 92 distributes the current obtained by sampling to the detection value distributor 921 of the three-phase current values Iuc, Ivc, and Iwc of the U, V, and W phases according to the three-phase AC voltage commands vu ^* ~vw ^* ; The signal from the detection value distributor 921 switches the three switches 922a, 922b, and 922c input from the current sampler 91; the subtractor 16 that calculates the difference (Imid) between the two current values Imax and Imin output from the current sampler 91 constitute.

In FIG. 4 , the current detection values Imax and Imin output from the current sampler 91 and Imid calculated in the current reproducer 92 are current values associated with the magnitude relationships of the three-phase AC voltage commands vu ^* to vw ^* , respectively. For example, when the relationship between the three-phase AC voltage commands vu ^* to vw ^* is vu ^* > vv ^* > vw ^* , Imax is the U-phase current, Imid is the V-phase current, and Imin is the W-phase current. This specific example will be described with reference to FIG. 5 .

In Figure 5, (a) is the three-phase AC voltage command vu ^* ~ vw ^* , the triangular wave and carrier wave used in the PWM signal, (b) is the waveform of the PWM pulse signal of each phase modulated by the pulse width, (c) is a switching pattern showing the switching state of the converter 3, (d) is the current waveform of the three-phase AC current flowing through the synchronous motor 5, and (e) is the current waveform of the converter current I0 detected by the current detector 6 , (f) is the waveform of the current Imax and Imin sampled by the current sampler 91 , and (g) is the waveform of the reproduced current Iuc to Iwc of each phase reproduced by the current reproducer 92 .

Fig. 5 shows an example in which the size relationship of the three-phase AC voltage command is vu ^* >vv ^* >vw ^* , if the triangular wave carrier frequency is sufficiently higher than the drive frequency ω1c of the synchronous motor 5, then the three-phase AC voltage command has a higher relative to the triangular wave carrier The period of one cycle of the waveform can be regarded as constant, and it becomes a waveform like (a). At this time, the waveform of the PWM pulse signal becomes as shown in (b). The PWM pulse waveform respectively means that when F=1 (the switching level is "1"), the switching elements Sup, Svp, and Swp on the upper side of the inverter 3 are turned on, and the switching elements Sun, Svn, and Swn on the lower side are turned off. broken. Now, assuming that the AC current of the synchronous motor 5 is (d), the inverter current I0 has a waveform like (e). Furthermore, in the case of (5), there are the following four switching patterns, and the current values in the respective patterns are as follows.

(1) Switch mode 1:

Switching element Sup=ON, Svp=ON, Swp=ON→I0=0

(2) Switch mode 2:

Sup=ON, Svp=ON, Swp=OFF→I0=Iu+Iv=-Iw

(3) Switch mode 3:

Sup=ON, Svp=OFF, Swp=OFF→I0=Iu

(4) Switch mode 4:

Sup=OFF, Svp=OFF, Swp=OFF→I0=0

That is, in switching mode 2, the current value of the phase with the smallest voltage command (W phase in this case) is observed, and in switching mode 3, the current value of the phase with the largest voltage command (in this case, U phase) is observed. Mutually). That is, within a half cycle of the triangular wave carrier, the commutator current I0 includes the current information of the "maximum voltage phase" and "minimum voltage phase".

Thus, if the commutator current I0 is sampled at the timing of the arrow (e), the current Imin of the phase with the lowest voltage (in this case, the W phase) and the current Imax of the phase with the highest voltage (in this case, W phase) can be sampled separately. is under the U phase) (Fig. 5(f)). The sampling timing is determined by the sampling time setter 911 . Through the sampling time setter 911, according to the relationship between the magnitude of the voltage command and the relationship between the switching mode, the sampling time for sampling the current of the phase with the largest voltage and the current of the phase with the smallest voltage is determined, and the two timers 912a are set according to the determined time. , 912b to set the sampling time. In the sample/hold units 913a and 913b, the inverter current I0 is sampled and held based on signals generated by the respective timers. In addition, since the sign of Imin is inverted, the sign is correctly corrected by the signal inverter 914 .

In addition, in the case of three-phase alternating current, as long as the neutral point is not connected, Iu+Iv+Iw=0 is established, so the current value of the voltage intermediate phase (V phase in this case) Imid can be calculated by using the subtractor 16 Imax Find the difference with Imin. In addition, in the current reproducer 92, Imax, Imin, and Imid are assigned to U, V, and W phases, respectively. That is, in the detection value distributor 921, the current detection value is distributed for each phase using the three switches 922a to 922c based on the magnitude relationship of the applied voltage command for each phase. When the current values of the respective phases are distributed, the current values Iuc, Ivc, and Iwc of the respective phases are converted into current components Idc, Iqc of the dc-qc axis using the dq coordinate converter 93 .

Based on the Iqc obtained in this way, Iq ^* is calculated according to (4), and the correction amount Δω1 is obtained by the ω1 corrector 15 from the difference between Iq ^* and Iqc obtained by the calculation, and the electrical angular frequency command Δω1 ^* is corrected by the correction amount Δω1 to generate The driving frequency ω1c can realize vector control.

Thus, in this embodiment, the operation of the detection current processor 9 is the most complicated. In particular, as the frequency of the triangular wave carrier increases, computing power becomes an important factor. However, in this embodiment, since it is a vector control with a different structure from the conventional "vector control with a sensor", it is possible to lengthen the calculation processing time for this control.

That is, in the control structure of this embodiment, as shown in FIG. 1 , the "feedback control" that is actually performed is only the control of the misalignment correction by the ω1 corrector 15 . The correction loop gain of ω1 may be a response time of several tens of milliseconds in applications such as fans, pumps, and air conditioner compressors, for example. Therefore, it is sufficient if the detection current processing of the detection current processor 9 is performed in a processing cycle of about 1/5 of the response time. That is, the current detection process can be performed in a cycle of several ms.

In contrast, in the conventional sensorless control based on "vector control with sensor" as the basic structure, since the feedback control system is composed of multiple speed controllers, current controllers, speed estimators, position estimators, etc., It is difficult to set the control response time of each element, and as a result, it is necessary to increase the calculation speed. As a result, the detection current processing also needs to be performed on the order of several hundreds of μs.

In this way, according to the present embodiment, since the correction amount Δω1 is obtained based on the difference between Iq ^* and Iqc, and the electrical angular frequency command ω1 ^* is corrected according to the correction amount Δω1 to obtain ω1c, the response time of the detection current processor 9 can be lengthened and the By making the synchronous motor 5 rotate stably and at high speed without using a magnetic pole position sensor or a current sensor, it is possible to minimize hardware components and simplify the control structure.

Implementation form 2

Next, Embodiment 2 of the speed control device for a synchronous motor according to the present invention will be described with reference to FIG. 6 . In this embodiment, instead of the ω1 corrector 15, the ω1 corrector 15B is used, and the other structures are the same as those in FIG. 1 . That is, in Embodiment 1, since the calculation of the axis error Δθ is simplified, and the axis error Δθ is controlled through the ω1 corrector 15, it is considered that if the conditions such as the rotation speed are different, the gain of the ω1 correction loop will change, which may damage the In order to control the stability of the system, the ω1 corrector 15B is used instead of the ω1 corrector 15 .

The ω1 corrector 15B consists of a shaft error calculator 19 that calculates the shaft error Δθ with high precision; setters 20a and 20b that set the winding resistance R as the motor constant; and setters that set the q-axis inductance Lq as the motor constant 21a, 21b; the multiplier 22a that multiplies Idc and ω1 ^* ; the multiplier 22b that multiplies ω1 ^* and Iqc; adds Vdc ^* , the output of the setter 20a, and the output of the setter 21b respectively An adder 17a as a subtractor for subtraction; an adder 17b as a subtractor for adding and subtracting the outputs of Vqc ^* and the setters 20b and 21a respectively; the arc is obtained from the outputs of the adder 17a and the adder 17b Arc tangent calculator 23 for tangent; zero command generator 24 that provides a command of "zero" for the axis error estimated value Δθc output by the operator 23; a subtractor that adds and subtracts the axis error estimated value Δθc and "zero" The adder 17c; Gain K as a proportional element of the gain setter 25 constitutes.

In the shaft error calculator 19, the shaft error Δθ is estimated and calculated according to the following equation (5) from Vdc ^* , Vqc ^* , ω1 ^* , Idc, and Iqc.

[Formula 5]

${\mathrm{<span\; class="notranslate">\; \&Delta;\&theta;</span>}}_{\mathrm{<span\; class="notranslate">\; C</span>}}<span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; the\; tan</span>}}^{<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}}\frac{{{\mathrm{<span\; class="notranslate">\; V</span>}}_{\mathrm{<span\; class="notranslate">\; dc</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; R</span>}<span\; class="notranslate">\; \&CenterDot;</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; dc</span>}}<span\; class="notranslate">\; +</span>{{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}{\mathrm{<span\; class="notranslate">\; L</span>}}_{\mathrm{<span\; class="notranslate">\; q</span>}}<span\; class="notranslate">\; \&CenterDot;</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; qc</span>}}}{{{\mathrm{<span\; class="notranslate">\; V</span>}}_{\mathrm{<span\; class="notranslate">\; qc</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; R</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; qc</span>}}<span\; class="notranslate">\; -</span>{{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}{\mathrm{<span\; class="notranslate">\; L</span>}}_{\mathrm{<span\; class="notranslate">\; q</span>}}<span\; class="notranslate">\; \&CenterDot;</span>{\mathrm{<span\; class="notranslate">\; I</span>}}_{\mathrm{<span\; class="notranslate">\; dc</span>}}}$

That is, in the shaft error calculation 19, the calculation of the numerator in the formula (5) is performed by the setter 20a, the multiplier 22b, the setter 21b, and the adder 17a, and the multiplier 22a, the setter 21a, and The unit 20b and the adder 17b calculate the denominator of the formula (5). The arc tangent calculation unit 23 calculates the axis error estimated value Δθc from the calculation result.

In this way, in this embodiment, when obtaining the estimated axis error value Δθc, more input information than that of the ω1 corrector 15 is used to obtain it, so that the estimated axis error value Δθc can be calculated more accurately than in the above-mentioned embodiment. Able to contribute in improving the performance of sensorless control.

In addition, in the ω1 corrector 15B, "zero" as the target value of the axis error estimated value is supplied from the zero command generator 24 to the adder 17c, and the output of the adder 17c is multiplied by the proportional gain K to perform correction, so the gain The set value of the setter 25 is an amount directly related to the response to determine the ω1 correction loop gain. As a result, there is no dependence of the control system on speed conditions or load conditions, and the overall response characteristics of the control system can be improved compared to the above-described embodiment.

Implementation form 3

Next, Embodiment 3 of the control device for a synchronous motor according to the present invention will be described with reference to FIG. 7 .

In this embodiment, the voltage phase calculating unit 26 as a voltage phase calculation means for calculating the voltage command phase θv based on the AC phase θdc output by the integrator 8 and the applied voltage command Vdc ^* is newly provided, and each specific phase of the voltage command phase θv is newly provided. The interrupt generator 27 as an interrupt signal generating means for outputting an interrupt signal S for instruction sampling by the detection current processor 9 constitutes a controller 2C, and other structures are the same as those in FIG. 1 .

The basic operation of the controller 2C in this embodiment is almost the same as that in the first embodiment. However, when the detected current processor 9 is operated, an interrupt signal is generated as a trigger pulse at a specific timing of the voltage command phase θv.

Specifically, the voltage command phase θv is calculated in the voltage phase calculator 26 according to the following equation (6).

[Formula 6]

$\mathrm{<span\; class="notranslate">\; \&theta;v</span>}<span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; \&theta;</span>}}_{\mathrm{<span\; class="notranslate">\; dc</span>}}<span\; class="notranslate">\; +</span>\frac{\mathrm{<span\; class="notranslate">\; \&pi;</span>}}{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; the\; tan</span>}}^{<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 1</span>}}<span\; class="notranslate">\; (</span><span\; class="notranslate">\; -</span>\frac{{{\mathrm{<span\; class="notranslate">\; V</span>}}_{\mathrm{<span\; class="notranslate">\; dc</span>}}}^{<span\; class="notranslate">\; *</span>}}{{{\mathrm{<span\; class="notranslate">\; V</span>}}_{\mathrm{<span\; class="notranslate">\; qc</span>}}}^{<span\; class="notranslate">\; *</span>}}<span\; class="notranslate">\; )</span>$

(6) The relationship between θv and voltage command phase is shown in Figure 8(a). In accordance with the value of the voltage command phase θv, in the interrupt generator 27, an interrupt signal S is generated at the timing shown in FIG. 8(b). The interruption signal S occurs at the time of θv=30 degrees, 90 degrees, 150 degrees, . . . , 330 degrees, respectively. If the interrupt signal S is input into the detected current processor 9, the current sampler 91 of the detected current processor 9 uses the interrupt signal S as a trigger pulse to sequentially sample the inverter current. If the current sampler 91 sequentially samples the inverter current at intervals of θv=60 degrees, the following effects can be obtained.

Specifically, as shown in Figure 9(A), in the three-phase AC voltage command, if the waveform of the inverter current I0 at point A (near θv=30 degrees) (Figure 9B) is compared with point B (θv = around 60 degrees) compared with the waveform (FIG. 9C) of the inverter current I0, it can be seen that a large difference occurs in the pulse width of the inverter current I0. In the case of sampling at point A, the currents of Iu and Iw have the same flow period as I0, and each has a wide pulse width. However, when sampling at point B, the flow period of Iu is short. In the actual waveform of the inverter current I0 , since coupling occurs due to the switching operation, it is extremely difficult to sample a current with a narrow pulse width like Iu in FIG. 9C . Also, when θv=60 degrees is correct, the period of Iu becomes completely 0. That is, when the detection current processing is performed near the point B, the motor currents of all three phases cannot be reproduced, and only Iw can be reproduced. The higher the carrier frequency, the wider the scope of the phenomenon of "current reproducibility not possible", which is an essential problem of the method of detecting the motor current using the inverter current I0.

However, in the case of the controller 2C in this embodiment, normally, the detection current processor 9 is operated only at the timing shown in FIG. 9B , so the undesirable situation shown in FIG. 9C does not occur. As shown in FIG. 9B, the current can always be detected at a timing with good conditions.

In addition, in the present embodiment, current detection processing is performed only 6 times (every 60 degrees) with respect to one cycle period (0<θv<360 degrees) of the voltage phase command. In this case, although there is concern about the influence of the current detection delay, it is not a problem with the configuration of the controller 2C in this embodiment. That is, as described in the first embodiment, in the controller 2C, since the feedback control is only the axis error control, it is easy to stabilize the system, and it is possible to stabilize the system even if the control response is lowered. Also, as described in Embodiment 1, the detected current processor 9 can be executed in a cycle of several ms. Assuming that the current detection process is performed every 5 ms, the basic frequency can be used as long as it is 33 Hz (1/(0.005×6)) or higher. In the case of the synchronous motor 5, since there are many applications for high-speed rotation with a fundamental frequency of several 100 Hz, the present invention can be applied in almost all frequency bands.

According to this embodiment, the synchronous motor 5 can always be rotated at high speed with a stable control system even without a magnetic pole position sensor or a current sensor.

Embodiment 4

Next, a fourth embodiment of the present invention will be described with reference to FIG. 10 . In this embodiment, a controller 2D is used instead of the controller 2, and the other structures are the same as those in FIG. 1 .

Specifically, the signs (polarities) of the AC voltage commands vu ^* to vw ^* of each phase are newly added, and a sign calculator 28 as a polarity calculation device that outputs polarity signals of each phase is newly added; according to the sign calculator 28 The output polarity signal is outputted to the interrupt generator 27D as an interrupt signal generating means which outputs the sampling signal S for command sampling to the detection current processor 9 .

Next, the operation of the controller 2D will be described. The basic operation of the controller 2D is almost the same as that of the first embodiment. However, when operating the detected current processor 9, it is characteristic that an interrupt signal is generated as a trigger pulse at the timing when the polarities of the three-phase AC voltage commands vu ^* to vw ^* change.

Specifically, in Embodiment 3, when generating the interrupt signal S, the voltage phase command θv is used, and as shown in the formula (6), in the calculation of θv, it is necessary to use the arc tangent, which is to obtain the value of θv Complicated processing takes time. Furthermore, in order to monitor the timing of the interrupt signal S, it is necessary to perform calculation every time the voltage command phase is updated. Therefore, in Embodiment 3, this processing becomes a bottleneck, and the overall calculation time, the value of the carrier frequency, and the like are limited.

On the other hand, in this embodiment, in order to solve the problem in the third embodiment, the polarity information of the AC voltage command is used. Specifically, as shown in FIG. 11 , with respect to the AC voltage commands vu ^* to vw ^* , as shown in FIGS. 11( b ) to 11 ( d ), the change in polarity of each phase is obtained by the sign calculator 28 . The applied voltage command for each phase occurs at the timing of θv = 30 degrees, 90 degrees, 150 degrees, . . . , 330 degrees. Therefore, the interrupt generator 27D uses the timing of θv=30 degrees, 90 degrees, 150 degrees, ..., 330 degrees, that is, the rising and falling timings of the polarity signals pu, pv, pw of each phase, as shown in (e) As shown, an interrupt signal S occurs. As a result, since the same signal as that of FIG. 8( b ) can be obtained, the interrupt signal S can be generated without the need for calculations such as the formula (6).

As described above, according to the present embodiment, it is possible to realize a high-performance speed control device for a synchronous motor with a simpler structure than that of the third embodiment.

Embodiment 5

Next, Embodiment 5 of the present invention will be described with reference to FIG. 12 . In this embodiment, an absolute value calculator 29 is used instead of the sign calculator 28, and an interrupt signal generator 27E is used instead of the interrupt signal generator 27D. The other structures are the same as those of the fourth embodiment.

The absolute value computing unit 29 is configured as an absolute value computing device that computes and outputs the absolute values of the three-phase AC voltage commands vu ^* to vw ^* , and the interrupt generator 27E is configured so that when there are two phases in the absolute value output by the absolute value computing unit 29 When the absolute value becomes an approximate value, an interrupt signal generator that outputs an interrupt signal S for instructing sampling to the detection current processor 9 .

In the fourth embodiment, the polarity of the voltage command phase is used, and an interrupt signal is generated when the sign of the phase is reversed. However, if this processing is implemented by software, a delay of an operation cycle will be generated in order to detect the reversed sign. That is, since the inversion of the sign becomes a comparison with the previous value, there will be a delay anyway. In particular, when the fundamental frequency is high and close to the carrier frequency, the delay increases, making it difficult to perform detection current processing under ideal conditions as shown in FIG. 9(A) .

Therefore, in Embodiment 5, in order to solve such a problem, the absolute value of the three-phase AC voltage command is calculated by the absolute value calculator 29, and the calculation result is output to the interrupt generator 27E. In the interrupt generator 27E, as shown in the figure As shown in 13, the absolute value of the applied voltage command of each phase is compared, two larger values are selected among the absolute values of each phase, and the difference between the two selected is calculated. When the difference becomes "zero", for example, or When the value falls below a predetermined value, that is, at a timing closer to both values, for example, at a timing faster than θv=30 degrees, 90 degrees, 150 degrees, . . . , 330 degrees, an interrupt signal S is generated. By performing such processing, as in the fourth embodiment, no delay occurs in the interrupt signal S, and conversely, the generation timing of the interrupt signal can be advanced.

In this manner, according to the present embodiment, the activation timing of the detection current processor 9 can be arbitrarily shifted according to circumstances, and the degree of freedom of setting can be increased. In addition, although the same effect can be obtained by using the third embodiment, the processing is complicated because the formula (6) is used. On the other hand, in this embodiment, only the magnitudes of the absolute values of the AC voltage commands are compared, so that the calculation process can be simplified.

As described above, according to the present embodiment, a speed control device for a synchronous motor with higher performance can be realized with a simpler structure than that of the fourth embodiment.

Embodiment 6

Next, the configuration of Embodiment 6 of the present invention will be described with reference to FIG. 14 . In this embodiment, a controller 2F is used instead of the controller 2, and the other structures are the same as those in FIG. 1 .

In the controller 2F of the present embodiment, a subtracter 35 as a subtracting means for computing the difference between the d-axis current command Id ^* and Idc, and a device for correcting the voltage command calculator 12F based on the output of the subtractor 35 are newly provided. A current controller 36 with a fixed value Ke. The current controller 36 is configured as a motor constant correcting device for correcting a motor constant for calculating an applied voltage command on the qc axis based on the output of the subtracter 35 .

That is, in each of the above-described embodiments, the feedback control system in the controller is only a control system for correcting the axis error, and as a result, the following problems may arise. For example, since there is no control over the magnitude of the current flowing into the synchronous motor 5, the setting value shown in (3) formula is the entire value, and if there is a deviation in the motor constant used here, the motor current becomes equal to the command different currents. For example, even if there is no load, excessive no-load current may occur, or under-voltage may occur under load, causing unfavorable conditions such as out-of-step. In particular, since the synchronous motor 5 is characterized as a high-performance electrode, it is not necessary to flow a large amount of reactive current. In order to maintain the maximum efficiency of the synchronous motor 5, a mechanism for controlling the d-axis current component in accordance with commands is required.

Therefore, in order to solve such a problem, in the sixth embodiment, the set value Ke in the voltage command calculator 12F is corrected using the subtractor 35 and the current controller 36 . In the current controller 36, in the difference between Id ^* and Idc, it is considered that the motor constant in the formula (3) is affected by the setting error of the electric power generation constant Ke of the motor, and the difference between Id ^* and Idc is used as The current controller 36 as an integral element calculates ΔKe, and corrects the set value Ke in the voltage command calculator 12F based on ΔKe. The term of the generation constant Ke greatly affects the applied voltage of the synchronous motor 5 . Therefore, in order to correct the term of the power generation constant Ke, it is most effective to make Idc coincide with Id ^* . Of course, in the low-speed range, since the R term has a greater influence than the Ke term, R may be corrected in this case as well.

In addition, the current controller 36 can be realized using only an integral element. In addition, in this control, in order to correct the deviation of the constant Idc, the control response may be delayed. That is, since it is possible to lag behind the shaft error control system, the configuration of the sixth embodiment can be applied to each of the above-mentioned embodiments.

As described above, according to the present embodiment, it is possible to cope with fluctuations in generator constants or setting errors in generator constants, so that a high-performance speed control device for a synchronous motor can be realized.

Implementation form 7

Next, the structure of Embodiment 7 of the present invention will be described with reference to FIG. 15 . In this embodiment, the controller 2, the inverter 3, the current detector 6 and the diode bridge 42 are integrated into one module. When carrying out this modularization, set the revolution command terminal from the revolution command generator 1 composed of microcomputer, the input terminal of AC power supply 41, the connection terminal of filter capacitor 43 and the connection terminal of synchronous motor 5, and all other parts contained within the module. A controller 2 using a microcomputer, an inverter 3 composed of a switch driver, a current detector 6 composed of a shunt resistor, and a diode bridge 42 are accommodated in the module.

When the controller 2 etc. are modularized, if the parts of the above-mentioned embodiments are used, the speed control device of the synchronous motor without a magnetic pole position sensor and a current sensor can be realized with high-performance parts, and at the same time, it can be realized with an inexpensive microcomputer. , which can be easily modularized.

In this manner, according to the present embodiment, the power module can be handled as a single component, which facilitates assembly and enables downsizing of the entire device.

Embodiment 8

Next, the configuration of Embodiment 8 of the present invention will be described with reference to FIG. 16. FIG. This embodiment is an example in which the speed control device for a synchronous motor of the present invention is applied to an outdoor unit of an air conditioner, and the speed control device used in any one of Embodiments 1 to 7 is installed inside the outdoor unit 37 of the air conditioner. At the same time, the synchronous motor 5 serving as a power source is accommodated in the air conditioner compressor 38 .

In order to make the inside of the compressor 38 a high-temperature and low-pressure environment, it is necessary to use a position sensorless motor installed inside.

Therefore, when applying a speed control device without a position sensor and without a current sensor to an outdoor unit of an air conditioner, the speed control device for a synchronous motor of the present invention is used. The speed control device for a synchronous motor of the present invention has the feature of being able to realize no magnetic pole position sensor, and also can realize no current sensor. As a result, the overall structure of the device can be simplified, and the control device itself can be downsized, while the wiring process inside the outdoor unit 37 can be shortened, and the overall size of the device can be realized.

As described above, according to the present invention, a synchronous motor can be stably rotated at high speed even if it is without a magnetic pole position sensor or a current sensor.

Claims (1)

1. A control device for a permanent magnet motor, which controls the permanent magnet motor through the motor current, wherein the above motor current is calculated according to the current value detected by the DC shunt resistor connected to the DC side of the inverter Motor current, characterized by: A current command value given to the torque shaft component of the motor is generated based on the torque shaft component of the motor current, a pulse width control signal for driving the inverter is generated based on the current command value, and the pulse width control signal is given to The above-mentioned inverter is used to drive the above-mentioned motor.

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