CN1327617C - Low-pass filter,feedback system and semiconductor integrated circuit - Google Patents

Abstract

The invention provides a low-pass filter suitably used as a loop filter for a PLL or a DLL that has a filtering characteristic equivalent to that of a conventional one and can be realized in a smaller circuit area. The low-pass filter includes first filtering means (31) for accepting, as an input, an input signal to the low-pass filter and outputting a first voltage; a circuit element (311) included in the first filtering means (31) for allowing a first current to flow in accordance with the first voltage; current generating means (32) for generating a second current at a given rate to the first current; second filtering means (33) for accepting, as an input, the second current and outputting a second voltage; and adding means (34) for adding the first voltage and the second voltage and outputting an output signal of the low-pass filter, in which the second current is set to be smaller than the first current.

Description

Low-pass filter circuit, reponse system and semiconductor integrated circuit

Technical field

The present invention relates to a kind of low-pass filter circuit, particularly

Belong to the low-pass filter circuit that is fit to very much to do the loop filter in phase-locked loop etc. and uses, technology with phase-locked loop of such low-pass filter circuit.

Background technology

Phase-locked loop (below, be called " PLL ") become indispensable inscape in the semiconductor IC system now, phase-locked loop all is installed among nearly all LSI.And its range of application also is to cross over various fields such as microprocessor, IC-card headed by the communicating machine.

Figure 32 represents the structure of general charge-pump type PLL.The summary of PLL is described with reference to this figure.Phase comparator 10 relatively offers the input clock CKin of PLL and the phase difference between the feedback clock CKdiv and output rises signal UP and falls signal DN with this phase difference is corresponding; Charge pump circuit 20 is according to rising signal UP and falling signal DN output charging current Ip; Loop filter 30 is exported as voltage Vout with charging current Ip smoothing and with it; Voltage-controlled oscillator 40 changes the frequency of the output clock CKout of PLL according to voltage Vout; Fractional-N frequency device 50 will be exported clock CKout and be divided into N frequency and feed back in the phase comparator 10 as feedback clock CKdiv with it.Repeat aforesaid operations, output clock CKout converges to the frequency of defined and locked gradually.

Loop filter 30 is the key element of particular importance in the inscape of above-mentioned PLL, we can say that the response characteristic of PLL is decided by the filtering characteristic of loop filter 30.

Figure 33 represents general loop filter.Shown in wherein (a) is passive filter, and shortcoming is that characteristic can be because at back level connecting circuit and variation etc.Be active filters for overcoming this shortcoming, be shown among the figure (b).Need mention, both transmission characteristics can be carried out equivalence transformation.Like this, generally speaking, be to realize loop filter 30 by the low-pass filter circuit that constitutes by combined resistance element R and capacity cell C.

According to the control theory of PLL, ideal situation is, maximum also be the responsive bandwidth with PLL be set in clock about 1/10th frequency on.According to this theory, the PLL as input just must make the cut-off frequency of loop filter lower, make responsive bandwidth narrower with the input clock of lower frequency.Therefore, the loop filter among the existing P LL has bigger time constant, and just bigger CR is long-pending.Long-pending for realizing bigger CR, generally be to make capacity cell bigger.

On the other hand, the response speed of PLL by damping coefficient about.Damping coefficient changes along with the frequency of the input clock of PLL, but is that the response of stablizing PLL also preferably makes it necessarily constant.Therefore, till now, be among the PLL of input at frequency input clock with broader bandwidth, use can change the loop filter of filtering characteristic and adjust damping coefficient.

Figure 34 shows the existing loop filter that can change filtering characteristic.Contain resistor ladder (resistor ladder circuit) 100 in the loop filter shown in Figure 34 (a).Shown in Figure 34 (b), resistor ladder 100 is made of a plurality of resistive elements and switch, just suitably control switch has various resistance values.Generally speaking, used the loop filter that contains such resistor ladder 100 among the PLL.

Figure 35 shows existing another loop filter that can change filtering characteristic.Loop filter 30 shown in this Fig comprises: integrating circuit 30-1, see-saw circuit 30-2 and adder 30-3.Integrating circuit 30-1 carries out integration to the charging current Ip1 from first charge pump circuit 20a output, and the voltage behind the output smoothing; See-saw circuit 30-2 is to from addition anti-phase amplification of the charging current Ip2 of second charge pump circuit 20b output; Adder 30-3 obtains the output voltage of loop filter 30 with the output of integrating circuit 30-1 and the output addition of see-saw circuit 30-2.In such loop filter 30,, just can adjust (for example speciallyying permit communique No. 2778421) by suitably changing the ratio of charging current Ip1 and charging current Ip2 to the damping coefficient of PLL with reference to Japan.

As mentioned above, be among the PLL of input at input clock with lower frequency, use large-scale capacity cell for the CR that increases loop filter is long-pending.And, in the PLL of input clock for input of the frequency of wide bandwidth, be to adjust damping coefficient, be necessary to use resistor ladder shown in Figure 34, a plurality of charge pump circuits and operational amplifier shown in Figure 35.These all are the main causes that circuit scale is increased.

For being difficult to will large-scale capacity cell to be arranged in the such product in outside in the application product of PLL, the circuit area that how to reduce PLL becomes extremely important.Particularly, in IC-card,, must avoid installing the parts of thickness more than card from the viewpoint of reliability.Therefore, come down to jumbo capacity cell can not to be installed to outside.The circuit area that reduces PLL just become must solution problem.And such LSI also is the same beyond the weld pad zone for PLL is installed in.

In addition, ideal situation is, among the LSI of chip on chip (chip on chip) structure, the PLL scale that is contained in the chip on upper strata is littler.And, because used a lot of PLL in the microprocessor, so the PLL circuit area is very big to the influence of whole microprocessor circuit area.

Summary of the invention

The present invention develops for addressing the above problem just.Its purpose is: a kind of filtering characteristic and existing the same, the low-pass filter circuit that can realize with littler circuit area are provided and have the PLL etc. that such low-pass filter circuit is made loop filter.Particularly, its purpose is: realize the capacity cell that low-pass filter circuit but need not be large-scale.Another object of the present invention is: make the filtering characteristic of such low-pass filter circuit variable and can adjust the damping coefficient of PLL.

Fig. 1 shows the structure as the low-pass filter circuit of the technical solution adopted in the present invention.Low-pass filter circuit 30 of the present invention comprises: the input signal with this low-pass filter circuit is input, is first filter 31 of output with first voltage; Be the electric circuit element that first filter 31 is had, the circuit element 311 that allows first electric current flow according to described first voltage; Produce the current generator 32 that becomes second electric current of certain ratio with described first electric current; With described second electric current is input, is second filter 33 of output with second voltage; With described first voltage and the described second voltage addition, obtain the adder 34 of the output signal of this low-pass filter circuit 30.

According to the present invention, the input signal of low-pass filter circuit has been made Filtering Processing for the first time by first filter 31, and exports first voltage.At this moment, produce second electric current by current generator 32, second electric current becomes certain ratio for first electric current that flows into the circuit element of first filter according to first voltage relatively.Because current generator 32 is not the device that amplifies first electric current itself, so the output of first filter 31 can not be affected owing to the generation of second electric current.And, not to flow into first electric current in second filter 33 of first filter, 31 back levels, but flow into second electric current that produces by current generator 32.Second electric current carries out Filtering Processing and export second voltage second time by second filter 33.At last, with first voltage and the second voltage addition, and obtain the output signal of this low-pass filter circuit by adder 34.

In the aforesaid low-pass filter circuit of structure, by changing certain ratio of relative first electric current of second electric current, change the transmission characteristic of second filter therefrom, just from the second original voltage of second filter output, low-pass filter circuit looks and just shows original transmission characteristic.In other words, though transmission characteristic since by changing second Filter Structures so that circuit scale is smaller changes, also can change and change above-mentioned certain ratio according to this, the result is that the filtering characteristic of low-pass filter circuit can not change.Therefore, according to the present invention, can realize that filtering characteristic is the same with prior art, circuit area but can be than the little low-pass filter circuit under the prior art.

In low-pass filter circuit of the present invention, preferably described certain ratio is the positive number less than 1.So, because generate second electric current littler, so just enough as the capacity cell that constitutes second filter with the capacity cell of little capacitance than first electric current.Therefore, can make second Filter Structures smaller, the result is the circuit area that can reduce low-pass filter circuit.

Particularly, the described current generator in the low-pass filter circuit of the present invention can be input in order to described first electric current, and described second electric current is realized for the current mirroring circuit of output.Input one side at this current mirroring circuit is provided with first semiconductor element that shows first coefficient of conductivity.This first semiconductor element also is the circuit element that first filter is had.Output one side at current mirroring circuit is provided with demonstration becomes second coefficient of conductivity of described certain ratio with described first coefficient of conductivity second semiconductor element.This circuit element can be described first semiconductor element of replacement, shows the resistive element of the resistance value that is equivalent to described first coefficient of conductivity.At this moment, described current mirroring circuit replaces described first electric current, is input with the 3rd electric current that is equivalent to described first electric current.

Particularly, the described circuit element in the low-pass filter circuit of the present invention can be by showing first coefficient of conductivity, being that first voltage current transformating circuit of described first electric current is realized with described first voltage transformation; Described current generator can be by showing second coefficient of conductivity, being that second voltage current transformating circuit of described second electric current is realized with described first voltage transformation.Only, the ratio of this first and second coefficient of conductivity is described certain ratio.

Described adder in the low-pass filter circuit of the present invention particularly, can be realized by operational amplifier, operation transconductance amplifier.When realizing with operational amplifier, it is the input to in-phase input end, the device that the output tertiary voltage is made the output signal of low-pass filter circuit for described second filter that has part ownership in negative feedback, with described first voltage.And when realizing with operation transconductance amplifier, be the device that an output the 3rd electric current is made the output signal of low-pass filter circuit.

In low-pass filter circuit of the present invention, preferably, described first semiconductor element, for: according to the first transistor that shows described first coefficient of conductivity for first bias current that comes; Described second semiconductor element, for: according to the transistor seconds that shows described second coefficient of conductivity for second bias current that comes; Described first and second bias current, its size changes according to shared bias control signal.The size variation of first and second bias current means that first and second coefficient of conductivity changes.After first and second coefficient of conductivity changes, just can dynamically change the filtering characteristic of low-pass filter circuit.And, the change of this filtering characteristic can based on first and second bias current shared bias control signal carry out.Therefore, just need be in order to the resistor ladder of the filtering characteristic that changes low-pass filter circuit, and can dwindle circuit scale further.And, under the situation that the loop filter in making phase-locked loop etc. is used, be easy to adjust damping coefficient, so very effective according to bias control signal.

Equally, preferably, described first and second voltage current transformating circuit also is based on the circuit that shared bias control signal changes described first and second coefficient of conductivity respectively.So, the same with above-mentioned reason, can further dwindle circuit scale.

Can the switch that have or not of the output of the electric current that switches described second to n the semiconductor element of inflow from the semiconductor element of described second to n (n is the natural number 3 or more), respectively be set in output one side of the described current mirroring circuit of low-pass filter circuit of the present invention; Any or any several total that this switch will flow in the electric current of described second to n semiconductor element are exported as described second electric current.So, just can ladder ground switch the size of second electric current, just digitally switch the size of second electric current.

Being in the structure of resistive element with described circuit element, can replace described resistive element, and being set, resistor ladder makes described circuit element.In this case, resistor ladder can change described resistance value according to the variation of the described coefficient of conductivity.So, just can allow the filtering characteristic of low-pass filter circuit dynamically change.

The technical solution adopted in the present invention is such: as reponse system, comprising: the charge pump circuit that produces charging current based on the phase difference of described input clock and the clock that fed back; With described charging current is the loop filter of input; And the output clock generator that produces described output clock based on output signal from described loop filter.Described loop filter comprises again: with described charging current is input, is first filter of output with first voltage; The circuit element that is had for described first filter, the circuit element that allows first electric current flow according to described first voltage; Generate the current generator that becomes second electric current of certain ratio with described first electric current; With described second electric current is input, is second filter of output with second voltage; And, export the adder of described output signal with described first voltage and the described second voltage addition.So, make the loop filter of reponse system, just can reduce the circuit area of reponse system by using structure low-pass filter circuit as shown in Figure 1.

The reponse system here, what say is: allow the output clock that produces based on input clock feed back (feedback clock), the feedback circuit that allows this output clock have desirable characteristic.Can list as its typical example: based on input clock produce the output clock of desirable frequency phase-locked loop, produce the delay-locked loop circuit etc. that relative input clock has the output clock of a desirable phase delay.

Particularly, described output clock generator, for: allow described output clock oscillation, the voltage-controlled oscillator that allows frequency of oscillation change based on described output signal from described loop filter.So, can realize the phase-locked loop that circuit area is less.

Particularly, described output clock generator, for: based on described input clock and from the described output signal of described loop filter, the voltage control delay circuit that allows the retardation of the described relatively input clock of described output clock change.So, can realize the delay-locked loop circuit that circuit area is less.

Preferably, described circuit element can change the shown coefficient of conductivity; Described reponse system comprises: the offset controller that allows the coefficient of conductivity of described circuit element and described charging current change according to shared bias control signal.So, because the shown coefficient of conductivity and the charging current of circuit element changes based on shared bias control signal, so damping coefficient can be remained on the certain value.

Particularly, described current generator, for: have according to first field-effect transistor that shows first coefficient of conductivity for first bias current that comes in input one side, having second bias current demonstration that supply according to institute in output one side, to become second field-effect transistor of second coefficient of conductivity of described certain ratio with described first coefficient of conductivity be input with described first electric current, with described second electric current current mirroring circuit that is output.At this moment, described circuit element is described first field-effect transistor.And described substrate bias controller is the controller that is allowed described first and second bias current, described charging current change by described bias voltage control signal.

Particularly, described circuit element, for: show first coefficient of conductivity, be first voltage current transformating circuit of described first electric current with described first voltage transformation; Described current generator, for: show become second coefficient of conductivity of described certain ratio with described first coefficient of conductivity, be second voltage current transformating circuit of described second electric current with described first voltage transformation.These first and second voltage current transformating circuits can change described first and second coefficient of conductivity respectively.And described substrate bias controller is the controller that is allowed described first and second coefficient of conductivity, described charging current change by described bias voltage control signal.

Preferably, described bias control signal is based on that described output signal from described loop filter produces.So, just can allow the coefficient of conductivity and the charging current of described circuit element of loop filter suitably change based on the output of loop filter.In other words, can realize based on the reponse system that suitably changes response characteristic from the output of loop filter.

Preferably, the described adder that described loop filter is had is an operational amplifier, makes described offset controller for change the device of described bandwidth of operational amplifiers characteristic according to described bias control signal.

Description of drawings

Fig. 1 is the structure chart of low-pass filter circuit of the present invention.

Fig. 2 is the circuit diagram of the low-pass filter circuit among first embodiment of the present invention.

Fig. 3 is the figure of the transmission characteristic of the low-pass filter circuit in the key diagram 2.

Fig. 4 is the circuit diagram the during reflectance of the current mirroring circuit in the low-pass filter circuit among Fig. 2 for a change.

Fig. 5 is the circuit diagram of a concrete offset compensator in the low-pass filter circuit of Fig. 2.

Fig. 6 is the circuit diagram of a concrete offset compensator in the low-pass filter circuit of Fig. 2.

Fig. 7 is provided with the circuit diagram of migration with the low-pass filter circuit of duplicate circuit.

Fig. 8 is the circuit diagram of a concrete bias regulator in the low-pass filter circuit of Fig. 2.

Fig. 9 is the circuit diagram of the temperature-compensation circuit used as the bias regulator in the low-pass filter circuit of Fig. 2.

Figure 10 is the variation of the low-pass filter circuit of first embodiment.

Figure 11 is the circuit diagram of the low-pass filter circuit of second embodiment of the present invention.

Figure 12 is the circuit diagram of the low-pass filter circuit of the 3rd embodiment of the present invention.

Figure 13 is the circuit diagram of the low-pass filter circuit of the 4th embodiment of the present invention.

Figure 14 is the circuit diagram of the low-pass filter circuit of the 5th embodiment of the present invention.

Figure 15 is the circuit diagram of the low-pass filter circuit of the 6th embodiment of the present invention.

Figure 16 is the circuit diagram of the low-pass filter circuit of the 7th embodiment of the present invention.

Figure 17 is the circuit diagram of the low-pass filter circuit of the 8th embodiment of the present invention.

Figure 18 is the circuit diagram of the low-pass filter circuit of the 9th embodiment of the present invention.

Figure 19 is the circuit diagram of the low-pass filter circuit of the of the present invention ten embodiment.

Figure 20 is the circuit diagram of the switched-capacitor circuit of parallel connection.

Figure 21 is the structure chart of the phase-locked loop of the 11 embodiment of the present invention.

Figure 22 is the structure chart of the delay lock loop back path of the 12 embodiment of the present invention.

Figure 23 is the structure chart of the phase-locked loop of the 13 embodiment of the present invention.

Figure 24 is the circuit diagram of a concrete starter in the phase-locked loop of Figure 23.

Figure 25 is the circuit diagram of a concrete starter of the corresponding test pattern of an energy.

Figure 26 is the structure chart of the reponse system of the 14 embodiment of the present invention.

Figure 27 is the structure chart of the phase-locked loop of the 15 embodiment of the present invention.

Figure 28 is for being applied to PLL of the present invention, DLL the example among the IC.

Figure 29 is for being applied to PLL of the present invention, DLL the example in the COC parts.

Figure 30 is for being installed to PLL of the present invention, DLL the example in LSI weld pad zone.

Figure 31 is for being installed to PLL of the present invention, DLL the example in the microprocessor.

Figure 32 is the structure chart of general PLL.

Figure 33 is the circuit diagram of general loop filter.

Figure 34 is the existing circuit diagram that can change the loop filter of filtering characteristic.

Figure 35 is existing another circuit diagram that can change the loop filter of filtering characteristic.

Embodiment

Below, with reference to the accompanying drawings, embodiments of the invention are described.

(first embodiment)

Fig. 2 shows the low-pass filter circuit of first embodiment of invention.Low-pass filter circuit 30A in the present embodiment is a secondary active filter.It comprises: capacity cell 312, current mirroring circuit 32A, capacity cell 33, operational amplifier 34A, current source 35a, 35b, offset compensator 36 and bias regulator 37.Need mention, low-pass filter circuit 30A can constitute semiconductor integrated circuit.

Low-pass filter circuit 30A can be applied among the PLL for example shown in Figure 32.In this case, since the charging current Ip (flow direction of electric current and shown in Figure 32 opposite) of self charge pump circuit 20 do the input of low-pass filter circuit 30A, be output as control voltage Vout, control voltage-controlled oscillator 40.

Current mirroring circuit 32A is equivalent to current generator of the present invention.Current mirroring circuit 32A has field-effect transistor 311A as first semiconductor element in input one side, has field-effect transistor 321A as second semiconductor element in output one side, is input with the first electric current I in, is output with the second electric current I out.Field-effect transistor 311A is according to showing that for the first bias current Ib1 that comes first coefficient of conductivity gm1 is as its electrical characteristic from current source 35a.Equally, field-effect transistor 321A, according to showing second coefficient of conductivity gm2 as its electrical characteristic from current source 35b for the second bias current Ib2 that comes, second coefficient of conductivity gm2 becomes certain ratio with first coefficient of conductivity gm1.In this embodiment, this certain ratio reflectance that promptly is current mirroring circuit 32A.

Capacity cell 312 and field-effect transistor 311A constitute the first filter 31A of the present invention together.Field-effect transistor 311A is equivalent to the circuit element that first filter is had that has of the present invention.The first filter 31A is that the input signal of low-pass filter circuit 30A is input with electric current I p, exports the first voltage Vp.

Capacity cell 33 is equivalent to second filter of the present invention.Capacity cell 33 is input with the second electric current I out from current mirroring circuit 32A output, exports second voltage.Under the situation of Fig. 2, second voltage is the voltage difference between voltage Vout and the voltage Vm.

Operational amplifier 34A is equivalent to adder of the present invention.Operational amplifier 34A is at the negative feedback capacity cell 33 that has part ownership, and as the input to in-phase input end, input tertiary voltage Vout is as the output signal of low-pass filter circuit 30A with the first voltage Vp.In other words, operational amplifier 34A constitutes active integrating circuit with capacity cell 33, imports the second electric current I out, and its integration is exported tertiary voltage Vout.

Describe offset compensator 36 and bias regulator 37 afterwards again in detail.

The working condition of the low-pass filter circuit 30A that presses above-mentioned formation then, is described.Need mention, the hypothesis of explanation does not have the situation of offset compensator 36 and bias regulator 37.

The electric current I p that supplies with low-pass filter circuit 30A has carried out Filtering Processing for the first time by the first filter 31A, exports the first voltage Vp.At this moment, the first electric current I in that flows through input one side of current mirroring circuit 32A is arrived output one side by mirrorization.Equate that in other words, reflectance (mirror ratio) is " 1 " if establish first coefficient of conductivity gm1 and second coefficient of conductivity gm2 here, then output one side of current mirroring circuit 32A just flows through the second anti-phase electric current I out that is equivalent to the first electric current I in.The second electric current I out has carried out Filtering Processing for the second time by capacity cell 33, exports second voltage.Because the inverting input of operational amplifier 34A and in-phase input end be because so-called imaginary short circuit and current potential is identical, so from operational amplifier 34A output has added second voltage among the first voltage Vp after and the tertiary voltage Vout that obtains.

Shared bias control signal CS1 is provided for current source 35a, 35b, first and second bias current Ib1, Ib2 change according to this bias control signal CS1.If first and second bias current Ib1, Ib2 change, first and second coefficient of conductivity gm1, the coefficient of conductivity gm2 of transistor 311A, 321A also change.In other words, without resistor ladder, low-pass filter circuit 30A just can momentarily change filtering characteristic under the effect of bias control signal CS1.

Then, show that low-pass filter circuit 30A has the transmission characteristic the same with general secondary active filter.

Fig. 3 is the figure in order to the transmission characteristic of explanation low-pass filter circuit 30A.Fig. 3 (a) shows the general secondary active filter shown in Figure 33 (b) again.Here, be all " R " in the resistance value of establishing resistive element 311A ', 321A ', the capacitance of capacity cell 312 is " Cx ", the capacitance of capacity cell 33 is " C ", when the magnification ratio of operational amplifier 34A was " A ", the modal equation formula (nodal equations) of some n and some m and voltage Vout were respectively suc as formula shown in (1)～formula (3).

$-\mathrm{Ip}+\mathrm{Vn}\·\mathrm{sCx}+\frac{\mathrm{Vn}-\mathrm{Vm}}{R}=0\·\·\·\left(1\right)$

$\frac{\mathrm{Vm}-\mathrm{Vn}}{R}+\frac{\mathrm{Vm}-\mathrm{<figure-callout\; id="1"\; label="Vout\; R\; +"\; filenames="C0380197100561.png,C0380197100571.png"\; state="\{\{state\}\}">Vout</figure-callout>}}{R+}=0\&CenterDot;\&CenterDot;\&CenterDot;\left(2\right)$

Vout＝A·(-Vm) … (3)

Here, suppose magnification ratio " A " infinity of operational amplifier 34A, obtain formula (4) and make transfer function Vout/Ip.

$\mathrm{Vout}/\mathrm{Ip}=-\frac{R\&CenterDot;\mathrm{sC}+1}{\mathrm{sC}(R\&CenterDot;\mathrm{sCx}+1)}\&CenterDot;\&CenterDot;\&CenterDot;\left(4\right)$

On the other hand, for low-pass filter circuit 30A, if the coefficient of conductivity of field- effect transistor 311A, 321A is all " gm ", the same the time, the modal equation formula of some n and some m and voltage Vout are respectively suc as formula shown in (5)～formula (7) with above-mentioned for the capacitance of capacity cell 312,33 and the magnification ratio of operational amplifier 34A.

Ip+Vp·sCx+Vp·gm＝0 … (5)

Vm·gm+(Vm-Vout)·sC＝0 … (6)

Vout＝A·(Vp-Vm) … (7)

Here,, suppose magnification ratio " A " infinity of operational amplifier 34, then obtain formula (8) and make transfer function Vout/Ip with above-mentioned the same.

$\mathrm{Vout}/\mathrm{Ip}=-\frac{\frac{\mathrm{sC}}{\mathrm{gm}}+1}{\mathrm{sC}(\frac{\mathrm{sCx}}{\mathrm{gm}}+1)}\&CenterDot;\&CenterDot;\&CenterDot;\left(8\right)$

If the gm=1/R in the formula (8), then formula (8) just becomes formula (4).In other words, as can be known: the transmission characteristic of low-pass filter circuit 30A is the same with the transmission characteristic of the general secondary active filter shown in Figure 33 (b).

Then, the minimizing of the circuit area of low-pass filter circuit 30A is described, the miniaturization of capacity cell 33 particularly is described.

Fig. 4 is the situation the during reflectance of the current mirroring circuit 32A among the low-pass filter circuit 30A for a change.As shown in the drawing, the coefficient of conductivity ratio of transistor 311A and transistor 321A is set at 1: α.The transistorized coefficient of conductivity can be adjusted by Change Example such as W/L (W: grid is wide, L: grid is long).And, with above-mentioned the same, the ratio of first bias current and second bias current also is set at 1: α.In other words, the reflectance with current mirroring circuit 32A is set at " α ".

As can be known: it is the same with original low-pass filter circuit 30A shown in Figure 2 to want to make reflectance with current mirroring circuit 32A to be set at the transmission characteristic of low-pass filter circuit 30A of " α ", and the capacitance with capacity cell 33 in formula (8) is set at " α C " and gets final product.Therefore, by " α " being set at positive number, just can make the capacitance of capacity cell 33 smaller less than 1.Need mention, allow under " α " smaller situation, the actual transmissions characteristic of low-pass filter circuit 30A has just departed from by the resulting theoretical value of formula (8), but can make " α " little by about to 1/10 to 1/100 at least.

Then, illustrate offset compensator 36 is how to compensate bias current.

Because used current mirroring circuit 32A, so there is following problem at low-pass filter circuit 30A.Promptly, can between the first electric current I in and the second electric current I out, produce error, promptly produce biasing (offset) owing to the characteristic deviation of current source 35a, 35b, the characteristic deviation of field- effect transistor 311A, 321A etc. in stable state.Therefore offset compensator 36 is set comes compensation offset current (offsetcurrent).Offset compensator 36, the voltage Vout that is produced during according to the first electric current I in that is input in blocking among the current mirroring circuit 32A changes the biasing of current source 35b, adjusts the second bias current Ib2 so that make voltage Vout become 0.

Fig. 5 is the concrete example of offset compensator.Offset compensator 36A contains: switch 361, voltage retainer 362 and signal inverter 363.Under the state of the blocking first electric current I in, by closing a switch 361, voltage Vout has just offered voltage retainer 362, keeps this voltage Vout by this voltage retainer 362.Voltage retainer 362 can be realized by for example sampling/holding circuit, perhaps only be realized by capacity cell.The voltage that voltage retainer 362 is kept has carried out voltage inversion by signal inverter 363, and exports as control voltage Vc1 with it.Signal inverter 363 can be realized by for example see-saw circuit.Control voltage Vc1 is fed the second bias current Ib2 that adjusts current source 35b to current source 35b.The feedback loop of Gou Chenging like this, drift current be 0 o'clock stable.Open switch 361 after feedback loop is stable, just can use low-pass filter circuit 30A.At this moment, the voltage when voltage retainer 362 is keeping feedback loop to stablize carries out the adjustment of the second bias current Ib2 according to this voltage.

But, the voltage that voltage retainer 362 is kept, the passing that is accompanied by the time is owing to leakage current of circuit etc. changes.Here, not that the voltage when feedback loop is stablized keeps as the analogue value, but as described belowly keep as digital value.

Fig. 6 shows another concrete example of offset compensator.Offset compensator 36B comprises: switch 361, comparator 364, increase/down counter 365 and D/A converter 366.Under the state of the blocking first electric current I in, by closing a switch 361, and Vout is exported to comparator 364.364 couples of voltage Vout of comparator are compared with the voltage (for example earthed voltage) that becomes benchmark, according to its big or small output supply voltage or earthed voltage.Increase/down counter 365 increases or reduces count value according to the output from comparator 364.D/A converter 366 will increase/and the count value of down counter 365 is converted to the control voltage Vc1 into the analogue value.Increase/down counter 365 and D/A converter 366 sample with shared control clock synchronization.Under this structure, the voltage when offset compensator 36B just stablizes with the form maintenance feedback loop of count value that is digital value.

As mentioned above, the shortcoming of offset compensator 36A is that the voltage that remains in the voltage retainer 362 can change, but can be made of the less circuit structure of scale.On the other hand, though offset compensator 36B circuit structure is bigger, compensation offset current accurately.And, by improving the bit accuracy of D/A converter 366, just can improve the precision of drift current compensation further.

Need mention, in the above description, Vc1 feeds back to current source 35b with control voltage, moreover, will control voltage Vc1 and feed back among the current source 35a and also be fine.Also can be by current source 35a, 35b control voltage Vc1.Any situation can both be received the effect the same with above-mentioned effect.

Duplicate circuit can be set come compensation offset current.Fig. 7 shows the low-pass filter circuit that is provided with the duplicate circuit that is used for compensating offset.Duplicate circuit 38 comprises: corresponding to the transistorized transistor 321A ' of transistor 321A, corresponding to the current source 35b ' of current source 35b, corresponding to the capacity cell 33 ' of capacity cell 33 and corresponding to the operational amplifier 34A ' of operational amplifier 34A.With the first voltage Vp is input, is output with the voltage Vout ' that is equivalent to tertiary voltage Vout.Offset compensator 36 will be anti-phase as the voltage Vout of input, as control voltage Vc1 output.The offset compensator 36 of this moment can be realized by the see-saw circuit with official hour constant.Control voltage Vc1 is fed to current source 35b ', and the bias current Ib2 that adjusts current source 35b ' serves as zero with the drift current of accomplishing duplicate circuit 38.Also offer current source 35b by controlling voltage Vc1, drift current that just can offset current mirror circuit 32A.So, if adopt the method used duplicate circuit 38, just can be in the relative accuracy scope drift current output corresponding to the element of these circuit elements in each circuit element among the low-pass filter circuit 30A and the duplicate circuit 38 be minimized.

As mentioned above, come the drift current that produced among the offset current mirror circuit 32A, thereby eliminate because the drift of the voltage Vout that drift current causes by offset compensator 36, duplicate circuit 38.Also just can improve the filtering accuracy of low-pass filter circuit 30A.

Secondly, illustrate bias regulator 37 is how to adjust biasing.

The first filter 31A among the low-pass filter circuit 30A uses transistor 311A to make the resistance circuit element.First coefficient of conductivity gm1 of transistor 311A is according to the first bias current Ib1 that comes is provided by current source 35a decision.But generally speaking, even the electric current of having setovered is certain, the shown coefficient of conductivity of transistor also changes along with variations in temperature.In other words, if variations in temperature, filtering characteristic also changes.For addressing this problem, bias regulator 37 is set in low-pass filter circuit 30A, and allows shown first and second coefficient of conductivity gm1 of transistor 311A, 321A and gm2 relative temperature change and necessarily constant.

Fig. 8 is the concrete example of bias voltage adjuster.Bias regulator 37A comprises: the current source 373 and the differential amplifier circuit 374 that also offer transistor 371a corresponding to the 3rd transistor 371a of transistor 311A, corresponding to the 4th transistor 371b of transistor 321A, corresponding to the current source 372a of current source 35a, corresponding to the current source 372b of current source 35b, with the bias current Iref that appends.Transistor 371a, 371b which be that diode connects.

Difference between current is that the bias current of Iref offers transistor 371a, the 371b among the bias regulator 37A respectively.So, just produced voltage difference delta V, i.e. the voltage difference of transistor 371a, 371b.Differential amplifier circuit 374 is input with voltage difference delta V, and output is controlled voltage Vc2 and allowed it become voltage Vref.Control voltage Vc2 is fed back to current source 372a, 372b, adjusts the bias current of each current source.Just the bias current of Control current source 372a, 372b, and accomplishing has been arranged after the feedback loop by above-mentioned formation: though variations in temperature, also difference between current Iref formation voltage difference Vref relatively, in other words, the coefficient of conductivity (Iref/Vref) is certain.Use control voltage Vc2, in other words, the adjustment situation of looking current source 372a, 372b is adjusted first and second bias current Ib1, the Ib2 of current source 35a, 35b, shown first and second coefficient of conductivity gm1 of transistor 311A, 321A and gm2 relative temperature is changed keep necessarily constant.

Generally, transistor has with the shown such tendency of coefficient of conductivity minimizing of the rising of temperature.When temperature rises, just the coefficient of conductivity can be kept necessarily constant by increasing bias current.Particularly, the serviceability temperature compensating circuit is made offset compensator and is got final product.Fig. 9 shows the temperature-compensation circuit as bias regulator.This temperature-compensation circuit 37B is the circuit that uses generally speaking, is a constant-current source circuit that certain electric current can be provided at supply voltage.People know: had after this temperature-compensation circuit 37B, output current just increases with absolute temperature with being directly proportional.

As mentioned above, can accomplish by bias regulator 37: the shown coefficient of conductivity relative temperature of transistor 311A, 321A changes and keeps necessarily constant.So, just no matter temperature how, can both make the frequency characteristic of low-pass filter circuit 30A necessarily constant.

As mentioned above, according to this embodiment,, just need not to install resistive element (resistive element 311A ', 321A ' in the low-pass filter circuit shown in Fig. 3 (a)), thereby can reduce circuit area by making active load with transistor 311A, 321A.And, by make 311A, 321A the shown coefficient of conductivity littler, just can make capacity cell 312,33 littler.

With regard to the reflectance " α " of current mirroring circuit 32A, set less than after 1 with " α ", just the capacitance of capacity cell 33 can be reduced to about 1/10～1/100.Generally speaking, be to use the large bulk capacitance element about 100～200pF to make capacity cell 33, for example its area ratio shared in PLL is about 50～70%.In the low-pass filter circuit 30A of this embodiment because capacity cell 33 capacitances are narrowed down to existing about 1/10～1/100, so can reduce circuit area significantly.By making the second electric current I out smaller, can reduce power consumption; The bias current that flows into operational amplifier 34A also diminishes, and the desired specification of operational amplifier 34A, for example percent of pass (through rate) etc. are relaxed.Need mention, because generally speaking, capacity cell 312 is the less capacitance (about 10～20pF) that the enough MOS of energy form, so there is no need to go to consider the miniaturization of this capacity cell 312 more.

Except capacity cell 312, capacity cell 33 can be made first filter and second filter, other capacity cell, resistive element etc. also can be made first filter and second filter.Also can use the parasitic capacitance of transistor 311A to make capacity cell 312.Even carry out such change, above-mentioned effect can not reduce yet.

In this embodiment, can come Control current source 35a, 35b by bias control signal CS1, but be not to do like this.And, can omit offset compensator 36, bias regulator 37.Can determine as required whether offset compensator 36, bias regulator 37 are set.

Yet,, just current mirroring circuit 32A can be replaced into resistive element by provide charging current and its negative-phase sequence curent simultaneously from charge pump circuit to low-pass filter circuit.Figure 10 shows the charge pump circuit 20 that can export charging current Ip1 and its negative-phase sequence curent Ip2 simultaneously and the low-pass filter circuit 30 that the current mirroring circuit 32A among the low-pass filter circuit 30A is replaced into resistive element 311A ', 321A '.In the charge pump circuit 20,24,25 groups on signal UP1, UP2 group and signal DN1, DN2 group 23,26 groups of control switchs of difference and switch.From charge pump circuit 20, export charging current Ip1 and negative-phase sequence curent Ip2 thereof simultaneously.On the other hand, by input charging current Ip1 and negative-phase sequence curent Ip2 thereof, the action that low-pass filter circuit 30 is carried out, the same with regard to the action of being carried out when in low-pass filter circuit 30A, importing the first electric current I in by current mirroring circuit 32A.

By PLL with such charge pump circuit 20 and low-pass filter circuit 30, homophase switching noise in the charge pump circuit 20 will be cancelled out each other by the in-phase input end and the inverting input that they are input to the operational amplifier 34A in the low-pass filter circuit 30.Like this, just, can reduce the shake composition that appears among the PLL.

(second embodiment)

Figure 11 shows the low-pass filter circuit of second embodiment of the present invention.Low-pass filter circuit 30B among this embodiment is the adder of the low-pass filter circuit 30A among first embodiment have been made some change back and formation.The place different with first embodiment below only is described.The inscape identical with first embodiment just do not done and illustrated that symbol is also with reference to the same-sign among the figure 2.

Among the low-pass filter circuit 30B, operational amplifier 331 at in-phase input end input reference voltage, is exported the second voltage V2 at the negative feedback capacity cell 33 that has part ownership.In other words, in this embodiment, capacity cell 33 and operational amplifier 331 are equivalent to second filter.

Adder 34B is equivalent to the adder among the embodiment.Adder 34B is input with the first voltage Vp and the second voltage V2, these voltages is added up export tertiary voltage Vout.

As mentioned above, according to this embodiment, the first voltage Vp and the second voltage V2 respectively can references.So, adder 34B is joined wherein as the part of the voltage-controlled oscillator among Figure 32 40, export first and second voltage respectively from low-pass filter circuit, can directly control voltage-controlled oscillator 40.

(the 3rd embodiment)

Figure 12 shows the low-pass filter circuit of the 3rd embodiment of the present invention.Low-pass filter circuit 30C among this embodiment is the adder of the low-pass filter circuit 30A among first embodiment have been made some change back and formation.The place different with first embodiment below only is described.The inscape identical with first embodiment just do not done and illustrated that symbol is also with reference to the same-sign among the figure 2.

Among the low-pass filter circuit 30C, an end of establishing capacity cell 33 is a reference voltage.So capacity cell 33 is input with the second electric current I out just, the second voltage V2 anti-phase with polarity is output.

Adder among this embodiment is operation transconductance amplifier (OTA) 34C.OTA34C as the input to in-phase input end, does input to inverting input with the anti-phase second voltage V2 of polarity with the first voltage Vp, exports the output signal that the 3rd electric current I out2 makes low-pass filter circuit 30C.The 3rd electric current I out2 is for being multiplied by the value that certain coefficient of conductivity obtains on the differential voltage between input terminal.In other words, in this embodiment, the voltage of the first voltage Vp and second voltage V2 total is transformed to electric current, exports as the 3rd electric current I out2.

As mentioned above, according to this embodiment,, just can output current signal make the output signal of low-pass filter circuit by making adder with operation transconductance amplifier 34C.And, because operational amplifier of no use is made adder,, can also reduce power consumption so can dwindle circuit scale.

(the 4th embodiment)

In the low-pass filter circuit 30A of first embodiment, transistor 311A, 321A are regarded as linear resistive element come usefulness, and say that closely transistor is non-linear element.Therefore, transistorized non-linear the non-linear of filtering characteristic that show as with chapter and verse.So, become linearity for making filtering characteristic, the low-pass filter circuit 30A among first embodiment is improved, the result after the improvement promptly is the low-pass filter circuit of the 4th embodiment of the present invention.

Figure 13 shows the low-pass filter circuit of this embodiment.The place different with first embodiment below only is described.The inscape identical with first embodiment just do not done and illustrated that symbol is also with reference to the same-sign among the figure 2.

The first filter 31D among the low-pass filter circuit 30D is made of capacity cell 312 and resistor ladder 311D.Resistor ladder 311D is equivalent to the circuit element that first filter of the present invention is had, and has shown the resistance value of first coefficient of conductivity gm1 that the transistor 311A that is equivalent among the current mirroring circuit 32A is shown.If the first shown coefficient of conductivity gm1 of transistor 311A changes, resistor ladder 311D also just changes resistance value along with this variation.And the bias current that will be equivalent to offer first bias current of transistor 311A by current source 35a offers resistor ladder 311D.

On the other hand, with current mirroring circuit 32A in transistor 311A arranged side by side be provided with capacity cell 312 ' corresponding to capacity cell 312.In other words, input one side of current mirroring circuit 32A and the first filter 31D equivalence.So, by the input electric current I p the same with the input of the first filter 31D, the voltage of input one side of current mirroring circuit 32A just becomes the voltage Vp ' that is equivalent to the first voltage Vp, and the electric current that flows into input one side becomes the 3rd electric current I in ' that is equivalent to the first electric current I in.Current mirroring circuit 32A is input with the 3rd electric current I in ', is output with the second electric current I out.Therefore, on showization of the current mirroring circuit 32A essence first electric current I among this embodiment produces the second electric current I out.

The in-phase input end of operational amplifier 34A has applied the i.e. output of the first filter 31D of the first voltage Vp.Export the tertiary voltage Vout that obtains after the first voltage Vp and the second voltage addition from operational amplifier 34A.

As mentioned above,, make resistive element among the first filter 31D, but make resistive element among the first filter 31D, so just can improve the linearity of the filtering characteristic of low-pass filter circuit 30D with resistor ladder 311D without transistor according to this embodiment; Change the resistance value of resistor ladder 311D by variation, just can dynamically change the filtering characteristic of low-pass filter circuit 30D according to the coefficient of conductivity of transistor 311A.

Need mention, not need dynamically to change under the situation of filtering characteristic, available simple resistive element replaces resistor ladder 311D.Do the linear such effect that also can receive the filtering characteristic that improves low-pass filter circuit 30D like this.

(the 5th embodiment)

Figure 14 shows the low-pass filter circuit of the 5th embodiment of the present invention.Low-pass filter circuit 30E among this embodiment is to constitute after being modified to the state that can switch by the filtering characteristic with the low-pass filter circuit 30A among first embodiment.The place different with first embodiment below only is described.The inscape identical with first embodiment just do not done and illustrated that symbol is also with reference to the same-sign among the figure 2.

Current mirroring circuit 32E among the low-pass filter circuit 30E has the first transistor 311A in input one side, has transistor 321Ab and these two transistors of the 3rd transistor 321Ac in output one side.Transistor 321Ab, 321Ac are setovered by current source 35a, 35b respectively.By allowing the first electric current I in input one side flow of current mirroring circuit 32E, the just output one side output current Ioutb from having transistor 321Ab is from the output one side output current Ioutc that has transistor 321Ac.

Contain among the low-pass filter circuit 30E: switch 322b, the 322c of switch current Ioutb, electric current I outc.The total that flows through the electric current of the switch 322b, the 322c that close offers capacity cell 33 as the second electric current I out.

As mentioned above, according to present embodiment,, just can switch to staged the filtering characteristic of low-pass filter circuit 30E, even if also can digitally switch the filtering characteristic of low-pass filter circuit 30E by the suitably open and close of control switch 322b, 322c.So, in the loop filter time spent of for example making PLL with low-pass filter circuit 30E, to the output clock locked till, increase the second electric current I out and accelerate to introduce, reduce the second electric current I out after the locking again and make bandwidth characteristic slow.

Need mention, in this embodiment, the transistor that is located at output one side of current mirroring circuit 32E is two, but the present invention is not limited to this.Also can have from the transistor of second to n (n is the natural number more than 3) in output one side of current mirroring circuit.

As for the switch that have or not of control from the output of a plurality of electric currents of current mirroring circuit 32E output, can from a plurality of electric currents, select any, also can select a plurality of simultaneously.

(the 6th embodiment)

Figure 15 is the low-pass filter circuit of the 6th embodiment of the present invention.Be to replace the current mirroring circuit 34A among the low-pass filter circuit 30A of first embodiment with the first voltage current transformating circuit 311F and the second voltage current transformating circuit 32F and constitute low-pass filter circuit 30F's among this embodiment.The place different with first embodiment below only is described.The inscape identical with first embodiment just do not done and illustrated that symbol is also with reference to the same-sign among the figure 2.

The first voltage current transformating circuit 311F is corresponding to transistor 311A among the low-pass filter circuit 30A of first embodiment and current source 35a.The first voltage current transformating circuit 311F shows first coefficient of conductivity gm1 the same with transistor 311A, and the first voltage Vp is transformed to the first electric current I in.Can change first coefficient of conductivity gm1 by bias control signal CS1.The first voltage current transformating circuit 311F is equivalent to the circuit element that first filter of the present invention is had, and constitutes the first filter 31F with capacity cell 312.

The second voltage current transformating circuit 32F is corresponding to transistor 321A among the low-pass filter circuit 30A of first embodiment and current source 35b.The second voltage current transformating circuit 32F is equivalent to current generator of the present invention, shows second coefficient of conductivity gm2 the same with transistor 321A, and the first voltage Vp is transformed to the second electric current I out.Can change second coefficient of conductivity gm2 by bias control signal CS1.

Press the transmission characteristic of the low-pass filter circuit 30F of above-mentioned formation, the same as the formula (8) with the low-pass filter circuit 30A of first embodiment.Working condition is also the same with low-pass filter circuit 30A.Therefore, if second coefficient of conductivity gm2 that the second voltage current transformating circuit 32F is shown sets to such an extent that shown first coefficient of conductivity gm1 is little than the first voltage current transformating circuit 311F, just can make capacity cell 33 miniaturizations, thereby can reduce circuit area significantly.

Voltage current transformating circuit 311F, 32F change first, second coefficient of conductivity gm1, gm2 respectively according to shared bias control signal CS1.So, just can dynamically change the filtering characteristic of low-pass filter circuit 30F.

As mentioned above, according to this embodiment,, just need not to install resistive element, thereby can reduce circuit area if make active load usefulness with the first voltage current transformating circuit 311F and the second voltage current transformating circuit 32F.And, by making the shown coefficient of conductivity of the first voltage current transformating circuit 311F, 32F littler, can make capacity cell 312,33 littler.If make the second shown coefficient of conductivity gm2 of the second voltage current transformating circuit 32F, first coefficient of conductivity gm1 more shown little than the first voltage current transformating circuit 311F, just can keep filtering characteristic certain in the limit, the limit is with capacity cell 33 miniaturizations.

The low-pass filter circuit 30F of this embodiment does not use current mirroring circuit to make current generator.Therefore, compare, be difficult to produce drift current, thereby can access the higher filtering characteristic of precision with using current mirroring circuit.

(the 7th embodiment)

Figure 16 shows the low-pass filter circuit of the 7th embodiment of the present invention.The low-pass filter circuit 30G of this embodiment comprises: the first low-pass filter circuit part 30a and the second low-pass filter circuit part 30b.With electric current I p ⁺I.e. first input signal and electric current I p ^-Promptly the differential wave of second input signal is input; With voltage Vout ⁺I.e. first output signal and voltage Vout ^-Promptly the differential wave of second output signal is output.

Among the low-pass filter circuit 30G, the structure of first and second low-pass filter circuit part 30a, the 30b all structure with the low-pass filter circuit 30A of first embodiment is the same.The first low-pass filter circuit part 30a is with electric current I p ⁺For input, with voltage Vout ^-Be output, and the second low-pass filter circuit part 30b is with electric current I p ^-For input, with voltage Vout ^-Be output.Need mention, first and second low-pass filter circuit part 30a, 30b structure and working condition separately all illustrated in first embodiment of the present invention, just no longer illustrated here.

As mentioned above, according to this embodiment,, just can improve the noise immunity of anti-homophase power supply noise if going into signal with the output of low-pass filter circuit is differential wave; If make first and second low-pass filter circuit part 30a, 30b with the low-pass filter circuit 30A among first embodiment, just can make first and second low-pass filter circuit part 30a, 30b capacity cell 33 miniaturizations separately, thereby reduce the circuit area of whole low-pass filter circuit 30G significantly.

Need mention, first and second low-pass filter circuit part 30a, 30b are not limited to the low-pass filter circuit 30A of first embodiment.All can make first and second low-pass filter circuit part 30a, 30b usefulness so long as have the device of identical filtering characteristic.Therefore, low-pass filter circuit 30A～30F of available first to the 6th embodiment makes any usefulness among first and second low-pass filter circuit part 30a, the 30b.And, if with low-pass filter circuit of the present invention work at least one of among first and second low-pass filter circuit part 30a, the 30b, just can receive the effect that the circuit area of low-pass filter circuit 30G reduces.

(the 8th embodiment)

Figure 17 shows the low-pass filter circuit of the 8th embodiment of the present invention.The low-pass filter circuit 30H of this embodiment also is with electric current I p ⁺I.e. first input signal and electric current I p ^-Promptly the differential wave of second input signal is input; Output voltage V out ⁺I.e. first output signal and voltage Vout ^-The differential wave of second output signal.Low-pass filter circuit 30H is the circuit with differentialization of low-pass filter circuit 30F of the 6th embodiment.

The first differential voltage current conversion circuit 311H is equivalent to the circuit element that first filter of the present invention is had.The first differential voltage current conversion circuit 311H, input voltage Vp ⁺With voltage Vp ^-Voltage difference make first voltage; Output current Iin ^-And electric current I in ^-Make first electric current.Electric current I in ⁺And electric current I in ^-Size decide by first coefficient of conductivity that the first differential voltage current conversion circuit 311H shows.The first differential voltage current conversion circuit 311H and capacity cell 312H constitute the first filter 31H together.

The second differential voltage current conversion circuit 32H is equivalent to current generator of the present invention.The second differential voltage current conversion circuit 32H, input voltage Vp ⁺With voltage Vp ^-Voltage difference make first voltage; Output current Iout ⁺And electric current I out ^-As second electric current.Electric current I out ⁺And electric current I out ^-Second coefficient of conductivity decision that shows by the second differential voltage current conversion circuit 32H of size.The first shown coefficient of conductivity of second coefficient of conductivity and the first differential voltage current conversion circuit 311H becomes certain ratio.

Capacity cell 33a, 33b are equivalent to second filter of the present invention.Capacity cell 33a, 33b be the electric current I out to be generated by the second differential voltage current conversion circuit 32H respectively ⁺And electric current I out ^-For input, export second voltage.At this moment, second voltage is voltage Vout ^-With voltage Vm ^-Difference and voltage Vout ⁺With voltage Vm ⁺Difference poor.

Differential operational amplifier 34H is equivalent to adder of the present invention.Differential operational amplifier 34H is at negative feedback have part ownership capacity cell 33a, 33b, with voltage Vp ⁺With voltage Vp ^-Voltage difference do input to in-phase input end, output voltage V out ⁺With voltage Vout ^-Voltage difference make the output signal of low-pass filter circuit 30H.

As mentioned above, according to present embodiment,, just can improve the noise immunity of anti-homophase power supply noise if going into signal with the output of low-pass filter circuit is differential wave.If make the second shown coefficient of conductivity of the second differential voltage current conversion circuit 32H less than the first shown coefficient of conductivity of the first differential voltage current conversion circuit 311H, just can be with capacity cell 33a, 33b miniaturization.Just can reduce the circuit area of low-pass filter circuit 30H thus significantly.

(the 9th embodiment)

In the low-pass filter circuit 30A of Fig. 2, used current mirroring circuit 32A to make to transmit the device of electric charge.But generally speaking, can produce by input one side in the current mirroring circuit and export the leakage current that the characteristics of transistor deviation etc. of a side is brought, in other words, drift current (offset current).Therefore, need in some cases offset compensator shown in Figure 2 36 to be set as mentioned above.

Also can use beyond the current mirroring circuit circuit for example switched-capacitor circuit make charge transfer device.And switched-capacitor circuit is different with current mirroring circuit, can not produce drift current.Below, consider with switched-capacitor circuit replacement current mirroring circuit and make the situation that charge transfer device constitutes low-pass filter circuit.

Figure 18 shows the low-pass filter circuit of the 9th embodiment of the present invention.The low-pass filter circuit 30I of this embodiment is to comprise: the secondary active filter of capacity cell 312, switched-capacitor circuit 311I and 32I, capacity cell 33 and operational amplifier 34A.Below, the place different with first embodiment only is described.The inscape identical with first embodiment just do not done and illustrated that symbol is also with reference to the same-sign among the figure 2.Need mention, low-pass filter circuit 30I can constitute semiconductor integrated circuit.

Switched-capacitor circuit 311I and 32I are that the circuit and the capacity cell 312 of non-sensitive type of parasitic capacitance or perhaps P.I. (Parasitic Insensitive) type constitutes the first filter 31I together.Switched-capacitor circuit 311I and 32I are the logic level of regulation for example when " H " at clock CK, are electrically connected (first connection status) with input one side of electric current I p.So, the first electric current I in just flows among switched-capacitor circuit 311I and the 32I according to the first voltage Vp.In other words, switched-capacitor circuit 311I and 32I are equivalent to the circuit element among the first filter 31I.On the other hand, when clock/CK is the anti-phase logic level for regulation of clock CK, for example when " H ", switched-capacitor circuit 311I is connected electrically on the reference voltage, and switched-capacitor circuit 32I is connected electrically on the capacity cell 33 (second connection status) simultaneously.So, capacity cell 33 receives the second electric current I out from switched-capacitor circuit 32 I.In other words, switched-capacitor circuit 32I is equivalent to current generator.

Here, the ratio with the total capacitance of switched-capacitor circuit 311I and 32I and the capacitance of switched-capacitor circuit 32I is set at 1: α (0＜α＜1).So the second electric current I out just becomes α times of the first electric current I in, can make the capacitance of capacity cell 33 smaller.As to the explanation that Fig. 4 did, " α " can be set in about 1/10 to 1/100 so little.

If the switching speed of switched-capacitor circuit 311I and 32I is out and away faster than the time constant of low-pass filter circuit 30I, the reception and registration characteristic of low-pass filter circuit 30I just with Fig. 4 in low-pass filter circuit 30A identical.Adorned low-pass filter circuit 30I in for example can using and made the input clock of PLL of loop filter as the switch of switched-capacitor circuit 311I and 32I clock CK.And because switching speed is fast more, the capacitance of capacity cell is more little in the switched-capacitor circuit, so preferably use the higher clock of output clock equifrequent of PLL.

As mentioned above, according to present embodiment, because do not produce drift current, so do not need offset compensator.Therefore, can make circuit scale also littler than the low-pass filter circuit 30A of Fig. 4.And switched-capacitor circuit does not generally comprise the resistance composition of the reason that becomes thermal noise etc.Therefore, compare with the situation of using current mirroring circuit, the noise resistance characteristic improves.

Need mention, in this embodiment, use the P.I. type to make switched-capacitor circuit 311I and 32I, moreover, also can use parasitic capacitance responsive type or perhaps P.S. (ParasiticSensitive) type.

Need not speak more, any one part among the first low-pass filter circuit part 30a among the low-pass filter circuit 30H of Figure 16 and the second low-pass filter circuit part 30b all can be used the low-pass filter circuit 30I of this embodiment.

(the tenth embodiment)

Figure 19 shows the low-pass filter circuit of the of the present invention ten embodiment.The low-pass filter circuit 30J of this embodiment obtains the low-pass filter circuit 30I of Figure 18 distortion back, is one to comprise the secondary active filter of capacity cell 312, P.S. type switched-capacitor circuit 311J and 32J, capacity cell 33 and adder 34J.Need mention, low-pass filter circuit 30J can constitute semiconductor integrated circuit.

Switched-capacitor circuit 311J and 32J and capacity cell 312 constitute the first filter 31J together.Because the working condition of switched-capacitor circuit 311J and 32J is the same with the working condition of switched-capacitor circuit 311I shown in Figure 180 and 32I,, explanation do not carry so omitting.

Adder 34J comprises voltage follower circuit 341.Particularly, in this embodiment, can use operational amplifier to constitute voltage follower circuit 341.

Voltage follower circuit 341 is with because the second electric current I out flows through the second voltage V2 that capacity cell 33 produced imports.And its output becomes and is connected on capacity cell 312 and the switched-capacitor circuit 311J reference voltage of an end separately.Therefore, the second voltage V2 that will produce at capacity cell 33 two ends from adder 34J output and at the add up voltage Vout of gained of first voltage that the first filter 31J produces.

As for low-pass filter circuit 30J, also the low-pass filter circuit 30I with Figure 18 is the same, if the ratio of the total capacitance of switched-capacitor circuit 311J and 32J and the capacitance of switched-capacitor circuit 32J is set at 1: α (0＜α＜1) just can make the capacitance of capacity cell 33 smaller.And, adorned input clock that low-pass filter circuit 30J makes the PLL of loop filter, output clock clock CK in available as the open and close of control switch condenser network 311J and 32J.

As mentioned above,, can receive and the 9th the same effect of embodiment the effect that can obtain dwindling circuit scale, improves the noise resistance characteristic according to present embodiment.

Yet, in the low-pass filter circuit 30J of the low-pass filter circuit 30I of Figure 18 and Figure 19, can 2 switched-capacitor circuits in parallel, be new switched-capacitor circuit.Figure 20 shows switched-capacitor circuit in parallel.Among Figure 20 (a), be the circuit that the P.S. type is together in parallel; Among Figure 20 (b), be the circuit that the P.I. type is together in parallel.Any pattern all is such, and when a side of parallel connection was connected electrically in terminal T1, the opposing party then was connected electrically on the terminal T2.Compare with independent switched-capacitor circuit, the switched-capacitor circuit that is together in parallel like this, sampling rate is fast 1 times, can not produce because the electric capacity change that switch causes.

Need mention, in low-pass filter circuit 30I shown in Figure 180, can use the P.I. type switched-capacitor circuit that is together in parallel and the P.S. type switched-capacitor circuit that is together in parallel in any circuit.

Need not speak more, any one among the first low-pass filter circuit part 30a among the low-pass filter circuit 30H of Figure 16 and the second low-pass filter circuit part 30b can be used the low-pass filter circuit 30J of this embodiment.

(the 11 embodiment)

Figure 21 shows the structure of the PLL of the 11 embodiment of the present invention.The PLL of this embodiment comprises: phase comparator 10, charge pump circuit 20, loop filter 30A, voltage-controlled oscillator 40, voltage current transformating circuit 41, biasing translation circuit 42 and Fractional-N frequency device 50.The PLL of this embodiment can be used as multiple (multiple) PLL, extract PLL, distortion (skew) out synchronously adjusts usefulness such as PLL.Under some application target, can save frequency divider 50 need not.And the PLL of this embodiment can constitute semiconductor integrated circuit.

The low-pass filter circuit 30A of available first embodiment makes the loop filter 30A among the PLL of this embodiment.But the offset compensator among the low-pass filter circuit 30A 36 and bias regulator 37 saved be out of use.In Figure 21, each circuit element among the loop filter 30A is represented with the symbol the same with Fig. 2, the structure of loop filter 30A and working condition have not been described yet.Loop filter 30A, the charging current Ip that is exported with charge pump circuit 20 is input, output voltage V out makes output signal.The capacity cell 33 miniaturization this point of loop filter 30A can be narrated in front.

Charge pump circuit 20 is according to producing charging current Ip by phase comparator 10 phase difference relatively, input clock CKin and feedback clock CKdiv.The current source 21,22 of charge pump circuit 20 is changed the size of charging current Ip respectively by bias control signal CS1, CS2 control.

Voltage-controlled oscillator 40 is equivalent to output clock generator of the present invention.The oscillator of voltage-controlled oscillator 40 for allowing output clock CKout vibrate changes frequency of oscillation according to the voltage Vout from loop filter 30A output.In fact voltage-controlled oscillator 40 is not directly to be controlled by voltage Vout, but imports with the bias control signal CS1 that is transformed to electric current by voltage current transformating circuit 41, allows output clock CKout change.

Voltage current transformating circuit 41 is equivalent to offset controller of the present invention.Voltage current transformating circuit 41 will be transformed to bias control signal CS1 from the voltage Vout of loop filter 30A output.Bias control signal CS1 also controls current source 21 in the charge pump circuit 20 and current source 35a, the 35b among the loop filter 30A except controlling above-mentioned voltage-controlled oscillator 40.And bias control signal CS1 is transformed to by biasing translation circuit 42 after the bias control signal CS2, the current source 22 in the control charge pump circuit 20.Be because the polarity of the biasing of current source 22 control and other the different biasing translation circuits 42 that just are provided with.Bias control signal CS1 is the bandwidth characteristic of the operational amplifier 34A among the control loop filter 30A.

, accomplish during at control above-mentioned bias control signal CS1, CS2: the same from the intensity of variation of the size of the electric current of these current sources outputs to current source 21,22,35a, 35b.In other words, become A doubly the time, also become A doubly from the size of first and second bias current Ib1, the Ib2 of current source 35a, 35b output from the size of the charging current Ip of current source 21,22 output.

Then, describe working condition in detail, particularly describe the method for adjustment of damping coefficient in detail by the PLL of above-mentioned formation.Should indicate, if above-mentioned loop filter 30A is used among the PLL, the ring progression of PLL (loop order) just becomes 3 grades, but because being difficult to resolve transfer function is the PLL of tertiary circulation (third-order loop), so explanation here is similar to the situation of two-stage ring (second-orderloop).

The response characteristic of PLL that secondary active type loop filter is housed is by the natural frequency ω of following formula (9) _nAnd the damping coefficient ξ of formula (10) decides.Here, Ko is the gain of voltage-controlled oscillator 40.

$\mathrm{\&omega;n}=\sqrt{\frac{\mathrm{Ko}\&CenterDot;\mathrm{Ip}}{2\mathrm{\&pi;C}}}\&CenterDot;\&CenterDot;\&CenterDot;\left(9\right)$

$\mathrm{\&zeta;}=\frac{\mathrm{CR}}{2}\&CenterDot;\sqrt{\frac{\mathrm{Ko}\&CenterDot;\mathrm{Ip}}{2\mathrm{\&pi;C}}}=\frac{\mathrm{CR}}{2}\&CenterDot;\mathrm{\&omega;n}\&CenterDot;\&CenterDot;\&CenterDot;\left(10\right)$

Natural frequency ω in the decision formula (9) _nVariable in be easy to from circuit structure change for charging current Ip.When the frequency of oscillation according to PLL changes the wide that is natural frequency ω of endless belt _nThe time, generally be to change charging current Ip.

If change charging current Ip, the damping coefficient ξ in the formula (10) also changes simultaneously.But also preferably damping coefficient ξ is necessarily constant for the response characteristic of stablizing PLL.Also have, in formula (10), must guarantee: the rate of change A of the relative charging current Ip of rate of change of capacitance " C " or resistance value " R " is

Here, the drain and gate of transistor 311A links to each other, and that is to say, is in the state that shows square-law (square-law) characteristic.And, because the grid of transistor 321A is identical with the current potential of the grid of transistor 311A, so transistor 321A also is in the state that shows square-law characteristic.

Current source 21,22 in charge pump circuit 20 allows under the situation that charging current Ip changes with rate of change A by executing the bias control signal CS1, the CS2 that come, and current source 35a, the 35b among the loop filter 30A also allows first and second bias current Ib1, Ib2 change with rate of change A by executing the bias control signal CS1 that comes.The result is, for transistor 311A, drain current becomes A doubly, and first coefficient of conductivity gm1 becomes Doubly, grid voltage Vp is Doubly.In other words, first filter time constant changes along with offset change.Equally, for transistor 321A, drain current becomes A doubly, and second coefficient of conductivity gm2 becomes Doubly, grid voltage Vm becomes Doubly.

Coefficient of conductivity gm1 and the gm2 of transistor 311A, 321A become

Doubly, the rate of change with resistance value R becomes

Equate.Therefore, in formula (10), the rate of change A of charging current Ip and the rate of change of resistance value R Cancel out each other, damping coefficient ξ is necessarily constant.

In the PLL of this embodiment, the frequency of oscillation of voltage-controlled oscillator 40 changes according to the bias control signal CS1 of output from voltage current transformating circuit 41, corresponding with this variation, from the charging current Ip of the current source in the charge pump circuit 20 21,22, also change from first and second bias current Ib1, the Ib2 of current source 35a, 35b among the loop filter 30A and the bandwidth characteristic of the operational amplifier 34A among the loop filter 30A.In other words, can allow offset change (the adaptive biasingization: adaptive bias method) of charge pump circuit 20 and loop filter 30A according to the variation of the frequency of oscillation of PLL.Particularly, under the lower situation of the frequency of oscillation of voltage-controlled oscillator 40, first and second bias current Ib1, the Ib2 of charging current Ip and loop filter 30A diminish; And under the frequency of oscillation condition with higher of voltage-controlled oscillator 40, it is big that first and second bias current Ib1, the Ib2 of charging current Ip and loop filter 30A become.

The PLL that above-mentioned adaptive biasing has been changed has been (reference literature 1:JohnG.Maneatis known to everybody, " Low-Jitter Process-Independent DLL and PLL Based onSelf-Bised Techniques ", IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL.31, No.11, NOVEMBER 1996, pp.1723-1732).Yet cited circuit mode only is the adaptive biasingization to the PLL that has 2 grades of responses in the document 1; And among the PLL in this embodiment, but be adaptiveization of PLL of 4 grades of responses of control of seeking in secondary loop filter 30A, to add three grades of responses of voltage-controlled oscillator 40 and adding the bandwidth characteristic of operational amplifier 34A again.

As mentioned above, according to this embodiment, can realize the miniaturization of the capacity cell 33 among the loop filter 30A, thereby can reduce the circuit area of whole PLL significantly; By first and second bias current Ib1 among the control loop filter 30A, the size of Ib2, just the damping coefficient ξ of PLL can be adjusted into necessarily constant, so, just no matter the frequency of oscillation of PLL how, but can remain best response characteristic; Also there is no need to install in order to the resistor ladder of adjusting damping coefficient ξ etc., just can reduce the circuit area of PLL.

First and second bias current Ib1, the Ib2 of charging current Ip, loop filter 30A and the bandwidth characteristic of operational amplifier 34A suitably change according to the frequency of oscillation of PLL.So, just can in wider surge frequency range, keep the response characteristic of PLL best.Can save the band-gap reference circuit (band gapreference) that reference voltage is offered current source 21,22,35a, 35b, thereby the circuit area of PLL is further reduced.

Should be pointed out that first and second bias current Ib1, the Ib2 of charging current Ip and loop filter 30A, is not must be based on the bias control signal CS1 control from voltage current transformating circuit 41 outputs.And the bandwidth characteristic of not controlling operational amplifier 133A also is fine.So, under situation not, also can reduce circuit area with the adaptive biasingization of PLL.

In this embodiment, use the low-pass filter circuit 30A among first embodiment to make loop filter 30A, but the present invention is not limited to this.For example, can use low-pass filter circuit 30B～30F among second to the 6th embodiment and low-pass filter circuit 30I and the 30J among the 9th and the tenth embodiment, also can use the low-pass filter circuit of structure in addition.

(the 12 embodiment)

Figure 22 shows the structure of the delay locked loop (DLL) of the 12 embodiment of the present invention.The DLL that obtain of this embodiment to liking the part displacement of the PLL among the 11 embodiment is back.Below only explanation and the 11 place that embodiment is different.Do not illustrated that with regard to not doing symbol is also with reference to the same-sign among Figure 21 with the 11 the identical inscape of embodiment.

Use the low-pass filter circuit 30F of the 6th embodiment to make loop filter 30F among the DLL of this embodiment.Among Figure 21, each circuit element of loop filter 30F is represented with the symbol identical with Figure 15, omits the explanation to structure and the working condition of loop filter 30F.Loop filter 30F is input with the charging current Ip from charge pump circuit 20 outputs, and output voltage V out makes output signal.This situation of miniaturization of capacity cell among the loop filter 30F 33 can be illustrated.

With respect to the rate of change A of the charging current Ip that is controlled by bias control signal CS1, shown first and second coefficient of conductivity gm1 and the gm2 of voltage current transformating circuit 311F, the 32F of loop filter 30F becomes

So, as the explanation of being done among the 11 embodiment, can make the variation of the relative charging current Ip of damping coefficient ξ but necessarily constant by adjustment.

Voltage control delay circuit 40A is equivalent to output clock generator of the present invention.Voltage control delay circuit 40A produces and will postpone the output clock CKout that the back obtains as the input clock CKin of input according to the voltage Vout from loop filter 30F output.In fact, voltage control delay circuit 40A is not directly by voltage Vout control, but produces the output clock CKout of the retardation with a biasing of being executed corresponding to the bias control signal CS1 that is transformed to electric current by voltage current transformating circuit 41.

As mentioned above,, can make capacity cell 33 miniaturizations of loop filter 30F, thereby reduce the circuit area of whole DLL significantly according to this embodiment.Also have, need not resistor ladder etc. be installed, just the damping coefficient ξ of DLL can be adjusted into necessarily by the shown coefficient of conductivity of voltage current transformating circuit 311F, 32F of control loop filter 30F.

Need mention, in this embodiment, be to use the 6th low-pass filter circuit 30F among the embodiment to make loop filter 30F, but the present invention be not limited to this.For example, can use low-pass filter circuit 30A～30E among first to the 5th embodiment and low-pass filter circuit 30I and the 30J among the 9th and the tenth embodiment, also can use the low-pass filter circuit of structure in addition.

(the 13 embodiment)

As the 11 embodiment, under situation with the adaptive biasingization of PLL, output current from voltage current transformating circuit 41 becomes at 0 o'clock, first and second bias current Ib1, Ib2 by bias control signal CS1 control charging current Ip, loop filter 30A also become 0, and PLL is under static state stable.Therefore, under the method that locks improving frequency of oscillation inchmeal, there is when starting system stable under 0 state, PLL problem such as be failure to actuate.At this moment the frequency of oscillation that the PLL of adaptive biasingization is taked to make it lowers and the method that locks inchmeal from maximum rating.So starter (startup means) is set in PLL.

Figure 23 shows the structure of the PLL of the 13 embodiment of the present invention.The PLL of this embodiment is to be provided with starter 60 and to constitute in the PLL of the 11 embodiment.Below, describe starter 60 in detail.

Starter 60 is switched voltage Vout between first state and second state.Under first state, be that the output signal of loop filter 30 is set at the output from the adder of loop filter 30 with voltage Vout; Under second state, be the starting voltage that the output signal of loop filter 30 is set at regulation with voltage Vout.Be set at second state during starting PLL, can be from the state-driven PLL of frequency of oscillation maximum.

During from the status triggering PLL of the frequency of oscillation maximum of voltage-controlled oscillator 40, if frequency rises excessively, Fractional-N frequency device 50 will misoperation, and the frequency of feedback clock Ckdiv will become 0.If such situation takes place, phase comparator 10 is absorbed in so-called dead-lock (dead lock) state for improving the frequency of oscillation control system.So the back level at current-voltage conversion circuit 41 is provided with restricting circuits 43, the biasing that adds to voltage-controlled oscillator 40 is applied restriction, so that frequency of oscillation can not rise excessively.

Figure 24 is the concrete example of starter.Starter 60A comprises: switching has, switch 61 that non-transformer is supplied with operational amplifier 34A, switches the short circuit between the input terminal of the switch 62 that whether output of loop filter 30 is set at voltage Vref2, switch operation amplifier 34A and the switch 63 of disconnection.Need mention, capacity cell 33 and operational amplifier 34A are second filter and the adder in the loop filter 30.

When initiating signal S_UP was first logic level (for example " H "), switch 61 was opened, simultaneously switch 62,63 closures.Because switch 61 is opened, and is cut off so offer the power supply of operational amplifier 34A, has not had the output from operational amplifier 34A.If close a switch 62, then the output voltage V out of loop filter 30 is set at the starting voltage Vref2 last (second state) of regulation.By closing a switch 63, the voltage of in-phase input end one side of operational amplifier 34A just imposes on capacity cell 33.This voltage is exactly first voltage of the present invention.In other words, initiating signal S_UP becomes after first logic level, and operational amplifier 34A just becomes non-action status, just starts PLL under the output of loop filter 30 is set at second state of voltage Vref2.Therefore, system is stable under 0 state in the time of can avoiding starting.

After the starting PLL, allow the initiating signal S_UP be second logic level (for example " L ").So, switch 61 closures, switch 62,63 is opened simultaneously.Because switch 62,63 is opened, the input terminal of operational amplifier 34A is separated, and, separate voltage Vref from output one side of loop filter 30.Meanwhile, shut after the switch 61, operational amplifier 34A just becomes operate condition, and the output of loop filter 30 is set to from the output of operational amplifier 34A (first state).So PLL just can work under stable state.

Operational amplifier is the amplifier that the potential difference between inverting input and the in-phase input end is amplified.Therefore, if when the operating state that operational amplifier 34A is switched to from the non operating state under second state under first state, produce potential difference between input terminal, then moment can produce excessive output, causes the disorder of system.Yet, in this embodiment, under second state, by switch 63 short circuits, become the state of no-voltage difference between the input terminal of operational amplifier 34A, so when switching to first state, do not have excessive output from operational amplifier 34A.So, the effect that switch 63 is had is exactly: not only under first state first voltage is offered capacity cell 33, and when switching to second state, also can make operational amplifier 34A not have excessive output.

Yet, generally speaking, be provided with the test pattern of voltage-frequency of oscillation characteristic among the PLL in order to measuring voltage control generator 40.Here, how to improve starter 60 and make it can corresponding test pattern with regard to considering.

Figure 25 is the concrete example of starter that can corresponding test pattern.Starter 60B appends switch 64,65 and obtains on the basis of starter 60A.Switch 65 is made the controlling object of initiating signal S_UP according to test signal TEST diverter switch 62 and switch 64.Whether switch 64 is set at the output of loop filter 30 between the voltage Vref3 switches.Need mention, voltage Vref3 is for coming from the outside.

If test signal TEST is set at first logic level (for example " H "), switch 65 just selector switch 64 is made the controlling object of initiating signal S_UP.So PLL becomes test pattern, the output of loop filter 30 can be set at voltage Vref3 and be external power source and start PLL.And, by being set at various voltages, voltage Vref3 starts PLL, just energy measurement voltage-frequency of oscillation characteristic.

On the other hand, if test signal TEST is set at second logic level (for example " L "), test pattern is just removed.So, just the output of loop filter 30 can be set at voltage Vref2 and be internal electric source and start PLL.

As mentioned above, according to this embodiment, the PLL that has changed for adaptive biasing is provided with after the starter 60, is stabilisation under 0 the state in system in the time of just avoiding starting, and PLL this situation of can not working takes place.And, can corresponding test pattern by making starter 60, the voltage of also easy measuring voltage control generator 40-frequency of oscillation characteristic.

Need mention, in this embodiment,, cut off the power supply supply by switch 61, but the present invention be not limited to this although stopped the output of operational amplifier 34A.For example, under first state, allow output one side of operational amplifier 34A become high impedance, also can stop the output of operational amplifier 34A in fact.Do like this, above-mentioned effect does not have any variation yet.

(the 14 embodiment)

Figure 26 shows the structure of the reponse system of the 14 embodiment of the present invention.Figure 26 (a) shows PL1, and Figure 26 (b) shows DLL.In the reponse system among this embodiment, using with the differential wave is that the device that output is gone into is made loop filter 30.And, use the circuit of doing to import with differential wave to make voltage-controlled oscillator 40 and voltage control delay circuit 40A.

For example can use the 7th among the embodiment low-pass filter circuit 30G or the low-pass filter circuit 30H among the 8th embodiment make this embodiment reponse system in loop filter 30 usefulness.So, can reduce the circuit area of whole reponse system significantly.

(the 15 embodiment)

Can produce this phenomenon of drift current when in loop filter, using current mirroring circuit and do explanation.If produce drift current in the loop filter,,, provide charging current from charge pump circuit to offset this drift current for the PLL of reponse system just starts working.So, will in the output clock of PLL, produce the stable phase angle error.Here just consideration formation one can be eliminated this part thing of PLL of this stable phase angle error.

Figure 27 shows the structure of the PLL of the 15 embodiment of the present invention.Among the PLL of this embodiment, comprising: make loop filter 30A with the low-pass filter circuit among first embodiment etc., also comprise stable phase angle error concealment circuit 70 with current mirroring circuit.Below only illustrate and the 11 place that embodiment is different, just do not do not illustrated that symbol is also with reference to the same-sign among Figure 21 with the 11 the identical inscape of embodiment.

Stable phase angle error concealment circuit 70 comprises: according to produce electric current I 3 that is the charge pump circuit 71 that produces electric current I 3 according to the signal UP and the DN of phase comparator 10 outputs by phase comparator 10 phase difference relatively, input clock CKin and feedback clock CKdiv, as the capacity cell 72 of the electric charge integrator of received current I3 and the Voltage-controlled Current Source 73 that produces electric current I 4 according to the voltage that in capacity cell 72, is produced.

The working condition of stable phase angle error concealment circuit 70 is as described below.At first, charge pump circuit 71 will be transformed to electric current I 3 as signal UP and the DN that the stable phase angle error occurs.Electric current I 3 produces electric current I 4 by capacity cell 72 integrations according to this integral voltage.Be fed back to output one side of the current mirroring circuit among the loop filter 30A by the electric current I 4 of Voltage-controlled Current Source 73 generations.As a result, the drift current that is produced in the loop filter is compensated.

More than, according to this embodiment, be provided with to loop filter among the PLL of current mirroring circuit, can automatically eliminate the stable phase angle error.

Need mention, in the above description, will feed back to output one side of the current mirroring circuit among the loop filter 30A, moreover, also it can be fed back to input one side by the electric current I 4 that Voltage-controlled Current Source 73 produces.Only, in this case, be necessary to make the flow direction of the electric current I 3 that charge pump circuit 71 produces opposite.

Can also constitute the DLL that comprises stable phase angle error concealment circuit 70.

(application examples of reponse system involved in the present invention)

Because PLL of the present invention and DLL do not need large-scale capacity cell, and circuit scale is changed on a small scale, it can be applied in the following product so especially wait in expectation.

Figure 28 shows PLL of the present invention and DLL is applied to IC-card with the example on the LSI.Be restricted on erection space owing to be used for the LSI of IC-card, therefore can constitute PLL of the present invention and DLL with littler circuit area and be fit to very much do IC-card usefulness.

Figure 29 is for being applied in PLL of the present invention and DLL the example on COC (the chip on chip) parts.In the COC structure, the circuit area of the semiconductor integrated circuit on upper strata is restricted.Therefore, PLL of the present invention and DLL are very effective.

Figure 30 is for being installed in PLL of the present invention and DLL the example of LSI gasket part.The same with the COC structure, the circuit area that can be installed in the LSI gasket part is limited.Therefore, PLL of the present invention and DLL are very effective.

The example of Figure 31 for PLL of the present invention and DLL are installed as the clock generator in the microprocessor.Now a lot of PLL and DLL are installed in the microprocessor.Also wish very much in microprocessor, to use PLL of the present invention and DLL, so that reduce the circuit area of whole microprocessor significantly.Therefore, it is very big PLL of the present invention and DLL to be used on the microprocessor the obtainable effect of institute.

Various embodiment of the present invention more than has been described.Need mention, in the above description, transistor 311A, the 321A among the current mirroring circuit 32A can be N channel-type or P channel-type.Though constitute with field-effect transistor (MOS transistor) here, also be fine with the bipolar transistor formation; MOS transistor and bipolar transistor combined to use also be fine.Import a side and also can be diode.Do such change, the effect that the present invention produced can not be affected yet.

Capacity cell 33,33a, 33b in capacity cell 312 in first filter, 312 ', 312H, second filter can be: use 2 layers of polysilicon for example, MIM electric capacity (metal-insulator-metal electric capacity) and one of having used in the mos capacitance etc. of MOS transistor.In use that they are combined, the effect that the present invention produced can not be affected yet.

In sum, according to the present invention, the same under the filtering characteristic of low-pass filter circuit and the prior art, but its circuit area has reduced significantly.Particularly, the capacity cell that uses as second filter can be narrowed down to about 1/10～1/100 of existing capacity cell, the possibility that reduces circuit area is very big.By using current generator to generate second electric current also littler, just can reduce power consumption than first electric current.

Resistor ladder etc. need not be installed, have the damping coefficient that above-mentioned low-pass filter circuit is made the reponse system of the phase-locked loop of loop filter, delay-locked loop circuit etc. with regard to adjusting.So, can further reduce the circuit area of reponse system.And, can suitably adjust the response characteristic of reponse system according to the output of loop filter.So, can in the frequency band of broad, keep response characteristic best.

Claims (44)

1. low-pass filter circuit is characterized in that:

Comprise:

Input signal with this low-pass filter circuit is input, is first filter of output with first voltage;

The circuit element that is had for described first filter, the circuit element that allows first electric current flow according to described first voltage;

Generate the current generator that becomes second electric current of certain ratio with described first electric current;

With described second electric current is input, is second filter of output with second voltage; And

With described first voltage and the described second voltage addition, export the adder of the output signal of this low-pass filter circuit.

2. low-pass filter circuit according to claim 1, wherein:

Described certain ratio is the positive number less than 1.

3. low-pass filter circuit according to claim 1, wherein:

Described current generator, be: have first semiconductor element that shows first coefficient of conductivity in input one side, have demonstration becomes second coefficient of conductivity of described certain ratio with described first coefficient of conductivity second semiconductor element in output one side, with described first electric current is input, with described second electric current current mirroring circuit that is output;

Described circuit element is described first semiconductor element.

4. low-pass filter circuit according to claim 3, wherein:

Described circuit element for replacing described first semiconductor element, shows the resistive element of the resistance value that is equivalent to described first coefficient of conductivity;

Described current mirroring circuit replaces described first electric current, is input with the 3rd electric current that is equivalent to described first electric current.

5. low-pass filter circuit according to claim 1, wherein:

Described circuit element, for: show first coefficient of conductivity, be first voltage current transformating circuit of described first electric current with described first voltage transformation;

Described current generator, for: show become second coefficient of conductivity of described certain ratio with described first coefficient of conductivity, be second voltage current transformating circuit of described second electric current with described first voltage transformation.

6. low-pass filter circuit according to claim 1, wherein:

Described adder, for: negative feedback have part ownership described second filter, with described first voltage for the input of in-phase input end, output tertiary voltage being made the operational amplifier of described output signal.

7. low-pass filter circuit according to claim 1, wherein:

Described adder, for: with described first and second voltage is that input, output the 3rd electric current are as the operation transconductance amplifier of described output signal.

8. low-pass filter circuit according to claim 3, wherein:

Described first semiconductor element, for: according to the first transistor that shows described first coefficient of conductivity for first bias current that comes;

Described second semiconductor element, for: according to the transistor seconds that shows described second coefficient of conductivity for second bias current that comes;

Described first and second bias current, its size changes according to shared bias control signal.

9. low-pass filter circuit according to claim 3, wherein:

Described current mirroring circuit, for: have from the semiconductor element of described second to n (n is the natural number more than 3) in output one side, have the switch that has or not that switches the output of the electric current that is flowing in described second to n semiconductor element respectively simultaneously;

Described switch is exported any or any several aggregate value that flow in the electric current of described second to n semiconductor element as described second electric current.

10. low-pass filter circuit according to claim 4, wherein:

Described first semiconductor element, for: according to the first transistor that shows described first coefficient of conductivity for first bias current that comes;

Described second semiconductor element, for: according to the transistor seconds that shows described second coefficient of conductivity for second bias current that comes;

Described first and second bias current, its size changes according to shared bias control signal;

Described circuit element, for: replace described resistive element, show the resistance value that is equivalent to described first coefficient of conductivity, the resistor ladder that can change described resistance value according to the variation of described first coefficient of conductivity.

11. low-pass filter circuit according to claim 5, wherein:

Described first, second current conversion circuit is changed described first, second coefficient of conductivity respectively according to shared bias control signal.

12. low-pass filter circuit according to claim 3, wherein:

At least one semiconductor element in described first and second semiconductor element, for: according to the transistor that shows at least one coefficient of conductivity in described first and second coefficient of conductivity for the bias current that comes;

This low-pass filter circuit comprises: the described output signal during according to described first failure of current is adjusted the offset compensator of described bias current.

13. low-pass filter circuit according to claim 12, wherein:

Described offset compensator has: the voltage retainer of the voltage of the described output signal when keeping described first failure of current;

Described offset compensator, the voltage that is kept based on described voltage retainer carries out described adjustment.

14. low-pass filter circuit according to claim 12, wherein:

Described offset compensator comprises:

The voltage of the described output signal during more described first failure of current and become the comparator of size of the voltage of benchmark,

Based on increasing/down counter that the output from described comparator allows count value increase or reduces, and

Described count value is transformed to the D/A converter of the analogue value;

Carry out described adjustment based on output from described D/A converter.

15. low-pass filter circuit according to claim 3, wherein:

Described second semiconductor element, for: according to the transistor that shows described second coefficient of conductivity for the bias current that comes;

This low-pass filter circuit comprises:

Duplicate circuit, its structure is the same with the part that is made of described second semiconductor element, described second filter and described adder, and

Offset compensator, it is based on the output from described duplicate circuit, in described duplicate circuit, adjust for the transistorized bias current of described second semiconductor element, simultaneously the described bias current of supplying with described second semiconductor element is adjusted supplying with.

16. low-pass filter circuit according to claim 15, wherein:

Described offset compensator is the see-saw circuit with regular hour constant.

17. low-pass filter circuit according to claim 3, wherein:

Described first semiconductor element, for: according to the first transistor that shows described first coefficient of conductivity for first bias current that comes;

Described second semiconductor element, for: according to the transistor seconds that shows described second coefficient of conductivity for second bias current that comes;

This low-pass filter circuit comprises: according to variations in temperature, and the bias regulator that described first and second bias current is adjusted.

18. low-pass filter circuit according to claim 17, wherein:

Described bias regulator comprises: corresponding to the 3rd transistor of described the first transistor with corresponding to the 4th transistor of described transistor seconds;

Described bias regulator, to have a predetermined electric current difference, provide the 3rd and the 4th transistorized described bias current to adjust respectively, accomplish: the voltage difference that is produced in the described the 3rd and the 4th transistor becomes the voltage difference of regulation, is also catering to this adjustment described first and second bias current is adjusted.

19. low-pass filter circuit according to claim 17, wherein:

Described bias regulator, for: the temperature-compensation circuit of described first and second bias current changed with variations in temperature with being directly proportional.

20. low-pass filter circuit according to claim 1, wherein:

Described circuit element, comprise: an end, the other end that first switched-capacitor circuit and terminates at described first switched-capacitor circuit is connected on the second switch condenser network on described second filter, when described first and second switched-capacitor circuit was in first connection status, described circuit element allowed described first electric current flow;

Described current generator, be described second switch condenser network, and its capacitance relatively total of it and the capacitance of described first switched-capacitor circuit becomes described certain ratio, when described first and second switched-capacitor circuit was in second connection status, described current generator produced described second electric current.

21. low-pass filter circuit according to claim 20, wherein:

Described adder is the voltage follower circuit that is input with described second voltage;

Described first filter is a benchmark with the output voltage of described voltage follower circuit, exports described first voltage.

22. low-pass filter circuit according to claim 20, wherein:

Connecting three switched-capacitor circuit in parallel at least one circuit in described first and second switched-capacitor circuit with this switched-capacitor circuit, when a switched-capacitor circuit in two switched-capacitor circuits that this parallel connection was connected electrically in that side of the first terminal, another switched-capacitor circuit just was connected that side of second terminal.

23. a low-pass filter circuit, its differential wave with first and second input signal be input, be output with first and second the differential wave of output signal, it is characterized in that:

Comprise: be input, be the first low-pass filter circuit part of output with described first input signal with described first output signal, and

Be input, be the second low-pass filter circuit part of output with described second output signal with described second input signal;

At least one low-pass filter circuit part in described first and second low-pass filter circuit part comprises:

Be input, be first filter of output with described first or second input signal with first voltage,

The circuit element that has for described first filter, the circuit element that allows first electric current flow according to described first voltage,

Generation becomes the current generator of second electric current of certain ratio to described first electric current,

With described second electric current is input, and second voltage is second filter of output, and

With described first voltage and the described second voltage addition, export described first or the adder of second output signal.

24. low-pass filter circuit according to claim 23, wherein:

Described current generator, for: have first semiconductor element that shows first coefficient of conductivity in input one side, have demonstration becomes second coefficient of conductivity of described certain ratio with described first coefficient of conductivity second semiconductor element in output one side, with described first electric current is input, with described second electric current current mirroring circuit that is output;

Described circuit element is described first semiconductor element.

25. low-pass filter circuit according to claim 23, wherein:

Described circuit element, for: show first coefficient of conductivity, be first voltage current transformating circuit of described first electric current with described first voltage transformation;

Described current generator, for: show become second coefficient of conductivity of described certain ratio with described first coefficient of conductivity, be second voltage current transformating circuit of described second electric current with described first voltage transformation.

26. a reponse system, it allows the output clock that produces based on input clock feed back, to make this output clock have desirable characteristic, it is characterized in that:

Comprise:

Produce the charge pump circuit of charging current based on the phase difference of described input clock and the clock that fed back,

With described charging current is the loop filter of input, and

Produce the output clock generator of described output clock based on output signal from described loop filter;

Described loop filter comprises again:

With described charging current is input, is first filter of output with first voltage,

The circuit element that is had for described first filter, the circuit element that allows first electric current flow according to described first voltage,

Generate the current generator that becomes second electric current of certain ratio with described first electric current,

With described second electric current is input, is second filter of output with second voltage, and

With described first voltage and the described second voltage addition, export the adder of described output signal.

27. reponse system according to claim 26, wherein:

Described output clock generator, for: described output clock oscillation allowed, the voltage-controlled oscillator that changes based on this frequency of oscillation of described output signal from described loop filter.

28. reponse system according to claim 26, wherein:

Described output clock generator, for: based on described input clock and from the described output signal of described loop filter, the voltage control delay circuit that allows the retardation of the described relatively input clock of described output clock change.

29. reponse system according to claim 26, wherein:

Comprise: the phase difference that has based on described input clock and the clock that fed back produces the 3rd current charge pump circuit, the electric charge integrator that receives described the 3rd electric current and the stable phase angle error concealment circuit that generates the Voltage-controlled Current Source of the 4th electric current according to the voltage that is produced in the described electric charge integrator;

Described current generator, for: have according to first field-effect transistor that shows first coefficient of conductivity for first bias current that comes in input one side, have second field-effect transistor that becomes second coefficient of conductivity of described certain ratio according to second bias current demonstration that is supplied with described first coefficient of conductivity in output one side, receive described first electric current at input one side joint and receive described the 4th electric current in input one side and the either side exported in the side simultaneously, with described second electric current current mirroring circuit that is output;

Described circuit element is described first field-effect transistor.

30. reponse system according to claim 26, wherein:

This circuit element can change the shown coefficient of conductivity;

Described reponse system comprises: the offset controller that allows the coefficient of conductivity of described circuit element and described charging current change according to shared bias control signal.

31. reponse system according to claim 30, wherein:

Described current generator, for: have according to first field-effect transistor that shows first coefficient of conductivity for first bias current that comes in input one side, have second field-effect transistor that becomes second coefficient of conductivity of described certain ratio according to second bias current demonstration that is supplied with described first coefficient of conductivity in output one side, with described first electric current is input, with described second electric current current mirroring circuit that is output;

Described circuit element is described first field-effect transistor;

Described substrate bias controller is the controller that is allowed described first and second bias current, described charging current change by described bias voltage control signal.

32. reponse system according to claim 30, wherein:

Described circuit element, for: show first coefficient of conductivity, be first voltage current transformating circuit of described first electric current with described first voltage transformation;

Described current generator, for: show become second coefficient of conductivity of described certain ratio with described first coefficient of conductivity, be second voltage current transformating circuit of described second electric current with described first voltage transformation;

Can change described first, second coefficient of conductivity of described first, second current conversion circuit;

Described substrate bias controller is the controller that is allowed described first and second coefficient of conductivity, described charging current change by described bias voltage control signal.

33. reponse system according to claim 30, wherein:

Described bias control signal is based on that described output signal from described loop filter produces.

34. reponse system according to claim 30, wherein:

Described adder is an operational amplifier;

Described offset controller is for being allowed the controller of described bandwidth of operational amplifiers characteristic variations by described bias control signal.

35. feedback circuit according to claim 33, wherein:

Comprise: be set at from first state of the output of described adder with the described output signal of described loop filter in described output signal and be set at the starter that switches between second state of certain voltage described loop filter;

Described starter is set at described second state when this reponse system of starting.

36. reponse system according to claim 35, wherein:

Described adder, for: negative feedback have part ownership described second filter, with described first voltage for the input of in-phase input terminal, output tertiary voltage being made the operational amplifier of described output signal;

Described starter has: the switch that switches between short circuit between the input terminal of described operational amplifier and disconnection;

Described switch, when described first state that described input terminal is separated, and when described second state with described input terminal between short circuit.

37. reponse system according to claim 35, wherein:

Described starter has the switch that switching internal electric source and external power source are made the power supply of described certain voltage.

38. a reponse system, it allows the output clock that produces based on input clock feed back, to make this output clock have desirable characteristic, it is characterized in that:

Comprise:

Produce the charge pump circuit of first and second charging current based on the phase difference of described input clock and the clock that fed back,

Differential wave with described first and second charging current is input, is the loop filter of output with first and second output signal,

Differential wave with described first and second output signal is input, produces the output clock generator of described output clock;

Described loop filter comprises: with described first charging current is input, with described first output signal be output first filter circuit, and be input with described second charging current, be second filter circuit of output with described second output signal;

At least one low-pass filter circuit in described first and second low-pass filter circuit comprises:

With at least one charging current in described first and second charging current is input, exports first filter of first voltage,

The circuit element that is had for described first filter, the circuit element that allows first electric current flow according to described first voltage,

Generate the current generator that becomes second electric current of certain ratio with described first electric current,

With described second electric current is input, is second filter of output with second voltage, and

With described first voltage and the described second voltage addition, export the adder of at least one output signal in described first and second output signal.

39. a semiconductor integrated circuit is characterized in that:

Be one to have the semiconductor integrated circuit of the described low-pass filter circuit of claim 1.

40. a semiconductor integrated circuit is characterized in that:

Be one to have the semiconductor integrated circuit of the described reponse system of claim 26.

41. according to the described semiconductor integrated circuit of claim 40, wherein:

This semiconductor integrated circuit is used for IC-card.

42. according to the described semiconductor integrated circuit of claim 40, wherein:

This semiconductor integrated circuit is chip on the chip (chip on chip) structure,

Described reponse system is installed in the top section of chip structure on the described chip.

43. according to the described semiconductor integrated circuit of claim 40, wherein:

Described reponse system is installed in the weld pad zone of this semiconductor integrated circuit.

44. according to the described semiconductor integrated circuit of claim 40, wherein:

This semiconductor integrated circuit is a microprocessor.

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