CN117060708B - Single-stage bridgeless PFC converter and control method - Google Patents

Single-stage bridgeless PFC converter and control method Download PDF

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Publication number
CN117060708B
CN117060708B CN202311056922.XA CN202311056922A CN117060708B CN 117060708 B CN117060708 B CN 117060708B CN 202311056922 A CN202311056922 A CN 202311056922A CN 117060708 B CN117060708 B CN 117060708B
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converter
transformer
mode
input
voltage
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CN117060708A (en
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李浩昱
丁明远
邢延林
王鹏飞
叶一舟
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Harbin Institute of Technology
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Harbin Institute of Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H02M1/4258—Arrangements for improving power factor of AC input using a single converter stage both for correction of AC input power factor and generation of a regulated and galvanically isolated DC output voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Rectifiers (AREA)
  • Dc-Dc Converters (AREA)

Abstract

The single-stage bridgeless PFC converter and the control method solve the problems that the single-stage PFC converter is complex in structure and limited in output voltage range, and belong to the topology field of the single-stage single-phase bridgeless PFC converter. The converter comprises a filter inductor, a bidirectional switch, an energy storage capacitor, a transformer, a diode D 1-D2, a filter capacitor C dc1-Cdc2 and a control circuit; when the converter works in a critical conduction state and works in a buck mode, the control circuit controls the converter according to a buck mode control equation V mTon＝|vac |; when the power-saving control circuit works in the boost mode, the control circuit controls the converter according to a boost mode control equation V mTon＝|vac|+nVdc/2; the switching between the buck mode and the boost mode can be flexibly switched, the conduction time T on of the converter is controlled according to control equations in different working modes, the problem of input current distortion in the buck mode is effectively solved, and the output voltage range is remarkably widened.

Description

Single-stage bridgeless PFC converter and control method
Technical Field
The invention relates to a single-stage bridgeless PFC converter with a wider output voltage range, and belongs to the topology field of single-stage single-phase bridgeless PFC converters.
Background
In the AC/DC conversion occasion, totem pole PFC, dual-boost bridgeless PFC and three-level bridgeless PFC are widely applied because of the higher efficiency of the bridgeless structure. However, these bridgeless PFC converters have evolved from Boost-type PFC converters, and their inherent Boost characteristics necessitate that the output voltage of such circuits be greater than the input ac voltage peak. For the application occasion requiring wider direct current output voltage range, a one-stage DC/DC conversion circuit needs to be added behind the direct current output voltage range to realize the voltage reduction function, and the two-stage conversion inevitably brings the defects of reduced efficiency, reduced power density and the like.
To widen the output voltage range of a single-stage bridgeless PFC converter, a single-stage PFC converter based on a SEPIC circuit is proposed. The converter is derived from a SEPIC circuit, and output voltage can be flexibly increased and decreased, but the topology is less at present due to the complexity of the circuit structure, and the converter mainly comprises a bridgeless PFC converter with double SEPIC conversion units and a simplified topology thereof. The bridgeless PFC converter with the double SEPIC conversion units adopts two groups of SEPIC circuits to respectively work on positive and negative half cycles of alternating current input voltage so as to realize power factor correction, and the defects of more components and complex structure are necessarily caused by adopting two groups of parallel structures. Therefore, the simplified topology of the bridgeless PFC converter with the double SEPIC conversion units is generated, the two groups of SEPIC units are simplified into one group, and the utilization rate of components is high. However, the transformer used in the simplified topology is a secondary double winding, and the design of the transformer is relatively complex; meanwhile, when the converter works in a buck mode, direct current components exist in the input current of the converter, the sine degree of the input current is seriously influenced, a good PFC effect cannot be achieved, and the lower limit of regulation of the output voltage of the converter is limited.
Disclosure of Invention
Aiming at the problems that the single-stage PFC converter is complex in structure and limited in output voltage range, the invention provides the single-stage bridgeless PFC converter which is simple in structure and wide in output voltage range.
The invention discloses a single-stage bridgeless PFC converter which comprises a filter inductor, a bidirectional switch, an energy storage capacitor, a transformer T, a diode D 1-D2 and a filter capacitor C dc1-Cdc2;
the positive electrode of the input power supply is connected with one end of the filter inductor, the other end of the filter inductor is simultaneously connected with one end of the two-way switch and one end of the energy storage capacitor, the other end of the energy storage capacitor is connected with the same-name end of the primary side of the transformer T, and the other end of the two-way switch is connected with the different-name end of the primary side of the transformer T;
The same-name end of the secondary side of the transformer T is connected with the anode of the diode D 1 and the cathode of the diode D 2 at the same time;
the synonym end of the secondary side of the transformer T is connected with the negative electrode of the filter capacitor C dc1 and the positive electrode of the filter capacitor C dc2 at the same time;
The cathode of the diode D 1 and the anode of the filter capacitor C dc1 are connected;
The negative electrode of the filter capacitor C dc2 and the anode of the diode D 2 are connected.
The control method of the single-stage bridgeless PFC converter comprises the following steps:
the converter works in a critical conduction mode;
if |v ac|>nVdc/2, the converter works in a buck mode, and the converter is controlled according to a buck mode control equation V mTon＝|vac |; if |v ac|<nVdc/2, the converter works in the boost mode, and the converter is controlled according to the boost mode control equation V mTon＝|vac|+nVdc/2;
T on represents the conduction time of the bidirectional switch tube in one switch period, the turns ratio of the primary side and the secondary side of the transformer T is 1, V dc is the load voltage, v ac is the input voltage of the input power supply, and r ac is the equivalent input resistance of the alternating current side; l m is the excitation inductance parameter of the transformer T,
The converter of the invention also comprises a control circuit, wherein the control circuit comprises a voltage compensator, a reset integrator, a zero crossing detection circuit, a comparator and an RS trigger;
The load voltage V dc is compared with the voltage given value V dc_ref, the difference value is subjected to closed loop regulation by a voltage compensator, the output value V m of the voltage compensator is used as an integrated signal of a reset integrator, a zero-crossing detection circuit is used for detecting a ZCD signal of the midpoint current i D of the filter capacitor at the output side and is used as a reset signal of the reset integrator to trigger a rising edge, and the ZCD signal is simultaneously input to the S end of the RS trigger; resetting the integral output of the integrator to V mTon;
the integral output and the comparison value of the reset integrator are respectively input to the positive input end and the negative input end of the comparator, the output of the comparator is connected with the R end of the RS trigger, and the Q end of the RS trigger outputs a driving signal of the bidirectional switching tube;
When the converter works in a buck mode, the comparison value is |v ac |;
When the converter is operated in boost mode, the comparison value is |v ac|+nVdc/2.
Preferably, if |v ac|>nVdc/2, it is considered to operate in buck mode, and if |v ac|<nVdc/2, it is considered to operate in boost mode.
Preferably, the switching period T S of the buck mode is:
where P ac is the input power, V ac_rms is the ac input voltage effective value, and L m is the excitation inductance parameter of the transformer T.
Preferably, the switching period T S of the boost mode is:
where P ac is the input power, V ac_rms is the ac input voltage effective value, and L m is the excitation inductance parameter of the transformer T.
Preferably, the excitation inductance parameter L m of the transformer T is:
Wherein T S_max is the upper limit of the switching period T S.
Preferably, the inductance value L of the filter inductance is:
Wherein α is a current ripple coefficient.
The invention has the beneficial effects that the power factor correction of the alternating current side and the wide voltage range output of the direct current side can be realized by adopting a simple circuit structure and a flexible control method. The advantages of the converter topology are mainly represented by: compared with 6 switching devices, 2 energy storage capacitors and 2 energy storage inductors of a bridgeless PFC converter with double SEPIC conversion units, the topology adopts 4 semiconductor power devices, 1 energy storage capacitor and 1 energy storage inductor, the number of the switching devices is reduced, and the number of the inductors and the number of the capacitors are halved; compared with the bridge-free PFC converter with double SEPIC conversion units, the topology is simplified, only one winding is needed on the secondary side of the transformer, and the transformer is simple in structure and easy to manufacture. In addition, the converter works in a critical conduction state, can be flexibly switched between a buck mode and a boost mode, and can control the conduction time T on of the converter according to control equations in different working modes, so that the problem of input current distortion in the buck mode is effectively solved, and the output voltage range is remarkably widened.
Drawings
Fig. 1 is a circuit diagram of a single-stage bridgeless PFC converter of the present invention;
fig. 2 is a diagram showing the division of the working area of the converter, wherein the ordinate v represents the magnitude of the input voltage, and the abscissa t represents the working time;
FIG. 3 is a diagram showing the waveforms of the inductor L, L m current, the diode D 1、D2 current, and the switching tube S 1、S2 voltage of the CRM buck mode positive input voltage stage converter during a switching cycle;
FIG. 4 is a diagram of two modes of the CRM buck mode positive input voltage stage converter during a switching cycle, wherein FIG. 4 (a) is mode I and FIG. 4 (b) is mode II;
FIG. 5 is a diagram showing the waveforms of the inductance L, L m current, the diode D 1、D2 current, and the switching tube S 1、S2 voltage of the CRM buck mode negative input voltage stage converter during a switching cycle;
FIG. 6 is a diagram of two modes of the CRM buck mode negative input voltage stage converter during a switching cycle, wherein FIG. 6 (a) is mode III and FIG. 6 (b) is mode IV;
FIG. 7 is a diagram showing the waveforms of the inductor L, L m current, the diode D 1、D2 current, and the switching tube S 1、S2 voltage of the CRM boost mode positive input voltage stage converter during a switching cycle;
FIG. 8 is a diagram of two modes of the CRM boost mode positive input voltage stage converter during a switching cycle, wherein FIG. 8 (a) is mode V and FIG. 8 (b) is mode VI;
FIG. 9 is a diagram showing the waveforms of the inductor L, L m current, the diode D 1、D2 current, and the switching tube S 1、S2 voltage of the CRM boost mode negative input voltage stage converter during a switching cycle;
FIG. 10 is a diagram of two modes of the CRM boost mode negative input voltage stage converter during a switching cycle, wherein FIG. 10 (a) is mode VII and FIG. 10 (b) is mode VIII;
FIG. 11 is a functional block diagram of a multi-mode switching CRM control method;
Fig. 12 is a graph comparing input voltage and current waveforms before and after the input of 100V/50Hz and the output of 100V/200W by the proposed control method, wherein fig. 12 (a) is an input voltage and current waveform controlled by DCM duty cycle, and fig. 12 (b) is an input voltage and current waveform controlled by the proposed multimode switching CRM control method.
Detailed Description
The following description of the embodiments of the present invention will be made clearly and completely with reference to the accompanying drawings, in which it is apparent that the embodiments described are only some embodiments of the present invention, but not all embodiments. All other embodiments, which can be made by those skilled in the art based on the embodiments of the invention without making any inventive effort, are intended to be within the scope of the invention.
It should be noted that, without conflict, the embodiments of the present invention and features of the embodiments may be combined with each other.
The invention is further described below with reference to the drawings and specific examples, which are not intended to be limiting.
The single-stage bridgeless PFC converter of the embodiment comprises a filter inductor L, a small-capacity energy storage capacitor C, a transformer T (excitation inductor L 2 and transformation ratio n: 1), two filter capacitors C dc1、Cdc2, a bidirectional switch (two groups of reverse series power switching tubes S 1 and S 2) and two diodes D 1、D2. The capacitance values of the two filter capacitors C dc1、Cdc2 are the same, and the switching actions of the switching tubes of the two-way switch are the same.
Before analyzing the working principle of the converter, the following description is made:
1) The converter operates in a critical conduction mode (CRM);
2) Neglecting the influence of parasitic parameters, conduction voltage drop and line parameters of the used components;
3) The capacitance of the filter capacitor C dc1、Cdc2 is large enough, and the partial pressure of the two filter capacitors is equal, namely v Cdc1＝vCdc2＝Vdc/2;
4) The alternating input voltage is an ideal sine wave, and the voltage at two ends of the energy storage capacitor is equal to the input alternating voltage, namely v C＝vac;
5) The switching frequency f S is far greater than the power frequency, and the input voltage and the energy storage capacitor voltage in the switching period T S are considered constant.
According to the instantaneous value of the AC input voltage of the converter, the mode of operation of the converter is divided into a buck mode and a boost mode by taking nV dc/2 as a dividing line. As shown in fig. 2, in a power frequency period, the instantaneous value of the ac input voltage in the intervals of [ t 1,t2 ] and [ t 3,t4 ] is greater than nV dc/2, and the converter works in a buck mode; the instantaneous value of the alternating current input voltage in other intervals is smaller than nV dc/2, and the converter works in a boosting mode.
According to the positive and negative intervals of the input voltage and the switching actions of the switching tubes S 1 and S 2 of the two-way switch, the converter has four working modes in the step-down mode. The main voltage and current waveforms of the positive input voltage stage circuit are shown in fig. 3, and the corresponding working mode diagram is shown in fig. 4; the main voltage and current waveforms of the negative input voltage stage circuit are shown in fig. 5, and the corresponding working mode diagram is shown in fig. 6.
Modality I (as shown in fig. 4 (a)): the mode starts at the conduction time of the switching tubes S 1 and S 2 of the bidirectional switch, the inductor L is charged linearly in the forward direction through the switching tube of the bidirectional switch, and the energy storage capacitor charges the exciting inductor of the transformer. At this time, the diode D 1 is turned off by the back voltage, the voltage across the diode D 2 is |v ac|/n-Vdc/2>0,D2, and the exciting inductance energy is released to the filter capacitor and the load through the secondary side diode D 2 of the transformer.
Modality II (as shown in fig. 4 (b)): the mode starts at the moment when the switching tubes S 1 and S 2 of the bidirectional switch are turned off, and the two ends of the switching tube S 1 bear pressure to be V DS_max. The diode D 1 is turned on, the filter inductor L and the exciting inductor L m are discharged to the load side together through the secondary side diode D 1 of the transformer, the current of the filter inductor L and the exciting inductor L m starts to linearly decrease, the current i D1 flowing through the diode D 1 linearly decreases from the peak value until i D1 decreases to 0, and the switching tubes S 1 and S 2 triggering the bidirectional switch are turned on to enter the next switching period.
Modality III (as shown in fig. 6 (a)): the mode starts at the conduction time of the switching tubes S 1 and S 2 of the bidirectional switch, the filter inductor L is charged in a reverse linear mode through the switching tubes of the bidirectional switch, and the energy storage capacitor charges the exciting inductor of the transformer. At this time, the diode D 2 is turned off by the back voltage, the voltage across the diode D 1 is |v ac|/n-Vdc/2>0,D1, and the exciting inductance energy is released to the filter capacitor and the load through the secondary side diode D 1 of the transformer.
Modality IV (as shown in fig. 6 (b)): the mode starts at the moment when the switching tubes S 1 and S 2 of the bidirectional switch are turned off, and the two ends of the switching tube S 2 bear pressure to be V DS_max. The diode D 2 is turned on, the filter inductor L and the exciting inductor L m are discharged to the load side together through the secondary side diode D 2 of the transformer, the current of the filter inductor L and the exciting inductor L m starts to linearly decrease, the current i D2 flowing through the diode D 2 linearly decreases from the peak value until i D2 decreases to 0, and the switching tubes S 1 and S 2 triggering the bidirectional switch are turned on to enter the next switching period.
During mode I/III, the secondary side diode D 2/D1 current increases linearly from 0 until the switching tubes S 1 and S 2 of the bi-directional switch turn off at peak I D_max1. During mode II/IV, the secondary side diode D 1/D2 current decreases linearly from the current peak i D_max2 until it drops to 0. The current peaks of the two diodes during the switching cycle can be expressed as:
Where T on represents the on-time of the switching tube of the bi-directional switch in one switching cycle.
By observing the waveforms of the two diode currents in the switching period, the waveforms of the two diode currents are triangular waves, so that the expression of the diode currents and the average value i D_avg in the switching period can be obtained:
Where d is the duty cycle and d=t on/TS.
Based on the conservation of input and output power, an expression of alternating input current is obtained:
to achieve unity power factor correction at the input side, the input current expression needs to satisfy:
Where r ac is the ac side equivalent input resistance.
By combining the converter input current expression and the ideal expression given above, the relationship satisfied by the on-time T on in buck mode can be obtained:
Order the Can be obtained by deformation of the upper part
VmTon＝|vac|
The control circuit controls the converter according to a buck mode control equation V mTon＝|vac, and the control circuit comprises a voltage compensator, a reset integrator, a zero crossing detection circuit, a comparator and an RS trigger; the load voltage V dc is compared with the voltage given value V dc_ref, the difference value is subjected to closed loop regulation by a voltage compensator, the output value V m of the voltage compensator is used as an integrated signal of a reset integrator, a zero-crossing detection circuit is used for detecting a ZCD signal of the midpoint current i D of the filter capacitor at the output side and is used as a reset signal of the reset integrator to trigger a rising edge, and the ZCD signal is simultaneously input to the S end of the RS trigger; resetting the integral output of the integrator to V mTon; the integral output and the comparison value of the reset integrator are respectively input to the positive input end and the negative input end of the comparator, the output of the comparator is connected with the R end of the RS trigger, and the Q end of the RS trigger outputs a driving signal of the bidirectional switching tube; a block diagram of the control method of the control circuit is shown in fig. 11. The control method is to conduct at fixed time and control turn-off, the conducting signal comes from Zero Crossing Detection (ZCD) signal of midpoint current i D of the filter capacitor at the output side, the conducting time is determined by the internal control method, and V m is obtained through the outer ring of the output voltage. The output side filter capacitor midpoint current I D is chosen because it is the sum of the diode D 1 and D 2 currents, and the zero crossing time of this current is the starting time of the mode I/III. The specific implementation process of the on-time control is as follows: the output voltage outer ring sampling converter outputs a DC output voltage V dc and compares the DC output voltage V dc with a voltage given value V dc_ref, and after the generated error value is subjected to closed loop adjustment by the voltage compensator, the output value V m of the voltage compensator is used as an integrated signal for resetting the integrator. The reset integrator has the capability of automatic reset every switching period, and the reset signal is from the ZCD signal of the midpoint current i D of the filter capacitor at the output side and is triggered by the rising edge. Meanwhile, the input voltage is sampled and the comparison value |v ac | is obtained through absolute value operation. When the ZCD signal arrives, the switching tube of the bidirectional switch is turned on, the integrator integrates the V m, when the output value V mTon reaches the comparison value, the comparator turns over, and the switching tube of the bidirectional switch is turned off, so that one switching period is completed.
When the instantaneous value of the AC input voltage is less than nV dc/2, the converter is operated in the boost mode. The converter has four working modes in CRM according to the positive and negative intervals of the input voltage and the switching actions of the switching tubes S 1 and S 2 of the two-way switch. The main voltage and current waveforms of the positive input voltage stage circuit are shown in fig. 7, and the corresponding working mode diagram is shown in fig. 8; the main voltage and current waveforms of the negative input voltage stage circuit are shown in fig. 9, and the corresponding working mode diagram is shown in fig. 10.
Modality V (as shown in fig. 8 (a)): the mode starts at the conduction time of the switching tubes S 1 and S 2 of the bidirectional switch, the inductor L is charged linearly in the forward direction through the switching tubes of the bidirectional switch, and the energy storage capacitor charges the excitation inductor of the transformer. At this time, the diode D 1 is turned off by the back voltage, the voltage across the diode D 2 is |v ac|/n-Vdc/2<0,D2, and the load is powered by the two filter capacitors.
Modality VI (as shown in fig. 8 (b)): the mode starts at the moment when the switching tubes S 1 and S 2 of the bidirectional switch are turned off, and the two ends of the switching tube S 1 bear pressure to be V DS_max. Diode D 1 is turned on, inductance L and L m are discharged to the load side together through transformer secondary side diode D 1, the current of L and L m starts to linearly decrease, current i D1 flowing through D 1 linearly decreases from the peak value until i D1 decreases to 0, and switching tubes S 1 and S 2 triggering the bidirectional switch are turned on to enter the next switching cycle.
Modality VII (as shown in fig. 10 (a)): the mode starts at the conduction time of the switching tubes S 1 and S 2 of the bidirectional switch, the inductor L is charged in a reverse linear mode through the switching tubes of the bidirectional switch, and the energy storage capacitor charges the excitation inductor of the transformer. At this time, the diode D 2 is turned off by the back voltage, the voltage across the diode D 1 is |v ac|/n-Vdc/2<0,D1, and the load is powered by the two filter capacitors.
Modality VIII (as shown in fig. 10 (b)): the mode starts at the moment when the switching tubes S 1 and S 2 of the bidirectional switch are turned off, and the two ends of the switching tube S 2 bear pressure to be V DS_max. Diode D 2 is turned on, inductance L and L m are discharged to the load side together through transformer secondary side diode D 2, the current of L and L m starts to linearly decrease, current i D2 flowing through D 2 linearly decreases from the peak value until i D2 decreases to 0, and switching tubes S 1 and S 2 triggering the bidirectional switch are turned on to enter the next switching cycle.
During mode V/VII, the secondary side diode D 2/D1 current is always 0. During mode VI/VIII, the secondary side diode D 1/D2 current decreases linearly from the current peak i D_max until it drops to 0. The current peak of the diode during the switching period can be expressed as:
By observing the waveform of the secondary side diode current in the switching period, the waveform is triangular, so that the expression of the diode current and the average value i D_avg in the switching period can be obtained:
based on the conservation of input and output power, an expression of alternating input current is obtained:
to achieve unity power factor correction at the input side, the input current expression also needs to satisfy:
combining the converter input current expression given above with the ideal expression, the relationship satisfied by the on-time T on in boost mode can be obtained:
Order the Can be obtained by deformation of the upper part
VmTon＝|vac|+nVdc/2
The above equation is a CRM control equation under the boost mode of the converter, the control circuit controls the converter according to the boost mode control equation V mTon＝|vac|+nVdc/2, and the control method has a structure block diagram shown in FIG. 11. The control method is also timing on and off, the on signal comes from Zero Crossing Detection (ZCD) signal of midpoint current i D of the filter capacitor at the output side, the on time is determined by the internal control method, and V m is obtained through the output voltage outer ring. The specific implementation process of the on-time control is as follows: the output voltage outer ring sampling converter outputs a DC output voltage V dc and compares the DC output voltage V dc with a voltage given value V dc_ref, and after the generated error value is subjected to closed loop adjustment by the voltage compensator, the output value V m of the voltage compensator is used as an integrated signal for resetting the integrator. The reset integrator has the capability of automatic reset every switching period, and the reset signal is from the ZCD signal of the midpoint current i D of the filter capacitor at the output side and is triggered by the rising edge. Meanwhile, input voltage is sampled, absolute value operation is carried out to obtain |v ac |, and comparison value |v ac|+nVdc/2 is obtained through operation with output voltage V dc. When the ZCD signal arrives, the switching tube of the bidirectional switch is turned on, the integrator starts to integrate the V m, when the output value V mTon reaches the comparison value, the comparator turns over, and the switching tube of the bidirectional switch is turned off, so that one switching period is completed.
Comparing the circuit operation in buck and boost modes, the two modes of control differ primarily in the CRM control equation. The overall control process of the converter is shown in fig. 11, firstly, the magnitude of the instantaneous value of the input voltage is judged, if |v ac|>nVdc/2, the converter is considered to work in a buck mode, and the converter is controlled according to a buck mode control equation V mTon＝|vac |; if |v ac|<nVdc/2, then it is considered to be operating in boost mode and controlled according to boost mode control equation V mTon＝|vac|+nVdc/2.
Meanwhile, the volt-second balance equation of the voltage switching period at two ends of the inductor L in two modes is the same equation, and the equation is expressed as follows:
The above simplification can be obtained:
From an observation of the converter output gain expression, when the transformer transformation ratio n is fixed, the output gain of the converter depends on the magnitude of the d value, and wide-range variation of the output voltage can be realized through closed-loop regulation of the output voltage.
Based on the previous analysis, the switching period T S in buck and boost modes, respectively, can be expressed as:
Where P ac is the input power and V ac_rms is the effective value of the AC input voltage.
Looking at the switching period expression, it can be seen that the switching period of the converter is directly related to the circuit operating index. Under the condition that the input voltage, the output voltage, the input power, the exciting inductance and the transformer transformation ratio are fixed, the switching period reaches the maximum at the peak value of the input voltage in the step-down mode, and the minimum near the zero crossing of the input voltage in the step-up mode. The excitation inductance parameter L m can be designed according to the above formula in combination with the upper limit of the switching period, and the design formula is as follows:
Wherein T S_max is the upper limit value of the designed switching period.
In addition, the design of the input filter inductance mainly considers the maximum current ripple. As can be seen from the foregoing operating principle, the maximum value of the current ripple occurs at the peak of the input voltage, which can be expressed as:
Thus, in the case of determining the excitation inductance size and the current ripple coefficient α, the filter inductance L can be designed by the following formula:
Wherein α is a current ripple coefficient.
The design of inductance parameters is carried out by taking 100V/50Hz of power frequency alternating current input voltage and 200W of power and 100V of output voltage as examples.
The design switching frequency is 50kHz, the transformer transformation ratio n=1, and the transformer transformation ratio is obtained by substituting an excitation inductance design formula:
and taking the current ripple coefficient alpha=0.2 to obtain the inductance value of the input inductor.
Based on the inductance parameters, simulation verification is performed on the input side power factor correction condition of the converter under the condition of adopting DCM (discontinuous control) duty cycle control and proposed multimode switching CRM (power factor correction), and simulation results are shown in fig. 12, wherein fig. 12 (a) is DCM duty cycle control, and fig. 12 (b) is multimode switching CRM control. From simulation results, the problem of distortion of the input current of the converter is effectively solved when the CRM control is switched in a multi-mode, the current presents sine waves with the same phase with the alternating input voltage, and the power factor correction effect of the input side is obviously improved.
Although the invention herein has been described with reference to particular embodiments, it is to be understood that these embodiments are merely illustrative of the principles and applications of the present invention. It is therefore to be understood that numerous modifications may be made to the illustrative embodiments and that other arrangements may be devised without departing from the spirit and scope of the present invention as defined by the appended claims. It should be understood that the different dependent claims and the features described herein may be combined in ways other than as described in the original claims. It is also to be understood that features described in connection with separate embodiments may be used in other described embodiments.

Claims (9)

1. The single-stage bridgeless PFC converter is characterized by comprising a filter inductor, a bidirectional switch, an energy storage capacitor, a transformer T, a diode D 1-D2 and a filter capacitor C dc1-Cdc2;
the positive electrode of the input power supply is connected with one end of the filter inductor, the other end of the filter inductor is simultaneously connected with one end of the two-way switch and one end of the energy storage capacitor, the other end of the energy storage capacitor is connected with the same-name end of the primary side of the transformer T, and the other end of the two-way switch is connected with the different-name end of the primary side of the transformer T;
The same-name end of the secondary side of the transformer T is connected with the anode of the diode D 1 and the cathode of the diode D 2 at the same time;
the synonym end of the secondary side of the transformer T is connected with the negative electrode of the filter capacitor C dc1 and the positive electrode of the filter capacitor C dc2 at the same time;
The cathode of the diode D 1 and the anode of the filter capacitor C dc1 are connected;
The negative electrode of the filter capacitor C dc2 and the anode of the diode D 2 are connected;
the converter further comprises a control circuit, wherein the control circuit comprises a voltage compensator, a reset integrator, a zero crossing detection circuit, a comparator and an RS trigger;
The load voltage V dc is compared with the voltage given value V dc_ref, the difference value is subjected to closed loop regulation by a voltage compensator, the output value V m of the voltage compensator is used as an integrated signal of a reset integrator, a zero-crossing detection circuit is used for detecting a ZCD signal of the midpoint current i D of the filter capacitor at the output side and is used as a reset signal of the reset integrator to trigger a rising edge, and the ZCD signal is simultaneously input to the S end of the R S trigger; resetting the integral output of the integrator to V mTon;Ton represents the conduction time of the bidirectional switching tube in one switching period;
the integral output and the comparison value of the reset integrator are respectively input to the positive input end and the negative input end of the comparator, the output of the comparator is connected with the R end of the R S trigger, and the Q end of the RS trigger outputs a driving signal of the bidirectional switching tube;
if |v ac|>nVdc/2, the converter is operated in buck mode, the comparison value is |v ac |;
If |v ac|<nVdc/2, the converter is operated in the boost mode, the comparison value is |v ac|+nVdc/2;
v ac is the input voltage of the input power supply, and the turns ratio of the primary side and the secondary side of the transformer T is 1.
2. The single-stage bridgeless PFC converter according to claim 1, wherein the switching period T S for buck mode is:
where P ac is the input power, V ac_rms is the ac input voltage effective value, and L m is the excitation inductance parameter of the transformer T.
3. The single-stage bridgeless PFC converter according to claim 1, wherein the boost mode switching period T S is:
where P ac is the input power, V ac_rms is the ac input voltage effective value, and L m is the excitation inductance parameter of the transformer T.
4. A single-stage bridgeless PFC converter according to claim 2 or claim 3, wherein the excitation inductance parameter L m of the transformer T is:
Wherein T S_max is the upper limit of the switching period T S.
5. The single-stage bridgeless PFC converter according to claim 4, wherein the inductance L of the filter inductor is:
Wherein α is a current ripple coefficient.
6. A method of controlling a single-stage bridgeless PFC converter according to claim 1, wherein said converter is operated in a critical conduction mode, said method comprising:
if |v ac|>nVdc/2, the converter works in a buck mode, and the converter is controlled according to a buck mode control equation V mTon＝|vac |; if |v ac|<nVdc/2, the converter works in the boost mode, and the converter is controlled according to the boost mode control equation V mTon＝|vac|+nVdc/2;
r ac is the equivalent input resistance of the alternating current side; l m is the excitation inductance parameter of the transformer T,
7. The control method according to claim 6, wherein the switching period T S of the buck mode is:
The switching period T S of the boost mode is:
where P ac is the input power, V ac_rms is the ac input voltage effective value, and L m is the excitation inductance parameter of the transformer T.
8. The control method according to claim 7, wherein the excitation inductance parameter L m of the transformer T is:
Wherein T S_max is the upper limit of the switching period T S.
9. The control method according to claim 8, wherein the inductance value L of the filter inductance is:
Wherein α is a current ripple coefficient.
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