CN115460509B

CN115460509B - Transducer bandwidth widening method and device using nonlinear non-foster system

Info

Publication number: CN115460509B
Application number: CN202211071042.5A
Authority: CN
Inventors: 杨鑫; 许梦伟; 张智贺; 李姝汛; 欧阳晓平
Original assignee: Changsha Semiconductor Technology And Application Innovation Research Institute; Hunan University
Current assignee: Changsha Semiconductor Technology And Application Innovation Research Institute; Hunan University
Priority date: 2022-09-02
Filing date: 2022-09-02
Publication date: 2024-06-04
Anticipated expiration: 2042-09-02
Also published as: CN115460509A

Abstract

The invention provides an electroacoustic transducer bandwidth widening method by utilizing a nonlinear non-foster system, which comprises the following steps: calculating a modulation degree m' and a modulation phase angle theta of the single-phase full-bridge inverter; in a first concentrated parameter model of the electroacoustic transducer with inductive input impedance, a first static equivalent resistance, an equivalent inductance model and a first vibration system equivalent circuit structure are mutually connected in series between two driving ends of the electroacoustic transducer; the equivalent inductance model is an eddy current impedance model formed by connecting a nonlinear inductance model and a nonlinear resistance model in series; the structure of the reactance model and the resistance model which are connected in series with each other and the structure of the first static equivalent resistance and equivalent inductance model which are connected in series with the equivalent circuit structure of the first vibration system are equivalent structures.

Description

Transducer bandwidth widening method and device using nonlinear non-foster system

Technical Field

The invention relates to the technical field of underwater acoustic communication, in particular to a method and a device for widening the bandwidth of a transducer by using a nonlinear non-foster system, which are used for broadband live acoustic emission.

Background

With the continuous development of underwater wireless communication technology and detection means, sound waves become the only carrier capable of transmitting information underwater in a long distance due to the advantages of high wave speed, low attenuation frequency and the like of the sound waves in an aqueous medium. Thus, the underwater sound technology capable of researching the underwater sound propagation rule and underwater signal processing is derived, and a key ring in the underwater sound technology is an electroacoustic transmitting device. The system can complete high-power energy conversion and realize remote transmission of information, and mainly comprises an electromagnetic energy storage element and a mechanical vibration system: the energy storage element is responsible for completing motor conversion under a certain physical effect, and the mechanical vibration system is responsible for outputting converted energy, namely sound energy.

In practical application of the electroacoustic transmitting device, relatively serious impedance mismatch exists between a driving power supply and the device, reactive power loss is generated, and the maximum power cannot be obtained from a power supply end, so that the output power and the signal transmission efficiency of a system are affected. In order to improve the transmission efficiency and quality of the high-power electroacoustic transmitting device, a broadband impedance matching network is required to be additionally arranged to offset reactive power loss caused by equivalent reactance of the broadband impedance matching network, and the resonance bandwidth of the system is expanded. The traditional passive matching mode is limited by the gain bandwidth theory, can only realize matching at a single frequency, and cannot meet the broadband matching requirement.

Existing non-foster matching systems are composed of transistors and operational amplifiers, which have extremely limited output capabilities and, due to limitations in their operating principles and internal structures, exhibit linear characteristics in their output characteristics. The linear non-foster matching system is only suitable for high-frequency-band micro-power sound sources, and is currently applied to miniature devices with almost unchanged input reactance in the working frequency range of miniature antennas and the like. However, due to the existence of dynamic mechanical impedance, the impedance characteristic of the high-power electroacoustic transmitting device commonly used in the field of underwater acoustic communication is not completely consistent with the static impedance of an electric end in the broadband operation process, so that the input impedance characteristic of the system is not fixed, and the system can be changed along with the change of frequency.

In addition, the high-power electroacoustic transmitting device is in a complex working environment, and a plurality of obvious nonlinear phenomena exist in the working process of the high-power electroacoustic transmitting device, so that the input impedance characteristic of the device presents nonlinear characteristics. This means that the transducer is often operated in a nonlinear state, energy loss is generated in the operation process, temperature disturbance is caused, nonlinear harmonic distortion is caused to the output response of the transducer, and the output performance of the transducer is seriously affected. The input reactance of the electroacoustic transducer can generate nonlinear change along with the change of working conditions and external disturbance, so that a constant output matching network cannot accurately match with the high-power electroacoustic transducer, the power magnitude and matching linearity (self-adaption degree) of the traditional linear non-foster matching system cannot meet the requirement of a large bandwidth of a high-power electroacoustic transmitting device, applicability is extremely limited, the requirement of broadband real-time adjustment of the high-power electroacoustic transmitting device cannot be met, and the energy transmission efficiency of the system is reduced.

Disclosure of Invention

The invention aims to solve the problem that a linear non-foster matching system formed by a transistor and an operational amplifier cannot meet the requirement of a large bandwidth of a high-power electroacoustic transducer in the prior art, and provides a bandwidth widening method and device for the electroacoustic transducer by using a switch-type non-linear non-foster system.

In order to solve the technical problems, the invention adopts the following technical scheme: an electroacoustic transducer bandwidth widening method utilizing a switch-type nonlinear non-foster system, wherein the input impedance of the electroacoustic transducer is inductive, a driving signal of the electroacoustic transducer is provided by a power amplifier, the electroacoustic transducer is provided with a first concentrated parameter model, and the first concentrated parameter model is composed of a first static equivalent resistor, an equivalent inductance model and a first vibration system equivalent circuit structure which are mutually connected in series between two driving ends of the electroacoustic transducer;

The switching type nonlinear non-foster system is characterized by comprising a first capacitor, a first inductor and a single-phase full-bridge inverter connected to the driving end of the electroacoustic transducer; two switching tubes of the single-phase full-bridge inverter are provided with first common connecting ends, and the other two switching tubes are provided with second common connecting ends; two input ends of the single-phase full-bridge inverter are respectively and correspondingly electrically connected with two output ends of the direct-current power supply module; the first common connection end is electrically connected with one end of the first inductor, and the second common connection end, one end of the first capacitor and one driving end of the electroacoustic transducer are electrically connected with each other;

the other end of the first inductor, the other end of the first capacitor and one output end of the power amplifier are electrically connected with each other at a first electrical connection point, and the other output end of the power amplifier and the other driving end of the electroacoustic transducer are electrically connected with each other;

the bandwidth widening method of the electroacoustic transducer comprises the following steps:

the modulation degree m' and the modulation phase angle theta of the single-phase full-bridge inverter are calculated by using the following formula, wherein

m’＝m+Δm；

Wherein, the expression of the angular frequency ω is ω=2pi F _s,V_dc, which is the output voltage of the dc power supply module, and L _s、C_s is the inductance value of the first inductor and the capacitance value of the first capacitor, respectively; e _m、F_s、θ_m is the amplitude, frequency and phase angle of the output voltage signal of the power amplifier respectively;

In a first concentrated parameter model of the electroacoustic transducer, a first static equivalent resistance, an equivalent inductance model and a first vibration system equivalent circuit structure are mutually connected in series between two driving ends of the electroacoustic transducer; the equivalent inductance model is an eddy current impedance model formed by connecting a nonlinear inductance model and a nonlinear resistance model in series;

U _eq (omega) is the ratio of the reactance value of the compensated reactance model to the angular frequency omega, and Z _r(ω)、θ_r (omega) is the impedance model and the impedance angle corresponding to the impedance expression of the residual impedance model respectively;

The compensated reactance model and the residual impedance model are connected in series between the two driving ends of the electroacoustic transducer (10), so as to form an equivalent structure of the first concentrated parameter model;

Δm is the modulation feedback amount calculated according to jωu _eq(ω)-Z_{Equivalent means}; z _{Equivalent means} is the measured value of the equivalent impedance of the switching nonlinear foster system between the first electrical connection point and the second common connection end, and j represents an imaginary unit;

(A) The first centralized parameter model is equivalent to a structure formed by connecting an equivalent reactance model and an equivalent resistance model in series, and the compensated reactance model and the residual impedance model are respectively corresponding to the equivalent reactance model and the equivalent resistance model; x _eq (omega) is the reactance value of the equivalent reactance model;

(B) And U _eq(ω)＝L_em (omega), wherein the compensated reactance model is a nonlinear inductance model, and the first static equivalent resistance, the nonlinear resistance model and the first vibration system equivalent circuit structure form the residual impedance model.

Through the arrangement, the modulation degree m' of the single-phase full-bridge inverter can track the frequency of the output voltage signal of the power amplifier, so that the resonance bandwidth of the system can be expanded, the modulation degree can be adjusted along with the frequency of the output voltage signal of the power amplifier, and the equivalent reactance can be prevented from being counteracted only under a single frequency, so that the method is suitable for broadband matching. The modulation feedback quantity is calculated by subtracting the value of target matching from the value obtained by actual measurement, so that closed-loop control can be realized, and the control precision is higher. Through the arrangement, the invention is provided with the non-Forst impedance matching network, and negative impedance is generated to counteract the equivalent reactance of the transducer, namely only an equivalent resistance model is arranged in the model of the electroacoustic transducer after the cancellation, so that the transmission efficiency and quality of output signals of the electroacoustic transducer can be improved.

In the invention, the modulation degree m' and the modulation phase angle theta of the single-phase full-bridge inverter are regulated through the formula, so that the equivalent impedance of a circuit structure formed by the single-phase full-bridge inverter, the first capacitor and the first inductor between the first electric connection point and the second common connection end counteracts the reactance value corresponding to the equivalent reactance model in the centralized parameter model.

In the invention, the equivalent inductance model is an eddy current impedance model formed by connecting a nonlinear inductance model and a nonlinear resistance model in series, so that the influence of eddy current factors is considered in the static inductance model, and the model is more accurate.

In the invention, the modulation degree m' and the modulation phase angle theta _s of the single-phase full-bridge inverter are adjusted and regulated through the formula, so that the equivalent impedance of a circuit structure formed by the single-phase full-bridge inverter, the first capacitor and the first inductor between the first electric connection point and the second common connection end counteracts the impedance value corresponding to the static equivalent inductance in the centralized parameter model. I.e. such that the voltage across the static equivalent inductance is cancelled out from the voltage between the first electrical connection point and the second common connection. In this way, the output voltage of the power amplifier is provided for the equivalent circuit structure of the vibration system as much as possible under the condition of not considering other losses, so that the performance of the vibration system of the transducer is prevented from being influenced as much as possible.

In the present invention, only the nonlinear inductance model may be used as the reactance model to be compensated, or the first concentrated parameter model may be equivalent to a structure formed by connecting the equivalent reactance model and the equivalent resistance model in series, and then the equivalent reactance model may be used as the reactance model to be compensated. When the equivalent reactance model is used as a compensated reactance model, and when the reactance of the electric end and the mechanical end of the electroacoustic transducer is changed due to the change of working conditions or external disturbance, a good matching effect can be still realized on the nonlinear reactance of the electric end and the reactance of the mechanical end of the electroacoustic transducer, so that the influence of the nonlinear change of the reactance is reduced, and the requirement of a electroacoustic transmitting device for a larger bandwidth is met.

The technical scheme is as follows: the inductance value L _em (omega) of the nonlinear inductance model (51) and the resistance value R _em (omega) of the nonlinear resistance model (52) are expressed as follows:

Wherein θ (ω) is a first phase angle value, L _ex is a scaling factor, p is a first factor, q is a second factor, 0 < p < 1,0 < q < 1; wherein ,p(ω)＝a₀+a₁ω+a₂ω²,q(ω)＝a₃+a₄ω+a₅ω²,a₀、a₁、a₂、a₃、a₄、a₅ are coefficients.

The technical scheme is as follows: and applying an excitation signal to the driving end of the electroacoustic transducer to obtain an electric input admittance curve/electric input impedance curve of the electroacoustic transducer, and fitting according to the electric input admittance curve/electric input impedance curve to obtain a first concentrated parameter model of the electroacoustic transducer.

In the invention, the first concentrated parameter model is obtained according to the electric input admittance curve/electric input impedance curve of the actual electroacoustic transducer, so that the established model is consistent with the performance of the actual transducer, and the effect of generating negative impedance by the switch type nonlinear non-foster system is better, thereby better counteracting the equivalent reactance of the transducer.

The technical scheme is as follows:

The first concentrated parameter model of the electroacoustic transducer is obtained through the following steps (M1) and (M2):

(M1) obtaining a second concentrated parameter model of the electroacoustic transducer according to the electric input admittance curve/electric input impedance curve fitting, wherein the second concentrated parameter model of the electroacoustic transducer is composed of a second static equivalent resistor, a static equivalent inductor and a second vibration system equivalent circuit structure which are mutually connected in series between two driving ends of the electroacoustic transducer; the first vibration system equivalent circuit structure and the second vibration system equivalent circuit structure have the same structure; the first vibration system equivalent circuit structure consists of a first vibration system equivalent resistor, a first vibration system equivalent inductor and a first vibration system equivalent capacitor which are connected in parallel; the second vibration system equivalent circuit structure consists of a second vibration system equivalent resistor, a second vibration system equivalent inductor and a second vibration system equivalent capacitor which are connected in parallel; the resistance value of the equivalent resistance of the second vibration system, the inductance value of the equivalent inductance of the second vibration system, the capacitance value of the equivalent capacitance of the second vibration system and the resistance value of the second static equivalent resistance are respectively used as the resistance value of the equivalent resistance of the first vibration system, the inductance value of the equivalent inductance of the first vibration system, the capacitance value of the equivalent capacitance of the first vibration system and the resistance value of the first static equivalent resistance;

(M2) fitting a first coefficient p, a second coefficient q, a scaling coefficient L _ex:

Constructing a virtual three-dimensional coordinate system oxyz, wherein the x-axis, the y-axis and the z-axis of the virtual three-dimensional coordinate system respectively represent a fitting value of a first coefficient p, a fitting value of a second coefficient q and a fitting value of a proportional coefficient L _ex; searching coordinate points around the first coordinate point in an oxyz coordinate system by taking the first coordinate point as a starting point, taking a coordinate point which is close to the first coordinate point and a coordinate point which is far from the first coordinate point as a searching sequence until the fitting value of a first coefficient p, the fitting value of a second coefficient q, the fitting value of a proportional coefficient L _ex corresponding to the searched coordinate point, the resistance value of the equivalent resistance of the first vibration system, the inductance value of the equivalent inductance of the first vibration system, the capacitance value of the equivalent capacitance of the first vibration system and the fitting error of an admittance curve/impedance curve corresponding to a model formed by the resistance value of the equivalent capacitance of the first static state relative to an electric input admittance curve/electric input impedance curve of the electroacoustic transducer are not larger than a fitting error set value, and respectively corresponding the fitting value of the first coefficient p, the fitting value of the second coefficient q and the fitting value of the proportional coefficient L _ex corresponding to the searched coordinate point to the value of the first coefficient p, the value of the second coefficient q and the fitting value of the proportional coefficient L _ex, so that the first parameter model is obtained;

Wherein the first coordinate point is a coordinate point composed of p ₀、q₀、L_ex0; p ₀＝1,q₀＝0,L_ex0 is the inductance value of the static equivalent inductance.

The applicant found at the time of the study that if fitting starts from p=arbitrary value, q=arbitrary value, L _ex =arbitrary value, the fitting speed is slow and the convergence time is long. The model formed by the set of parameters p ₀＝1、q₀＝0、L_ex0 and the result of step (M1), i.e. the second concentration parameter model, can at least coincide with the admittance/impedance values of the electrical input admittance/impedance curve of the electroacoustic transducer (10) at the resonance point. That is, the coordinate points corresponding to the true values of p, q, and L _ex can be considered to be relatively close to the first coordinate point formed by p ₀＝1、q₀＝0、L_ex0. In practice, if the search is started from the first coordinate point and the order from the coordinate point closer to the first coordinate point to the coordinate point farther from the first coordinate point is taken as the search order, the search efficiency can be greatly improved, and compared with the method of taking the values of p, q and L _ex to start fitting, the search time can be greatly reduced.

In the above technical scheme, Z _{Equivalent means} isAnd/>Ratio of/(I)For the difference between the measured voltage at the other end of the first capacitor and the voltage at the other end of the first capacitor,/>The current is output for the measured power amplifier.

In the above technical solution, Δm is calculated by a proportional-integral adjustment method according to jX _eq(ω)-Z_{Equivalent means}.

In the invention, the PI controller is adopted to make the response fast, and the static difference-free adjustment can be realized.

In the above technical scheme, the inductance value L _s of the first inductor and the capacitance value C _s of the first capacitor satisfy the following formulas:

Wherein F _n is the resonant frequency of the LC filter formed by the first inductor and the first capacitor; f _p is the switching frequency of a switching tube of the single-phase full-bridge inverter; Δi _{ac_max} is 30% of the nominal output current effective value of the power amplifier.

In the above technical scheme, the direct current power supply module is connected with a second capacitor in parallel.

The invention also provides an electroacoustic transducer bandwidth widening device utilizing a switched nonlinear non-foster system, comprising a processor configured for performing the steps of the electroacoustic transducer bandwidth widening method as described in any of the above.

Compared with the prior art, the invention has the following advantages:

1. The invention uses the active matching network constructed by the nonlinear non-Foster control system to realize the broadband matching of high-power electroacoustic emission impedance, so that the electroacoustic transducer can work in a larger bandwidth range, and the power application level of the non-Foster control system is improved; the system carries out self-adaptive tracking on the parameters of a control system according to the mechanical dynamic impedance of the electroacoustic transmitting device, thereby realizing a nonlinear non-foster matching circuit;

2. The system has good matching effect on the change of impedance characteristics of the electroacoustic transmitting device, particularly on the characteristic dynamic mechanical impedance characteristics of electroacoustic transmission, can realize the matching of impedance in a broadband, and widens the application range of a non-foster circuit matching network;

3. The system performs real-time matching aiming at high-power electroacoustic emission in a working state, realizes on-line automatic impedance matching, has high matching precision and good continuity, expands the output bandwidth while ensuring high-power output, greatly improves the output efficiency of the electroacoustic emission device, and has practical theoretical value and engineering significance;

4. The invention adopts a feedforward control method, can effectively reduce detection and control quantity, improves convergence speed and response speed, reduces cost, improves system reliability, and has practical theoretical value and engineering significance.

Drawings

In order to more clearly illustrate the technical solutions of the embodiments of the present application, the drawings that are needed in the description of the embodiments will be briefly described below, it being obvious that the drawings in the following description are only some embodiments of the present application, and that other drawings may be obtained according to these drawings without inventive effort to a person skilled in the art.

FIG. 1 is a simplified diagram of a prior art focused parametric model of an electroacoustic transducer with an input impedance of inductive nature;

Fig. 2 is a schematic diagram of circuit connection between the bandwidth widening device of electroacoustic transducer, power amplifier and electroacoustic transducer in embodiments 1 and 2 of the present invention;

FIG. 3 is a schematic diagram of the principle analysis of embodiment 1 of the present invention;

FIG. 4 is a control block diagram of embodiment 1 of the present invention;

FIG. 5 is a schematic diagram of the method steps of example 1 of the present invention;

fig. 6 is a schematic diagram of reactance curves before and after impedance matching of the electroacoustic transducer according to embodiment 1 of the present invention;

FIG. 7 is a schematic diagram of the principle analysis of embodiment 2 of the present invention;

fig. 8 is a control block diagram of embodiment 2 of the present invention;

FIG. 9 is a schematic diagram of the method steps of example 2 of the present invention;

Fig. 10 is a schematic diagram of reactance curves before and after impedance matching of the electroacoustic transducer according to embodiment 2 of the present invention.

Detailed Description

For the purpose of making the objects, technical solutions and advantages of the embodiments of the present invention more apparent, the following description of the technical solutions of the present invention will be made by way of specific examples with reference to the accompanying drawings. It will be apparent that the described embodiments are some, but not all, embodiments of the invention. All other embodiments, which can be made by those skilled in the art based on the embodiments of the invention without making any inventive effort, are intended to be within the scope of the invention.

The existing impedance matching network can generally complete matching only on a single-point frequency, so that the broadband matching of the load cannot be realized, and the application uses a non-foster network to complete the broadband matching of the load. The existing non-foster network is composed of elements with limited output capacity such as operational amplifiers and the like, and can only realize impedance matching of milliwatt-level low-power loads such as micro antennas and the like. In the application, the output capability of the switch type nonlinear non-foster system is stronger, the impedance matching of a high-power sound source with the power of more than 100W such as a transducer can be completed, and the output signal frequency of the power amplifier 20 can be tracked, so that the static impedance matching of the electroacoustic transducer is realized in the whole output signal frequency range, and the bandwidth of the electroacoustic transducer is expanded.

When the input impedance of the electroacoustic transducer is inductive, a centralized parameter model of the electroacoustic transducer and a simplified diagram thereof are generally adopted as shown in fig. 1. The model is derived according to a motor analogy method and comprises a driving system equivalent circuit and a vibration system equivalent circuit, wherein the driving system equivalent circuit and the vibration system equivalent circuit are connected through a transformer which converts the ratio into an electromechanical conversion coefficient. The driving system equivalent circuit comprises a second static equivalent resistor 57 '(the resistance value is R _e) and a static equivalent inductor 56' (the inductance value is L _ex0) which are connected in series, and the vibration system equivalent circuit is formed by connecting a dynamic inductor L _m, a dynamic capacitor C _m, a dynamic resistor R ₁ and a load impedance Z _L in series. The inductive transducer concentration parameter model may also be simplified into a second concentration parameter model, i.e. a five-parameter element equivalent circuit composed of a second static equivalent resistor 57 '(with a resistance value of R _e, for example, a driving coil direct current resistor), a static equivalent inductor 56' (with an inductance value of L _ex0), a second vibration system equivalent resistor 53 '(with a resistance value of R _mes), a second vibration system equivalent inductor 54' (with an inductance value of L _mes), and a second vibration system equivalent capacitor (with a capacitance value of C _mes). Namely, a second vibration system equivalent resistor 53', a second vibration system equivalent inductor 54', and a second vibration system equivalent capacitor 55' constitute a second vibration system equivalent circuit structure V2.

Without the addition of a matching network, the equivalent inductance of the transducer would share the power provided by the power amplifier 20, resulting in a lower output power of the transducer at multiple frequencies. The scheme of the invention is that under the condition of maintaining the output voltage of the power amplifier 20 unchanged, the equivalent admittance (impedance) of the input position of the transducer is regulated in a wider frequency range by using a non-foster matching network, the output power of the transducer is improved, and the wide-frequency output of the transducer is realized.

Example 1

This embodiment 1 provides an electroacoustic transducer bandwidth widening device for high power broadband live acoustic emission using a switching type nonlinear non-foster system, which is provided between a power amplifier 20 and high power electroacoustic emission, comprising:

The inversion module comprises a voltage-adjustable power supply, an inverter, a filter and the like, and an output end is connected with the power amplifier 20 and the high-power electroacoustic emission in series; the device is used for outputting corresponding negative reactance voltage according to the PWM excitation pulse signal generated by the PWM modulation module, so as to realize broadband impedance matching with high-power electroacoustic emission;

the signal detection module is used for detecting the voltage and the output current of the two ends of the power amplifier 20 in real time; outputting the detected signal frequency, amplitude and phase information to a phase locking module and a PWM calculation module for calculation processing;

the phase locking module is used for carrying out real-time phase locking calculation on the detected voltage signals, realizing angle tracking control and finally outputting the angle tracking control to the PWM calculation module;

The PWM calculation module is used for calculating the modulation degree and the modulation phase angle of PWM and taking the modulation degree and the modulation phase angle as the input quantity of the self-adaptive correction module;

The self-adaptive correction module is used for calculating and logically judging a PWM modulation degree and a phase angle according to the mechanical dynamic impedance characteristic of the electroacoustic transmitting device, automatically adjusting the modulation degree and the phase angle in real time according to a judging result, and using the modulation degree and the phase angle as a feedforward target quantity to compensate the output phase and the amplitude of the inverter in real time by using the PI controller so as to achieve the optimal broadband matching effect of the high-power electroacoustic transmission; the self-adaptive correction module is finally output to the PWM modulation module;

the input end of the PWM modulation module is connected with the self-adaptive correction module, and the output end of the PWM modulation module is connected with the single-phase full-bridge inverter 30; for outputting suitable PWM pulses, and controlling the operation states of the internal components of the single-phase full-bridge inverter 3030.

The control panel is used for controlling the amplitude and the frequency of the output voltage of the power amplifier 20, realizing the monitoring of the waveform and the numerical value of the key parameters such as the current and the voltage after matching, and realizing the man-machine interaction.

The PI controller input is the difference between the theoretical compensation reactance and the actual compensation reactance. The modulation degree m' is controlled to vary by a controller. The PI controller parameters are mainly divided into a proportional coefficient Kp and an integral coefficient Ki, and are determined by combining theoretical calculation and actual condition tuning. In the actual process, the proportional coefficient and the integral coefficient can be obtained through debugging, for example, the proportional coefficient and the integral coefficient are connected into a control system in the experiment, and the proportional coefficient and the integral coefficient are determined according to the quality of an output waveform and the magnitude of a steady-state error. In the present invention, the proportional coefficient kp=10 and the integral coefficient ki=5 are preferable in the proportional-integral adjustment. The PI controller of the invention has quick response and can perform no static difference adjustment.

In an ideal state, the PWM waveform obtained according to the target modulation degree and the target modulation phase angle can meet the requirements, but in practice, the negative impedance may not completely counteract the static reactance of the transducer due to the error existing in the control of the single-phase full-bridge inverter 30, so that the single-phase full-bridge inverter 30 collects the output ac, and then feeds back to perform compensation calculation and PI control, and then calculates the required target modulation degree m'.

According to the electroacoustic transducer bandwidth widening device of the high-power broadband live acoustic emission, provided by the scheme, the amplitude and the frequency of the output voltage of the power amplifier 20 in the control panel can be given, the power amplifier 20 is connected, then the voltage and the output current at two ends of the power amplifier 20 are detected in real time by the signal detection module, then the detected voltage signal is subjected to real-time phase locking calculation by the phase locking module, then the modulation degree and the modulation phase angle of PWM are calculated preliminarily by the PWM calculation module according to the output data of the signal detection module and the phase locking module, then the adaptive correction module is operated according to the mechanical dynamic impedance characteristics of the electroacoustic emission device to automatically adjust the target modulation degree and the phase angle, and the PWM modulation module and the inversion module are used for outputting corresponding negative reactance under the working condition, so that broadband impedance matching with the high-power electroacoustic emission device is realized.

The PWM calculation module calculates the following steps:

Assuming that the nonlinear reactance of the high-power electroacoustic transducer to be matched is X _eq (omega), the residual impedance amplitude phasor of the electroacoustic transmitting device after the complete matching can be expressed as The power amplifier output current amplitude phasor is:

According to fig. 3, the equivalent inductance model 56 is a structure formed by connecting the nonlinear inductance model 51 and the nonlinear resistance model 52 in series with each other. The structure formed by the first static equivalent resistor 57, the equivalent inductance model 56 and the first vibration system equivalent circuit structure V1 connected in series is equivalent to the structure formed by the equivalent reactance model E1 and the equivalent resistance model E2 connected in series.

In the invention, the nonlinear impedance to be matched refers to the reactance value X _eq (omega) of the equivalent reactance model E1, and the residual impedance after matching is the resistance value R _eq (omega) of the equivalent resistance model E2.

The target negative reactance of the single-phase full-bridge inverter 30 is-X _eq (ω), and the output voltage of the single-phase full-bridge inverter 30 is:

assuming that the modulation degree of the PWM waveform is m, the voltage amplitude phasor of the AC side of the single-phase full-bridge inverter 30 is The ac side current amplitude phasor is/>The voltage on the dc side of the single-phase full-bridge inverter 30 is V _dc, and when the power switch tube bridge loss is not counted, the relationship between the ac and dc side voltages is:

according to kirchhoff's voltage law, the ac side loop of the single-phase full-bridge inverter 30 satisfies that the ac side voltage is equal to the sum of the voltage across the first inductor 2 (the inductance value is L _s) and the voltage across the first capacitor 1 (the capacitance value is C _s), that is:

meanwhile, the voltage across the first capacitor 1 is equal to the target output voltage of the single-phase full-bridge inverter 30:

The combined type (1), (2), (3), (4) and (5) can be obtained by solving:

The modulation degree of the single-phase full-bridge inverter 30 (i.e., the modulation degree of the output PWM waveform) is:

m’＝m+Δm；

Wherein:

the modulation phase angle of the single-phase full-bridge inverter 30 (i.e., the modulation phase angle of the output PWM waveform) is:

referring to fig. 2, fig. 2 is a schematic diagram of a system structure where high-power electroacoustic emission is located according to an embodiment of the invention. Wherein the power amplifier 20 provides voltage and current signals of variable amplitude and frequency for electroacoustic emissions. The bandwidth widening device of the electroacoustic transducer for high-power broadband live acoustic emission is connected in series between the electroacoustic transmitting device and the power amplifier 20.

The input impedance of the electroacoustic transducer 10 is inductive, the driving signal of the electroacoustic transducer 10 is provided by a power amplifier 20, and the first concentrated parameter model of the electroacoustic transducer 10 has a first vibration system equivalent circuit structure V1. The first and second concentrated parametric models are one parametric model and another parametric model of the electroacoustic transducer 10.

The switching type nonlinear non-foster system comprises a first capacitor 1, a first inductor 2 and a single-phase full-bridge inverter 30 connected to the driving end of the electroacoustic transducer 10; two switching tubes of the single-phase full-bridge inverter 30 have a first common connection terminal PA, and the other two switching tubes have a second common connection terminal PB; two input ends of the single-phase full-bridge inverter 30 are respectively and correspondingly electrically connected with two output ends of the direct-current power supply module 4; the first common connection terminal PA is electrically connected with one end of the first inductor 2, and the second common connection terminal PB, one end of the first capacitor 1, and one driving end of the electroacoustic transducer 10 are electrically connected with each other. The other end of the first inductor 2, the other end of the first capacitor 1 and one output end of the power amplifier 20 are electrically connected to the first electrical connection point P1, and the other output end of the power amplifier 20 and the other driving end of the electroacoustic transducer 10 are electrically connected to each other.

the modulation degree m' and the modulation phase angle theta of the single-phase full-bridge inverter (30) are calculated by the following formula, wherein

m’＝m+Δm；

Wherein, the expression of the angular frequency ω is ω=2pi F _s,V_dc, which is the output voltage of the dc power supply module 4, and L _s、C_s is the inductance value of the first inductor 2 and the capacitance value of the first capacitor 1, respectively; e _m、F_s、θ_m is the amplitude, frequency, phase angle of the output voltage signal of the power amplifier 20, respectively.

In the first concentrated parameter model of the electroacoustic transducer 10, a first static equivalent resistor 57, an equivalent inductance model 56 and a first vibration system equivalent circuit structure V1 are mutually connected in series between two driving ends of the electroacoustic transducer 10; the equivalent inductance model 56 is an eddy current impedance model formed by connecting the nonlinear inductance model 51 and the nonlinear resistance model 52 in series with each other.

The structure formed by the first static equivalent resistor 57, the equivalent inductance model 56 and the first vibration system equivalent circuit structure V1 connected in series is equivalent to the structure formed by the equivalent reactance model E1 and the equivalent resistance model E2 connected in series.

Z _r(ω)、θ_r (omega) is the impedance mode and the impedance angle corresponding to the impedance expression of the equivalent resistance model E2 respectively; x _eq (ω) is the reactance value of the equivalent reactance model E1. Δm is the modulation feedback amount calculated according to jX _eq(ω)-Z_{Equivalent means}; z _{Equivalent means} is an actual measurement value of the equivalent impedance of the switching nonlinear foster system between the first electrical connection point P1 and the second common connection point PB, and j represents an imaginary unit.

Fig. 6 is a graph of the matching effect of the nonlinear non-foster system of the present application, and after the matching is completed, the input reactance of the system is completely eliminated theoretically, so as to achieve the control objective. The reactance comprised by the vibration system is also part of the input reactance. In the application, the reactance of the mechanical end is matched through a non-foster system, so that complete impedance matching can be realized near the resonance point of the transducer, which cannot be realized in a mode of only matching static inductance (namely static equivalent inductance 56'). By the matching method, wider-band matching can be realized, and the maximum output power of the transducer is improved.

Referring to fig. 4, fig. 4 is a schematic structural diagram of an electroacoustic transducer bandwidth widening device of high-power broadband live acoustic emission according to the present invention. The amplitude and the frequency of the output voltage of the power amplifier 20 in a given control panel can be used for switching on the power amplifier 20, then the voltage and the output current at two ends of the power amplifier 20 are detected in real time by a signal detection module, then the detected voltage signal is subjected to real-time phase locking calculation by a phase locking module, then the modulation degree and the modulation phase angle of PWM are calculated preliminarily by a PWM calculation module according to the output data of the signal detection module and the phase locking module, then an adaptive correction module is operated according to the mechanical dynamic impedance characteristic of the electroacoustic transmitting device to automatically adjust the target modulation degree and the phase angle, and the PWM modulation module and an inversion module are used for outputting corresponding negative reactance under the working condition, so that the broadband impedance matching with the high-power electroacoustic transmitting device is realized.

Fig. 5 is a schematic step diagram of the bandwidth expansion method of the electroacoustic transducer of the present embodiment. When the active matching network constructed by the electroacoustic transducer bandwidth widening device based on high-power broadband live acoustic emission realizes broadband matching of high-power sound source impedance, the specific operation steps are as follows:

S1: before the high-power electroacoustic transmitting device is connected to work, the impedance curve of the device under specific power is required to be measured so as to acquire impedance information under the required working condition, and theoretical calculation and parameter identification are particularly carried out on the specific dynamic mechanical impedance characteristic of electroacoustic transmission;

In the step S1, the centralized parameter model established according to the high-power electroacoustic emission structure is also suitable for wideband electroacoustic emission with multi-mode design, and is different from the single-mode electroacoustic emission centralized parameter model in that the mechanical end is formed by connecting a plurality of mechanical capacitors, mechanical inductors and mechanical resistors in parallel, and the model parameters can be calculated and fitted according to the multi-resonance point experimental impedance curve theory.

The step S1 specifically includes:

s101: applying a voltage signal and a current signal to high-power electroacoustic emission according to specific power, and acquiring an electric input impedance curve of the high-power sound source under a high-signal condition;

S102: deducing a high-power electroacoustic emission concentration parameter model based on the coupling of multiple physical fields such as electroacoustic emission, electromechanical emission, acoustic emission and the like and the working principle of a system in the device;

In order to be able to describe electroacoustic emission impedance characteristics with a more accurate application of a focused parametric model, a nonlinear impedance model may be used instead of the elements in the model.

In the high-power emission concentration parameter model according to the embodiment of the present invention, the influence of the eddy current factor is considered, so in this embodiment, the static equivalent inductance 56' (the inductance value is L _ex0) in the driving system is replaced by an eddy current impedance model, that is, a nonlinear inductance model 51 (the inductance value is L _em (ω)) and a nonlinear resistance model 52 (the resistance value is R _em (ω)) are used to represent, where ω is the operating angular frequency.

When p=1, q=0, L _em(ω)＝L_ex0.L_em (ω) represents the static inductance as a function of frequency. p=1, q=0, L _em(ω)＝L_ex0 are a set of initial parameters extracted from the impedance curve without considering the eddy current loss, providing an algorithmic search initial for the subsequent power-of-eddy current loss characterization model. As can be seen from the power pattern expression, L _em(ω)＝L_ex0 when p=1, q=0. I.e. when p=1, q=0, L _em (ω) is a frequency invariant value, which is consistent with the linear model conditions, so that the linear result can be taken as a special point of the nonlinear model.

In the invention, the nonlinear phenomenon exists in the working process of the electroacoustic transducer, but the nonlinear phenomenon such as the vortex phenomenon exists in the actual working condition of the transducer, so that the analog adaptation of the linear equivalent circuit model to the transducer is poor. In the invention, the nonlinear inductance model 51 and the nonlinear resistance model 52 are used for replacing the static equivalent inductance 56' in the original model, so that nonlinear processing is realized, and the model is more close to the working condition of the transducer.

The first focused parametric model of the electroacoustic transducer 10 is obtained by the following steps (M1) and (M2):

(M1) obtaining a second concentrated parameter model of the electroacoustic transducer 10 according to the electric input admittance curve/electric input impedance curve fitting, wherein the second concentrated parameter model of the electroacoustic transducer 10 is composed of a second static equivalent resistor 57', a static equivalent inductor 56', and a second vibration system equivalent circuit V2 structure which are mutually connected in series between two driving ends of the electroacoustic transducer 10; the first vibration system equivalent circuit structure V1 and the second vibration system equivalent circuit structure V2 have the same structure; the first vibration system equivalent circuit structure V1 consists of a first vibration system equivalent resistor 53, a first vibration system equivalent inductor 54 and a first vibration system equivalent capacitor 55 which are mutually connected in parallel; the second vibration system equivalent circuit structure V2 is composed of a second vibration system equivalent resistor 53', a second vibration system equivalent inductor 54', and a second vibration system equivalent capacitor 55' which are connected in parallel; the resistance value of the second vibration system equivalent resistor 53', the inductance value of the second vibration system equivalent inductor 54', the capacitance value of the second vibration system equivalent capacitor 55', and the resistance value of the second static equivalent resistor 57' are respectively used as the resistance value of the first vibration system equivalent resistor 53, the inductance value of the first vibration system equivalent inductor 54, the capacitance value of the first vibration system equivalent capacitor 55, and the resistance value of the first static equivalent resistor 57;

Constructing a virtual three-dimensional coordinate system oxyz, wherein the x-axis, the y-axis and the z-axis of the virtual three-dimensional coordinate system respectively represent a fitting value of a first coefficient p, a fitting value of a second coefficient q and a fitting value of a proportional coefficient L _ex; searching coordinate points around the first coordinate point by taking the first coordinate point as a starting point in an oxyz coordinate system, taking a coordinate point which is close to the first coordinate point to a coordinate point which is far from the first coordinate point as a searching sequence until a fitting value of a first coefficient p, a fitting value of a second coefficient q and a fitting value of a proportional coefficient L _ex corresponding to the searched coordinate point are matched with a model formed by the resistance value of the first vibration system equivalent resistor 53, the inductance value of the first vibration system equivalent inductor 54, the capacitance value of the first vibration system equivalent capacitor 55 and the resistance value of the first static equivalent resistor 57 obtained in the step (M1) (namely, obtaining a nonlinear inductor model 51 and a nonlinear resistor model 52 according to the fitting value of p, the fitting value of q and the fitting value of L _ex, combining the resistance value of the first vibration system equivalent resistor 53, the inductance value of the first vibration system equivalent inductor 54, the capacitance value of the first vibration system equivalent capacitor 55 and the resistance value of the first static equivalent resistor 57 obtained in the step (M1), wherein the fitting error of the admittance curve/impedance curve corresponding to the electric input admittance curve/electric input impedance curve of the electroacoustic transducer 10 is not larger than the fitting error set value, and the fitting value of the first coefficient p, the fitting value of the second coefficient q and the fitting value of the proportionality coefficient L _ex corresponding to the searched coordinate point are respectively used as the value of the first coefficient p, the value of the second coefficient q and the value of the proportionality coefficient L _ex, so that a first concentrated parameter model of the electroacoustic transducer 10 is obtained;

Wherein the first coordinate point is a coordinate point composed of p ₀、q₀、L_ex0; p ₀＝1,q₀＝0,L_ex0 is the inductance value of the static equivalent inductance 56'.

In the present application, in step (M1), the resistance of the equivalent resistor 53 of the first vibration system, the inductance of the equivalent inductor 54 of the first vibration system, the capacitance of the equivalent capacitor 55 of the first vibration system, and the resistance of the equivalent resistor 57 of the first static state are obtained first, that is, according to the values of the parameters of the equivalent circuit structure of the second vibration system and the resistance of the equivalent resistor of the first static state in the second centralized parameter model, the values of the parameters of the equivalent circuit structure of the first vibration system and the resistance of the equivalent resistor of the first static state in the first centralized parameter model. In step (M2), the fitting value of the first coefficient p, the fitting value of the second coefficient q, and the fitting value of the scaling factor L _ex are obtained, that is, the expressions of the nonlinear inductance model 51 and the nonlinear resistance model 52 can be obtained according to step (M2). Combining the step (M1) and the step (M2) to obtain a first concentrated parameter model.

S103: firstly, primarily calculating the parameter value of an impedance element of a lumped parameter model of the device according to the electroacoustic emission resonance point frequency and a corresponding impedance theoretical calculation method thereof;

In the step S103, the theoretical calculation method of the electroacoustic emission resonant point frequency and the corresponding impedance thereof mainly includes:

(1) When the working frequency F _s =0 Hz, the starting point of the impedance curve amplitude is the high-power electroacoustic emission driving dc resistor R _e (i.e. the resistance of the first static equivalent resistor 57).

(2) When high-power electroacoustic emission generates series resonance, resonance frequencyOnly the static equivalent inductance 56' (inductance value L _ex0) remains in the circuit, i.e. when F _s＝f_s corresponds to reactance value ω _sL_ex0.

(3) When parallel resonance occurs in high-power electroacoustic emission, the resonance frequencyAt this time, the theoretical transmitting reactance value is zero, and only the first static equivalent resistor 57 (the resistance value is R _e) and the first vibration system equivalent resistor 53 (the resistance value is R _mes) remain in the circuit, that is, when the resistance value corresponding to F _s＝f_p is R _e+R_mes.

The initial values of the element parameters in the parameter model in the transmitting device set can be obtained through a theoretical calculation method, and the initial values are shown in the following table.

TABLE 1

Component parameters	R_e/Ω	R_mes/Ω	L_mes/mH	C_mes/mF	L_ex0/mH
						Fitting parameters	7.4	34	32.6	0.318	4.2

S104: and (3) taking the element impedance parameter value theoretically calculated in the step (S103) as an initial parameter of a centralized parameter model identification algorithm, and fitting the element parameter value in the model according to an electric input impedance experimental curve.

In the process of fitting model element parameters, the eddy current impedance model has practical physical significance, and the change trend of the eddy current impedance along with frequency can be accurately simulated. Therefore, in the embodiment, the power finger model is selected to fit the eddy current impedance model in the high-power electroacoustic emission model, and the expression of the power finger model is as follows:

The fitting of L _ex, p, q is also obtained according to the above method, i.e. the working frequency is different, the corresponding impedance is different (e.g. fs=0 Hz, the impedance is Re), i.e. the model parameters are obtained according to the established mathematical model by the impedance values at different frequencies.

In practical calculations, the number of combinations of p, q, L _ex is very large. p=1, q=0, and L _ex =4.2 are one group of parameters that fit to the impedance value at the resonance point, but the area after the resonance point is poorly effective. In practice, therefore, a parameter trial can be performed at this point in a range (p, q, L _ex near the set of values) to finally obtain a satisfactory set of parameters p=0.7483, q=0.0398, L _ex =43.9 (the fitting error is smaller than the set fitting error). If fitting is performed from p=arbitrary value, q=arbitrary value, and L _ex =arbitrary value, the fitting speed is slow and the convergence time is long. For example, when the values of the parameters p, q, and L _ex are solved by using the least square method, when the least square method solves that the error of the impedance curve fitted under the parameters is within the set error compared with the actual curve, the parameters are derived.

Preferably, in the step S104, a fitting algorithm such as a least squares algorithm or a particle swarm algorithm may be selected to fit the electrical input impedance curve, but the fitting algorithm is not limited to these two algorithms.

And fitting the model element parameter values according to the electric input impedance curve through a fitting algorithm, and when the fitting error is smaller than a set value, the element fitting parameters are shown in the following table.

TABLE 2

S2: when the high-power electroacoustic emission works, the power amplifier 20 can be switched on, and the amplitude and frequency of the output voltage of the power amplifier 20 and the voltage of the voltage-adjustable power supply (for example, the voltage can be set through a control panel) can be set;

S3: the bandwidth widening device of the electroacoustic transducer starts to work, realizes impedance matching among the power amplifier, the high-power electroacoustic emission and the control system by utilizing the conversion of output negative reactance, and ensures the maximum active power output of the device all the time;

Referring to fig. 3, fig. 3 is a schematic diagram of an active matching network where the bandwidth widening device of the electroacoustic transducer of the present example is located. Inverter port output voltage Voltage/>, at two ends of equivalent reactance model E1 (reactance value is X _eq (omega)) to be matchedThe relation between the two is: /(I)Current flowing into an inverter from an inverter port/>The currents flowing through the equivalent reactance models E1 to be matched are all/>Due to inverter output impedance/>The impedance value corresponding to the reactance to be matched is calculated by the voltage V _L and the current I _L at the two ends of the equivalent inductance, namelyZ _non＝-Z_eq (ω) can be obtained to achieve a perfect match to the high-power electroacoustic emission impedance.

The step S3 specifically includes:

S301: according to element parameter values in the electroacoustic emission lumped parameter model, giving a negative reactance value required to be simulated by an output port of the inverter;

s302: by controlling the PWM phase and modulation degree, the output voltage of the inverter is kept to be negative reactance voltage, so that broadband matching of transmitting impedance is realized.

S303: the voltage signal of the power amplifier 20 is detected in real time by a signal detection module to obtain the output voltage amplitude of the power amplifier with the amplitude of E _m, the frequency of F _s and the initial phase angle of theta _m, namely the output voltage amplitude phasor of the power amplifier

S304: the phase-locked module is used for carrying out phase-locked calculation on the detected voltage signals, tracking control is carried out on angles, and finally the angles are output to the PWM calculation module;

s305: calculating a modulation degree and a modulation phase angle of PWM by using a PWM calculation module according to the impedance characteristic of the electroacoustic transmitting device to obtain a modulation degree m and a modulation phase angle theta;

s306: and inputting the calculation result into the self-adaptive correction module, logically distinguishing the input quantity according to the mechanical dynamic impedance characteristic of the electroacoustic transmitting device, and automatically adjusting the modulation degree and the phase angle in real time according to the distinguishing result.

When m is more than 0, controlling the adaptive control module to output a modulation degree and a phase angle unchanged; when m < 0, the control output modulation degree is output as the absolute value of m, and the modulation phase angle is increased by pi.

Because the existing controller generally cannot output a negative value, i.e., the modulation degree cannot output a negative value. In the invention, when m is more than 0, the modulation degree and phase angle of the output can be controlled to be unchanged. When m < 0, the modulation degree output can be made to be the absolute value (i.e. positive value) of m, and the modulation phase angle output can be made to be theta+pi, so that the signal output is facilitated.

S307: and the PWM modulation module outputs proper PWM pulse according to the calculation and control results in the step S304, the step S305 and the step S306, controls the working state of the nonlinear non-Foster matching system, and keeps the output voltage of the inverter to be negative reactance voltage, thereby realizing the on-line broadband matching of high-power electroacoustic emission impedance.

Preferably, in order to filter out higher harmonics around the switching frequency, the inverter module adopts the principle shown in the formula (6) and the formula (7) to select an LC filter (i.e. a filter formed by the first capacitor 1 and the first inductor 2):

Wherein F _n is the resonant frequency of the LC filter; f _p is the carrier frequency of PWM; Δi _{ac_max} is 30% of the effective value of the rated current output by the power amplifier, in this embodiment, the inductance value L _s of the first inductor 2 may be 2mH; the capacitance value C _s of the first capacitor 1 is 10 μf.

S4: when the working condition of high-power electroacoustic emission is changed (for example, the high-power electroacoustic emission can be adjusted through a control panel), the step S3 is repeated, and self-adaptive tracking is carried out according to the mechanical dynamic impedance of the electroacoustic emission device, so that the control system always outputs corresponding negative reactance voltage, and the maximum active power output of the device is ensured all the time.

Referring to fig. 6, fig. 6 is a schematic diagram of reactance curves before and after impedance matching for high-power electroacoustic emission in an embodiment of the present application. The solid line (-) represents the output reactance curve of the power amplifier before matching, and the graph shows that the output reactance of the power amplifier is rapidly increased along with the increase of the frequency, so that the broadband output of high-power transmission is severely limited; the dashed line (-) represents the output reactance curve of the switch-type nonlinear non-foster system, so that the influence of mechanical reactance is completely eliminated on the basis of eliminating static reactance of the non-foster system, and the output target of the system is realized; in the switch-type nonlinear non-foster system, the output reactance curve between the P1 point and the negative output end of the power amplifier 20 is shown as a broken line (-s) in fig. 6, and under the condition of completely eliminating the static reactance and the dynamic reactance of the emission, the output reactance of the power amplifier can be kept to be zero all the time, so that the output efficiency of the emission is improved, and the output bandwidth of the high-power emission is expanded.

In this embodiment 1, the nonlinear non-foster system matches both the static reactance (i.e., the reactance portion in the equivalent inductance model) and the mechanical reactance (i.e., the reactance portion in the equivalent circuit structure of the first vibration system), corresponding to the result of fig. 6.

The invention provides a bandwidth widening device of an electroacoustic transducer of high-power broadband live acoustic emission and an implementation method thereof.

The nonlinear non-foster system can theoretically realize infinite bandwidth, and breaks through the limitation of gain-bandwidth theory on high-power electroacoustic emission bandwidth; compared with the power capability of a common non-Forster system based on an operational amplifier, the system can be improved by more than 1000 times, and the power application level of a non-Forster circuit matching network is improved; the system has good matching effect on the change of the electroacoustic emission impedance characteristics under different frequencies, particularly on the characteristic of the dynamic mechanical impedance of electroacoustic emission, can realize the complete matching of the impedance in a broadband, and widens the application range of a non-foster circuit matching network; the system performs real-time matching aiming at high-power electroacoustic emission in a working state, has high matching precision and good continuity, expands output bandwidth while ensuring high-power output, greatly improves the output efficiency of the electroacoustic emission device, and has practical theoretical value and engineering significance.

In example 1, the reactance model and the residual impedance model to be compensated correspond to the equivalent reactance model (E1) and the equivalent resistance model (E2), respectively

Example 2

In embodiment 2, U _eq(ω)＝L_em (ω), the compensated reactance model is a nonlinear inductance model 51, and the first static equivalent resistance 57, the nonlinear resistance model 52 and the first vibration system equivalent circuit structure V1 form the residual impedance model. The first concentrated parameter model of this embodiment 2 is the same as that of embodiment 1, and is composed of a first static equivalent resistor 57, a nonlinear inductance model 51, a nonlinear resistance model 52, and a first vibration system equivalent circuit structure V1, which are connected in series with each other between the two driving ends of the electroacoustic transducer 10.

This embodiment 2 differs from embodiment 1 mainly in the calculation of m, the formula, and the calculation of L _em(ω)、R_em (ω), and other portions can be referred to embodiment 1.

Fig. 8 is a control block diagram of embodiment 2 of the present invention. The transducer bandwidth widening device comprises an equivalent circuit model building module, a model identification module, an inversion module, a signal detection module, a phase locking module, a PWM calculation module and a PWM modulation module. The equivalent circuit model construction module derives a nonlinear equivalent circuit model of the high-power transducer based on the electric-mechanical-acoustic multi-physical-field coupling of the electroacoustic transducer and the working principle of the internal system of the transducer and the nonlinear impedance model; the model identification module is based on a transducer nonlinear model, and determines parameters of each element of the nonlinear model by adopting methods such as theoretical calculation, parameter identification and the like according to high-power impedance test experimental data; the input end of the inversion module is connected with a direct-current voltage source, and the output end of the inversion module is connected with the power amplifier and the high-power sound source in series; the signal detection module is used for detecting the voltage and the output current at two ends of the power amplifier; the phase locking module is used for carrying out phase locking calculation on the detected signals, carrying out angle tracking control and finally outputting the signals to the PWM calculation module; the PWM calculation module is used for calculating a target modulation degree and a phase angle of PWM according to the identification parameter result of the model identification module, using the target modulation degree and the phase angle as feedforward quantity and using a PI controller to compensate the output phase and the amplitude of the inverter, and finally outputting the output phase and the amplitude to the PWM modulation module; the input end of the PWM modulation module is connected with the PWM calculation module, and the output end of the PWM modulation module is connected with the inverter module and is used for outputting proper PWM pulses and controlling the working state of internal elements of the inverter.

Fig. 9 is a schematic diagram of specific operation steps when the active matching network of the transducer bandwidth widening device of the nonlinear transducer of the high-power electroacoustic transducer is used for implementing the broadband matching of the nonlinear impedance of the high-power sound source, and the specific steps are as follows:

s1: applying high-power voltage signals and current signals to the electroacoustic transducer under different working conditions, so as to obtain an electric input impedance curve of the transducer;

s1, applying high-power excitation signals to a transducer under different frequencies, and obtaining impedance amplitude values and phase angles of the transducer under each frequency to obtain a relation curve of transducer resistance and reactance with respect to the frequency;

s2: building a high-power nonlinear equivalent circuit model of the electroacoustic transducer through a system equivalent circuit model building module in the technical scheme 1, obtaining a total impedance expression based on frequency-dependent mathematical models of various impedances in the nonlinear impedance model, and determining parameter values of elements in the nonlinear model through a parameter identification module according to impedance test data in the step S1;

s201: based on the electroacoustic transducer electric-mechanical-acoustic multi-physical field analogy method and the working principle of the internal system of the transducer, a linear centralized parameter model of the high-power electroacoustic transducer is built;

s202: according to the resonant point frequency of the electroacoustic transducer and the corresponding impedance theoretical calculation method, the parameters of the transducer linear concentrated parameter model element are calculated preliminarily;

the element parameters in the transducer linear model can be obtained through a theoretical calculation method, and the element parameters are shown in the following table.

TABLE 3 fitting values of parameters of the parametric model elements in the second set

Component parameters	R_e/Ω	R_mes/Ω	L_mes/mH	C_mes/mF	L_ex0/mH
						Fitting value	7.4	34	32.6	0.318	4.2

S203: based on the multi-physical field coupling of electroacoustic transducer such as electro-mechanical-acoustic and the working principle of the system in the transducer, the high-power nonlinear equivalent circuit model of the electroacoustic transducer is deduced by combining the influence of the vortex nonlinearity generated by high-frequency operation on the impedance characteristic of the transducer and the frequency-dependent mathematical models of various impedances in the nonlinear impedance model;

The impact of electroacoustic transducer nonlinearity on transducer impedance characteristics includes: under the condition of large signal excitation, under the high-frequency condition, various parameters and conversion coefficients of active components in the transducer can be greatly changed, so that the impedance parameters in the transducer centralized parameter model are changed.

The high-power nonlinear equivalent circuit model of the electroacoustic transducer is compared with the linear model in the step S201, and the linear elements in the equivalent circuit of the driving system and the vibration system are replaced by nonlinear impedance models.

The frequency-dependent mathematical model which considers the nonlinearity of the eddy current generated at high frequency selects an eddy current impedance model which combines a power exponent model and a polynomial model. The model has practical physical significance, and can accurately simulate the variation trend of the eddy current impedance along with the frequency as follows:

Wherein p (ω) =a ₀+a₁ω+a₂ω²,q(ω)＝a₃+a₄ω+a₅ω².

In calculating the value of L _ex、a₀、a₁、a₂、a₃、a₄、a₅, a method similar to that of calculation of L _ex, p, q in example 1 may be adopted, that is, each value of L _ex、a₀、a₁、a₂、a₃、a₄、a₅ that can fit the impedance at the upper resonance point is found first (as a search initial value), then the search initial value is used as a starting point, the search range is gradually expanded around until the error of the impedance curve fitted by each obtained parameter reaches within the set error compared with the actual curve, that is, the value of each parameter L _ex、a₀、a₁、a₂、a₃、a₄、a₅ is derived,

S204: and (3) taking the element parameter values in the step S202 as the initialization parameters of the nonlinear model of the transducer in the step S203, and determining the element parameter values in the nonlinear model through an identification algorithm according to the frequency-dependent electric input impedance experimental curve measured in the step S1. In the example, the nonlinear model related parameters are obtained through least square fitting as follows:

TABLE 4 first set of parametric model element parameter fits

Aiming at possible factors which can cause the transducer to work to present nonlinearity in the working process, the specific method for determining the parameters of each element in the nonlinear impedance model through an identification algorithm according to an electric input impedance experimental curve is as follows: when the transducer is operated under high-frequency excitation conditions, vortex nonlinearity can occur, and nonlinear changes can be generated in impedance parameters in the transducer concentration parameter model. And (2) obtaining a relation curve of transducer resistance and reactance with respect to frequency according to the transducer impedance amplitude and phase angle under different frequencies measured in the step (S1), obtaining a total impedance expression based on a frequency-dependent mathematical model of various impedances in the nonlinear impedance model, and obtaining various impedance parameters in the nonlinear impedance model when the transducer works at high frequency according to fitting of the expression and the measured impedance curve.

The related mathematical model of each factor can be selected from a polynomial model, a power exponent model, a logarithmic function model and the like to characterize each impedance characteristic, but the method is not limited to the models.

In step S2, a centralized parameter model is built according to the high-power electroacoustic transducer structure, and the centralized parameter model is also suitable for a wideband electroacoustic transducer with multi-mode design, and is different from a single-mode electroacoustic transducer centralized parameter model in that a mechanical end of the centralized parameter model is formed by connecting a plurality of mechanical capacitors, mechanical inductors and mechanical resistors in parallel, and model parameters of the centralized parameter model can be calculated and fitted according to a multi-resonance point experimental impedance curve theory.

S3: according to the nonlinear reactance in the nonlinear equivalent circuit model in the step S2, giving a negative reactance value required to be simulated by an output port of the non-foster system;

S4: by operating each module of the non-foster system in the technical scheme 1 to obtain PWM phase and modulation degree under corresponding working conditions, the output voltage of the system is kept to be negative reactance voltage, so that the broadband matching of nonlinear impedance of the transducer is realized.

S401: detecting the voltage signal of the power amplifier by using a signal detection module to obtain the amplitude of the output voltage of the power amplifier with the amplitude of E _m, the frequency of F _s and the initial phase angle of theta _m, namely the amplitude phasor of the output voltage of the power amplifier

S402: the phase-locked module is used for carrying out phase-locked calculation on the detected voltage signals, tracking control is carried out on angles, and finally the angles are output to the PWM calculation module;

s403: the PWM calculation module is used for calculating a target modulation degree and a phase angle of PWM according to the identification parameter result of the model identification module, using the target modulation degree and the phase angle as feedforward quantity and using a PI controller to compensate the output phase and the amplitude of the inverter, and outputting the output phase and the amplitude to the PWM modulation module;

The PWM calculation module calculates the following steps:

according to fig. 7, the equivalent inductance model 56 is a structure in which the nonlinear inductance model 51 and the nonlinear resistance model 52 are connected in series with each other. The structure formed by the first static equivalent resistor 57, the equivalent inductance model 56 and the first vibration system equivalent circuit structure V1 connected in series is equivalent to the structure formed by the equivalent reactance model E1 and the equivalent resistance model E2 connected in series.

In the invention, the nonlinear impedance to be matched refers to the reactance value L _em (omega) of the equivalent reactance model E1, and the amplitude phasor of the residual impedance after the matching can be expressed as

The target negative reactance of the single-phase full-bridge inverter 30 is-L _em (ω), and the output voltage of the single-phase full-bridge inverter 30 is:

The combined type (1), (2), (3), (4) and (5) can be obtained by solving:

Namely:

PWM target modulation degree: m' =m+Δm; wherein,

PWM target modulation phase angle:

s404: and the PWM modulation module outputs proper PWM pulse according to the calculation and control results in the step S402 and the step S403, controls the working state of the switching type non-Forst feedforward control system, and keeps the output voltage of the inverter to be negative reactance voltage, thereby realizing the matching of nonlinear impedance of the high-power electroacoustic transducer.

Fig. 7 is a schematic diagram of an active matching network where a nonlinear transducer bandwidth widening device of a high-power electroacoustic transducer is located in the present embodiment. The relationship between the inverter port output voltage V _non and the voltage V _L across the nonlinear reactance L _em (ω) is: v _non＝-V_L; the relationship between the current I _non flowing from the inverter port into the inverter and the current I _L flowing through the nonlinear reactance L _em (ω) is: i _non＝I_L; because of the inverter output impedance Z _non＝V_non/I_non, the nonlinear reactance Z _L＝V_L/I_L, Z _non＝-Z_L can be obtained, thereby achieving broadband matching of the transducer nonlinear impedance.

The invention provides a transducer bandwidth widening device for considering nonlinearity of a high-power electroacoustic transducer and an implementation method thereof. By constructing a nonlinear equivalent circuit model under the high-power condition of the electroacoustic transducer, the system can track the nonlinear impedance of the transducer more accurately and rapidly and provide negative reactance, so that the broadband matching of the high-power electroacoustic transducer is realized; the system can theoretically realize infinite bandwidth, and breaks through the limitation of gain-bandwidth theory on the bandwidth of the high-power electroacoustic transducer; compared with the common non-Forster system based on an operational amplifier, the system has the advantages that the power capacity can be improved by more than 1000 times, the power application level of a non-Forster circuit matching network is improved, and the application range of the non-Forster system is widened; the invention expands the output bandwidth while ensuring the high-power output of the transducer, greatly improves the output efficiency of the electroacoustic transducer, and has practical theoretical value, engineering significance and wide applicability.

Referring to fig. 10, fig. 10 is a control effect diagram of the transducer bandwidth widening method of the present invention. The solid line (-) represents the output reactance curve of the power amplifier before matching, and the graph shows that the output reactance of the power amplifier is rapidly increased along with the increase of the frequency, so that the broadband output of the high-power transducer is severely limited; the dashed line (-) indicates an output reactance curve of the power amplifier after matching of an active matching network composed of a transducer bandwidth widening device based on a high-power electroacoustic transducer linear model, and as can be seen from fig. 10, although the matching effect of the linear system is obvious, static inductance is not completely matched due to the influence of nonlinear factors such as eddy current effect; the dashed line (-) represents the output reactance curve of a switch-type nonlinear non-foster system introduced in the electroacoustic transducer system, and the negative reactance of the system output is not in a linear relation with the frequency, so that the output target of the system is realized; after the control system is introduced, that is, the output reactance (namely, the reactance between the P1 point and the negative output end of the power amplifier 20) of the power amplifier is shown as a graph dotted line (-), the output reactance of the power amplifier can be kept close to zero under the high-frequency condition under the condition of completely eliminating the nonlinear static reactance of the transducer, so that the output efficiency of the transducer is improved, and the output bandwidth of the high-power transducer is expanded.

In this embodiment 2, the nonlinear foster system matches the static reactance in fig. 7 (i.e., the reactance portion in the equivalent inductance model of the element 51 shown in fig. 7), which is an equivalent circuit considering the nonlinear eddy current effect, and corresponds to the result of fig. 10.

It should be noted that, in the present specification, each embodiment is described in a progressive manner, and each embodiment is mainly described as different from other embodiments, and identical and similar parts between the embodiments are all enough to be referred to each other.

The foregoing describes the embodiments of the present application in detail, but the description is only a preferred embodiment of the present application and should not be construed as limiting the scope of the application. All equivalent changes and modifications within the scope of the present application are intended to be covered by this patent. Modifications of the application which are equivalent to various embodiments of the application will occur to those skilled in the art upon reading the application, and are within the scope of the application as defined in the appended claims. Embodiments of the application and features of the embodiments may be combined with each other without conflict.

Claims

1. An electroacoustic transducer bandwidth widening method utilizing a switching type nonlinear non-foster system, wherein the input impedance of the electroacoustic transducer (10) is inductive, a driving signal of the electroacoustic transducer (10) is provided by a power amplifier (20), the electroacoustic transducer (10) is provided with a first centralized parameter model, and the first centralized parameter model is composed of a first static equivalent resistor (57), an equivalent inductance model (56) and a first vibration system equivalent circuit structure (V1) which are mutually connected in series between two driving ends of the electroacoustic transducer (10);

The switching type nonlinear non-foster system is characterized by comprising a first capacitor (1), a first inductor (2) and a single-phase full-bridge inverter (30) connected to the driving end of the electroacoustic transducer (10); two switching tubes of the single-phase full-bridge inverter (30) are provided with a first common connection end (PA), and the other two switching tubes are provided with a second common connection end (PB); two input ends of the single-phase full-bridge inverter (30) are respectively and correspondingly electrically connected with two output ends of the direct-current power supply module (4); the first public connection end (PA) is electrically connected with one end of the first inductor (2), and the second public connection end (PB), one end of the first capacitor (1) and one driving end of the electroacoustic transducer (10) are electrically connected with each other; the other end of the first inductor (2), the other end of the first capacitor (1) and one output end of the power amplifier (20) are electrically connected to a first electrical connection point (P1), and the other output end of the power amplifier (20) and the other driving end of the electroacoustic transducer (10) are electrically connected to each other;

m’＝m+Δm；

Wherein, the expression of the angular frequency ω is ω=2pi F _s,V_dc, which is the output voltage of the dc power supply module (4), and L _s、C_s is the inductance value of the first inductor (2) and the capacitance value of the first capacitor (1), respectively; e _m、F_s、θ_m is the amplitude, frequency and phase angle of the output voltage signal of the power amplifier (20);

The equivalent inductance model (56) is an eddy current impedance model formed by connecting a nonlinear inductance model (51) and a nonlinear resistance model (52) in series;

Δm is the modulation feedback amount calculated according to jωu _eq(ω)-Z_{Equivalent means}; z _{Equivalent means} is the measured value of the equivalent impedance of the switching nonlinear Foster system between the first electrical connection point (P1) and the second common connection Point (PB), j represents an imaginary unit;

(A) The first centralized parameter model is equivalent to a structure formed by mutually connecting an equivalent reactance model (E1) and an equivalent resistance model (E2) in series, and the compensated reactance model and the residual impedance model are respectively corresponding to the equivalent reactance model (E1) and the equivalent resistance model (E2); X _eq (omega) is the reactance value of the equivalent reactance model (E1);

(B) U _eq(ω)＝L_em (omega), wherein the compensated reactance model is a nonlinear inductance model (51), and the first static equivalent resistance (57), the nonlinear resistance model (52) and the first vibration system equivalent circuit structure (V1) form the residual impedance model.

2. The electroacoustic transducer bandwidth widening method according to claim 1, wherein: the inductance value L _em (omega) of the nonlinear inductance model (51) and the resistance value R _em (omega) of the nonlinear resistance model (52) are expressed as follows:

Or/>

3. The electroacoustic transducer bandwidth widening method according to claim 2, characterized in that: and obtaining an electric input admittance curve/electric input impedance curve of the electroacoustic transducer (10) by applying an excitation signal to the driving end of the electroacoustic transducer (10), and obtaining a first concentrated parameter model of the electroacoustic transducer (10) according to electric input admittance curve/electric input impedance curve fitting.

4. A bandwidth widening method for electroacoustic transducers according to claim 3, characterized in that:

The first concentrated parameter model of the electroacoustic transducer (10) is obtained by the following steps (M1) and (M2):

(M1) obtaining a second concentrated parameter model of the electroacoustic transducer (10) according to the electric input admittance curve/electric input impedance curve fitting, wherein the second concentrated parameter model of the electroacoustic transducer (10) is composed of a second static equivalent resistor (57 '), a static equivalent inductor (56') and a second vibration system equivalent circuit (V2) which are mutually connected in series between two driving ends of the electroacoustic transducer (10); the first vibration system equivalent circuit structure (V1) and the second vibration system equivalent circuit structure (V2) have the same structure; the first vibration system equivalent circuit structure (V1) consists of a first vibration system equivalent resistor (53), a first vibration system equivalent inductor (54) and a first vibration system equivalent capacitor (55) which are connected in parallel; the second vibration system equivalent circuit structure (V2) consists of a second vibration system equivalent resistor (53 '), a second vibration system equivalent inductor (54 ') and a second vibration system equivalent capacitor (55 ') which are mutually connected in parallel; the resistance value of the second vibration system equivalent resistor (53 '), the inductance value of the second vibration system equivalent inductor (54'), the capacitance value of the second vibration system equivalent capacitor (55 ') and the resistance value of the second static equivalent resistor (57') are respectively used as the resistance value of the first vibration system equivalent resistor (53), the inductance value of the first vibration system equivalent inductor (54), the capacitance value of the first vibration system equivalent capacitor (55) and the resistance value of the first static equivalent resistor (57);

Constructing a virtual three-dimensional coordinate system oxyz, wherein the x-axis, the y-axis and the z-axis of the virtual three-dimensional coordinate system respectively represent a fitting value of a first coefficient p, a fitting value of a second coefficient q and a fitting value of a proportional coefficient L _ex; searching coordinate points around the first coordinate point in an oxyz coordinate system by taking the first coordinate point as a starting point, taking a search sequence from a coordinate point close to the first coordinate point to a coordinate point far from the first coordinate point until a fitting error of the first coefficient p corresponding to the searched coordinate point, a fitting value of the second coefficient q, a fitting value of the proportionality coefficient L _ex and a resistance value of the first vibration system equivalent resistor (53), an inductance value of the first vibration system equivalent inductor (54), a capacitance value of the first vibration system equivalent capacitor (55) and a fitting error of an admittance curve/impedance curve corresponding to a model formed by a resistance value of the first static equivalent resistor (57) relative to an electric input admittance curve/electric input impedance curve of the electroacoustic transducer (10) is not larger than a fitting error set value, and fitting values of the first coefficient p corresponding to the searched coordinate point, the second coefficient q and the proportionality coefficient L _ex are respectively corresponding to the fitting value of the first coefficient p, the second coefficient q and the proportionality coefficient L _ex, so that the fitting error is concentrated to obtain a first model (_ex) of the electroacoustic transducer;

wherein the first coordinate point is a coordinate point composed of p ₀、q₀、L_ex0; p ₀＝1,q₀＝0,L_ex0 is the inductance value of the static equivalent inductance (56').

5. The bandwidth widening method for an electroacoustic transducer according to any one of claims 1 to 4, wherein: z _{Equivalent means} isAnd/>Ratio of/(I)For the difference between the measured voltage at the other end of the first capacitor (1) and the voltage at one end of the first capacitor (1),Outputting a current for the measured power amplifier (20).

6. The bandwidth widening method for an electroacoustic transducer according to any one of claims 1 to 4, wherein: Δm is calculated by a proportional-integral adjustment method from jωu _eq(ω)-Z_{Equivalent means}.

7. The bandwidth widening method for an electroacoustic transducer according to any one of claims 1 to 4, wherein: the inductance value L _s of the first inductor (2) and the capacitance value C _s of the first capacitor (1) satisfy the following formula:

Wherein F _n is the resonant frequency of the LC filter formed by the first inductor (2) and the first capacitor (1); f _p is the switching frequency of a switching tube of the single-phase full-bridge inverter (30); Δi _{ac_max} is 30% of the nominal output current effective value of the power amplifier (20).

8. The bandwidth widening method for an electroacoustic transducer according to any one of claims 1 to 4, wherein: the direct current power supply module (4) is connected with a second capacitor (3) in parallel.

9. An electroacoustic transducer bandwidth widening device utilizing a switched nonlinear non-foster system, characterized by comprising a processor configured for performing the steps of the electroacoustic transducer bandwidth widening method as claimed in any of the claims 1-8.