CN114243951B - Magnetic coupling type wireless power transmission system without parameter identification - Google Patents
Magnetic coupling type wireless power transmission system without parameter identification Download PDFInfo
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- CN114243951B CN114243951B CN202210159762.0A CN202210159762A CN114243951B CN 114243951 B CN114243951 B CN 114243951B CN 202210159762 A CN202210159762 A CN 202210159762A CN 114243951 B CN114243951 B CN 114243951B
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
- H02J50/00—Circuit arrangements or systems for wireless supply or distribution of electric power
- H02J50/10—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling
- H02J50/12—Circuit arrangements or systems for wireless supply or distribution of electric power using inductive coupling of the resonant type
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
- H02M1/083—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the ignition at the zero crossing of the voltage or the current
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
- H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/33507—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters
- H02M3/33523—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters with galvanic isolation between input and output of both the power stage and the feedback loop
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/3353—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having at least two simultaneously operating switches on the input side, e.g. "double forward" or "double (switched) flyback" converter
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
- H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
- H02M7/06—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
- H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
- H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
- H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
- H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
- Y02T90/00—Enabling technologies or technologies with a potential or indirect contribution to GHG emissions mitigation
- Y02T90/10—Technologies relating to charging of electric vehicles
- Y02T90/14—Plug-in electric vehicles
Abstract
The invention discloses a magnetic coupling type wireless power transmission system without parameter identification, which comprises a wireless power transmission module, a Boost converter, a first control module and a second control module, wherein the Boost converter is connected with the wireless power transmission module; the wireless power transmission module is used for outputting the input voltage to the Boost converter in a magnetic coupling type wireless power transmission mode; the Boost converter is used for transforming the input voltage and then providing the transformed input voltage to a load; the first control module is used for adjusting the input voltage of the wireless power transmission module according to the input voltage of the Boost converter, so that the voltage gain of the wireless power transmission module is an optimal value; the second control module is used for adjusting a driving signal provided for the Boost converter according to the output voltage of the Boost converter, so that the output voltage of the Boost converter is stable. The invention does not need mutual inductance and load identification, and has the advantages of high operation reliability and good output dynamic response.
Description
Technical Field
The invention belongs to the technical field of wireless power transmission, and particularly relates to a magnetic coupling type wireless power transmission system without parameter identification.
Background
The magnetic coupling type wireless energy transfer system (MCR-WPT) utilizes inductance-capacitance resonance and near-field energy coupling to enable electric energy transmission to get rid of physical constraint, high-efficiency transmission at medium distance is achieved, and the magnetic coupling type wireless energy transfer system is widely applied to the fields of underwater equipment, implanted medical equipment, unmanned aerial vehicles, electric vehicles and the like.
In a wireless energy transmission system, besides the need of ensuring the output performance, the system efficiency needs to be improved to reduce the power transmission cost and avoid the influence of a large amount of energy dissipated into heat energy on the performance of system devices. The improvement of the system efficiency can be mainly developed from the following two aspects: firstly, in the design stage of the system, the efficiency can be improved to the maximum extent by coil optimization, parameter design, device type selection and the like; in the operation stage of the system, due to the real-time change of the coil spacing and the load, the magnetic coupling resonator deviates from the optimal working point, and an impedance matching control strategy needs to be established for tracking with the maximum efficiency. Impedance matching control strategies can be divided into direct and indirect types according to different implementation ideas.
Most of the existing impedance matching control strategies are direct, mutual inductance and output load values need to be identified in real time for a receiving side converter to calculate duty ratio, the input equivalent resistance of a rectifier is directly modulated to an optimal value to track the maximum efficiency of a resonator, and meanwhile, output voltage is controlled at a transmitting end. In prior art 1, a half-controlled rectifier bridge is used to modulate an equivalent resistor to an optimal value, and a communication device is used to transmit a load output voltage to a front-end inverter to control output voltage stabilization. In the prior art 2, a rear-stage DC-DC control rectifier bridge is used to optimize an input equivalent resistance, and a first-order Linear Active Disturbance Rejection Controller (LADRC) is designed on a primary side to perform closed-loop control on an output voltage, so that the PI controller has the advantages of good dynamic performance, strong disturbance rejection capability and the like. This type of direct-type matching strategy suffers from the following problems: 1. the mutual inductance and the load identification make the system complicated, and the identification precision is difficult to ensure; 2. by utilizing a mode that a transmitting terminal controls the output voltage, the reliability of system operation and the output dynamic characteristic are very dependent on communication; 3. considering only the loss of the magnetic coupling resonator, the switch tube and the diode are assumed to be ideal devices, and actually, the inverter loss is closely related to the total loss of the system.
The indirect impedance matching control strategy is a matching strategy formulated according to the characteristics of electric parameters such as voltage and current when the output efficiency is maximum. In the prior art 3, a rear-end converter is used for controlling a load to output a constant voltage, the minimum value of the input power of a system is tracked by a disturbance observation method to realize the optimal efficiency of the system, and the influence of the loss of the converter is included in the equivalent resistance of a resonator, so that the impedance matching is equivalently realized.
Disclosure of Invention
In view of at least one of the defects or improvement requirements of the prior art, the invention provides a magnetic coupling type wireless power transmission system without parameter identification, which does not need mutual inductance and load identification, and has the advantages of high operation reliability and good output dynamic response.
In order to achieve the above object, the present invention provides a magnetic coupling wireless power transmission system without parameter identification, including a wireless power transmission module, a Boost converter, a first control module and a second control module;
the wireless power transmission module is used for outputting the input voltage of the wireless power transmission module to the Boost converter in a magnetic coupling type wireless power transmission mode;
the Boost converter is used for transforming the input voltage and then providing the transformed input voltage to a load;
the first control module is used for adjusting the input voltage of the wireless power transmission module according to the input voltage of the Boost converter, so that the voltage gain of the wireless power transmission module is an optimal value;
the second control module is used for adjusting a driving signal provided for the Boost converter according to the output voltage of the Boost converter, so that the output voltage of the Boost converter is stable.
Further, the wireless power transmission module includes a resonant network, and the operating frequency of the wireless power transmission module is set so that a transmitting end of the resonant network is in a full-harmonic state and a receiving end of the resonant network is in a detuned state.
Further, the first control module and the second control module perform multi-wheel regulation until the outputs of the wireless power transmission module and the Boost converter are in a stable state.
Further, the calculation of the optimal value of the voltage gain of the wireless power transmission module includes the steps of:
determining a system efficiency expression of the wireless power transmission module, wherein the system efficiency expression is a function of the detuning rate;
determining an optimal equivalent resistance expression corresponding to the maximum efficiency according to the system efficiency expression;
calculating the optimal equivalent resistance values under different detuning rates;
performing linear fitting on the detuning rate and the optimal equivalent resistance value to determine a fitting expression of the optimal equivalent resistance;
and determining a calculation formula of the optimal voltage gain value of the wireless power transmission module according to the fitting expression of the optimal equivalent resistance.
Further, the calculation formula of the optimal voltage gain value of the wireless power transmission module is a function with the detuning rate of the receiving end of the resonant network, the internal resistance of the transmitting coil and the internal resistance of the receiving coil of the resonant network as parameters.
Further, the calculation formula of the optimal value of the voltage gain of the wireless power transmission module is as follows:
Further, the wireless power transmission module further comprises an inverter and a rectification module, the inverter is used for converting input direct-current voltage into alternating-current voltage and supplying the alternating-current voltage to the resonant network, the rectification module is used for rectifying the output of the resonant network, and the inverter and the rectification module are both in 180-degree complementary conduction.
Further, the input voltage of the wireless power transmission module is a ratio of the input voltage of the Boost converter to the optimal voltage gain value of the wireless power transmission module.
In general, compared with the prior art, the invention has the following beneficial effects: the input voltage of the wireless power transmission module and the Boost converter cascaded at the receiving end are utilized to provide two control degrees of freedom, the first control module adjusts the input voltage of the wireless power transmission module according to the input voltage of the Boost converter, so that the voltage gain of the wireless power transmission module is an optimal value, the optimal value of the voltage gain of the wireless power transmission module is not related to mutual inductance of a resonant network and system load, and therefore mutual inductance and load identification are not needed, and the wireless power transmission module has the advantages of high operation reliability and good output dynamic response.
Drawings
Fig. 1 is a structural diagram of a magnetically-coupled wireless power transfer system of an embodiment of the present invention;
fig. 2 is an equivalent circuit phasor model of a wireless power transmission module of the magnetically coupled wireless power transmission system according to an embodiment of the present invention;
FIG. 3 is a linear fit of the optimal equivalent resistance for an embodiment of the invention;
fig. 4 is a schematic control loop diagram of a magnetically coupled wireless power transmission system according to an embodiment of the invention.
Detailed Description
In order to make the objects, technical solutions and advantages of the present invention more apparent, the present invention is described in further detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely illustrative of the invention and are not intended to limit the invention. In addition, the technical features involved in the embodiments of the present invention described below may be combined with each other as long as they do not conflict with each other.
In order to realize that the system still keeps high-efficiency operation under the dynamic change of load and mutual inductance, the embodiment of the invention provides a magnetic coupling type wireless electric energy transmission system without parameter identification.
The magnetic coupling type wireless power transmission system without parameter identification comprises a wireless power transmission module, a Boost converter, a first control module and a second control module; the wireless power transmission module is used for outputting the input voltage to the Boost converter in a magnetic coupling type wireless power transmission mode; the Boost converter is used for transforming the input voltage and then providing the transformed input voltage to a load; the first control module is used for adjusting the input voltage of the wireless power transmission module according to the input voltage of the Boost converter, so that the voltage gain of the wireless power transmission module is an optimal value; the second control module is used for adjusting a driving signal provided for the Boost converter according to the output voltage of the Boost converter, so that the output voltage of the Boost converter is stable.
Further, the wireless power transmission module includes a resonant network, and the operating frequency of the wireless power transmission module is set so that the transmitting end of the resonant network is in a full-harmonic state and the receiving end is in a detuned state.
In the magnetic coupling type wireless power transmission system, the wireless power transmission module can adopt various resonant networks such as SS, LCC-S and the like. Similar control strategies may be used in different types of resonant networks, but the specific calculation formulas may differ.
The working principle of the system is described below by taking the wireless power transmission module as an SS-type resonant network as an example. Firstly, a mutual inductance circuit model of the SS type MCR-WPT system is established, and the working principle of the system and the implementation method of the inverter soft switch under the unilateral detuning design are analyzed. And then analyzing the maximum efficiency tracking principle of the SS type resonator, and deducing an optimal voltage gain expression when the soft switching of the inverter and the maximum efficiency of the resonator are realized simultaneously. And finally, the receiving end Boost converter is used for outputting voltage stabilization, the transmitting end can adjust the DC power supply to coordinate and control to realize optimal voltage gain, and the indirect impedance matching strategy does not need mutual inductance and load identification and has the advantages of high operation reliability and good output dynamic response.
1. General framework and working principle of system
According to the magnetic coupling type wireless power transmission system without parameter identification, soft switching of an inverter is achieved through a secondary side detuning design, a composite control strategy of optimal load indirect control and constant voltage output without parameter identification is formulated, namely a receiving end Boost converter is used for outputting stable voltage, a transmitting end can adjust a direct current source to coordinately control optimal voltage gain to achieve impedance matching indirectly, and the structure diagram of the system is shown in fig. 1.
The topological structure of the main circuit comprises a wireless power transmission module and a Boost converter. The wireless power transmission module comprises an adjustable direct-current power supply, a high-frequency full-bridge inverter, an SS type resonant network and a Boost converter of a rectifier. In the context of figure 1 of the drawings,V DCis an adjustable DC power supply, 4 MOS tubesQ 1~Q 4To form a primary side inverter,u 1is the output port voltage of the inverter and,L 1andL 2the self-inductance of the transmit coil and the receive coil respectively,R 1is the internal resistance of the transmitting coil and,R 2is the internal resistance of the receiving coil and,Mis the mutual inductance between the coils,C 1is the compensation capacitance of the transmitting coil and,C 2is the compensation capacitance of the receiving coil and,i 1is the transmit coil current flowing into the end of the same name,i 2is the current of a receiving coil flowing into the same name terminal, 4 diodesVD 1~VD 4The secondary side rectifier is formed by the two parts,C infor the output of the filter capacitor of the rectifier,R eis the equivalent resistance of the input port of the rectifier,u 2is input port voltage of rectifier, inductanceL 3And MOS tubeQ 5Diode, and method of manufacturing the sameVD 5Filter capacitorC oThe Boost converter is formed to have a Boost converter,R inis an input resistor of the Boost converter,V inin order to input the voltage to the Boost converter,I infor the input current of the Boost converter,R Las a result of the output resistance, the resistance,V oin order to output the voltage, the voltage is,I oto output a current.
A system control circuit is designed that may include a first control module and a second control module. In the embodiment of the invention, the first control module is a transmitting end adjustable direct current source control circuit, and the second control module is a receiving end Boost control circuit. The two control loops are mutually independent and run in a decoupling mode.
The embodiment of the invention utilizes the process that the back-end Boost converter outputs the stabilized voltage to be independent of communication equipment, and the reliability of system operation and the dynamic output characteristic are better. Regulating a controllable direct current sourceV DCThe process of realizing the optimal voltage gain of the resonator to indirectly realize the impedance matching does not need mutual inductance and load identification, simplifies a control algorithm and is suitable for the working condition of variable coupling coefficients.
2. SS-WPT system modeling and unilateral detuning principle analysis
a) Working principle of SS-WPT system
Since the resonant network has a filtering effect on higher harmonics, the current in the resonant network has almost only a fundamental component, and the system is analyzed by a fundamental analysis method, and fig. 2 is an equivalent circuit phasor model of the main circuit in fig. 1. Wherein the content of the first and second substances,ωin order to obtain the angular frequency of operation of the system,
is an equivalent alternating voltage source of the output port of the inverter,
for an equivalent voltage at the input port of the rectifier,
is the current of the primary side,
is the current of the secondary side, and the current of the secondary side,U 1source of alternating voltage for output port of inverterThe effective value of the effective value is,U 2is the effective value of the voltage at the input port of the rectifier,I 1is the effective value of the primary side current,I 2the effective value of the secondary side current.
According to kirchhoff's voltage law, the voltage equation for the write-back path can be written:
solving the loop current:
wherein the content of the first and second substances,Z 11is the self-impedance of the primary side,Z 22is the self-impedance of the secondary side,Z Min order to be a mutual impedance,
，Z 22=R 2+jωL 2+1/ jωC 2，Z M=jωM，R eis the equivalent resistance of the input port of the rectifier.
Resonant frequency at primary side of
The resonance frequency at the secondary side is
In order to ensure the integral resonance of the system and reduce the reactive power content, the primary side and the secondary side are in a resonance state, and the working frequency of the inverter is equal to the resonance frequency of the two sides.
ω=ω p=ω s （3）
At this time, the process of the present invention,Z 11=R 1，Z 22=R 2，Z M=jωMsubstitution into the loop current equation (2)Simplifying:
is provided with
=U 1The angle of a point vector is represented by < 0 degrees and < represents, a primary current and an equivalent alternating current voltage source are in the same phase, a system impedance angle is pure resistance, and the inverter cannot realize soft switching. Internal resistance of coilR 1、R 2Is far less thanωMEffective value of secondary side currentI 2Can be approximated as:
equivalent resistanceR eWhen the system is changed, the secondary side current is almost unchanged, high-power transmission can be performed, and in addition, due to the output constant current characteristic of the SS resonator, the system can still work safely when secondary side short-circuit faults such as rectifier direct connection, capacitor breakdown and the like occur. Voltage gainG VCan be approximated as:
the voltage gain is approximately proportional to the equivalent resistance. Output efficiencyη：
b) Single side detuning design principle
In order to realize the soft switching of the inverter, the detuning design of a resonant network is needed, so that the equivalent input impedance angle of the system is weak. To simplify the analysis, the present invention considers only a single-sided detuning design approach.
Defining detuning rateμ:
Primary side full-harmonic secondary side detuning design:
at this time, the inverter operating frequency is equal to the primary side resonant frequency,
ω=ω p （9）
at this time, the process of the present invention,Z 11=R 1，Z 22=R 2+jωL 2 μ，Z M=jωMsubstituting into a loop current equation (2) for simplification:
input impedance angle of systemφ 1Comprises the following steps:
μwhen the pressure is higher than 0, the pressure is higher,φ 1＞0；μwhen the ratio is less than 0, the reaction mixture is,φ 1less than 0, and the secondary side is in detuning design, so that the input port of the resonant network is in weak inductanceμIs less than 0. Due to the fact thatωL 2 R 1Small component and detuning rateμAnd equivalent resistanceR eTo pairφ 1Non-direct influence, soft switching characteristics on detuning rateμLow sensitivity in disturbance and equivalent resistanceR eThe influence is small when the range is changed.
Due to internal resistance of the coilR 1、R 2And detuning rateμAre all in the order of 10-1Is far less thanωMThe effective secondary current value may be approximated as:
also due toωL 2 R 1Small component, secondary side currentI 2For detuning rateμLow sensitivity at disturbance and equivalent resistanceR eWidely varying pairI 2The influence is small, and the secondary side is approximately constant current. Voltage gainG V Can be approximated as:
the voltage gain is still approximately proportional to the equivalent resistance under the secondary side detuned design. Output efficiency:
primary side detuning secondary side full harmonic design:
at this time, the inverter operating frequency is equal to the secondary side resonance frequency,
ω=ω s （15）
at this time, the process of the present invention,Z 11=R 1+jωL 1 μ，Z 22=R 2，Z M=jωMand substituting the loop current equation (2) for simplification:
system input impedance angleφ 1Comprises the following steps:
μwhen the pressure is higher than 0, the pressure is higher,φ 1＜0，μwhen the ratio is less than 0, the reaction mixture is,φ 1the condition that the input port of the resonant network is weak inductance is met under the primary side detuning design when the input port of the resonant network is larger than 0 DEG CμIs greater than 0. In generalL 1≥L 2，（R e+R 2）＞＞R 1，ωL 1（R e+R 2) Far greater thanωL 2 R 1Component, slight detuning rate variation ΔμCan causeφ 1Large variation, soft switching characteristic versus detuning rateμThe disturbance is very sensitive, the factors such as the manufacturing process and the like can cause the detuning capacitance to have errors,μthe deviation from the design value results in low tolerance of the primary side detuning design method to the detuning capacitor;R esmall detuning rate to make system impedance angle weakμIn aR eWhen the voltage is increased, the voltage will be inductive, and the soft switching characteristic is subject to the equivalent loadR eThe influence of fluctuation is large.
The effective value of the secondary side current is as follows:
also due toωL 1（R e+R 2) High component, secondary currentI 2For detuning parametersμHigh sensitivity to disturbanceμLower equivalent loadR eChange also toI 2There is also a greater effect.
In summary, compared with the primary side detuning design, the secondary side detuning design has soft switching characteristic and output constant current characteristic affected by the equivalent resistanceR eSmall influence on detuning parameters in wide rangeμThe sensitivity is low during disturbance, and the tolerance degree to the secondary side detuning capacitance is higher, so the embodiment of the invention adopts the design of transmitting end full tuning and receiving end detuning to realize softThe switches reduce inverter losses. In addition, when the primary side parameters cannot be accurately configured, the primary side can be made to be fully resonant by fine-tuning the operating frequency of the system.
3. Maximum efficiency tracking principle and composite control analysis
a) Principle of maximum efficiency tracking
And the corresponding voltage gain is an optimal value when the output efficiency of the wireless electric energy transmission module is maximum. The optimal value of the voltage gain of the wireless power transmission module has no correlation with the mutual inductance of the resonant network and the system load.
The calculation of the optimal value of the voltage gain of the wireless power transmission module comprises the following steps:
(1) determining a system efficiency expression of the wireless power transmission module, wherein the system efficiency expression is a function of the detuning rate;
(2) determining an optimal equivalent resistance expression corresponding to the maximum efficiency according to the system efficiency expression;
(3) calculating the optimal equivalent resistance values under different detuning rates;
(4) performing linear fitting on the detuning rate and the optimal equivalent resistance value, and determining a fitting expression of the optimal equivalent resistance;
(5) and determining a calculation formula of the optimal voltage gain value of the wireless power transmission module according to the fitting expression of the optimal equivalent resistance.
Recording the corresponding optimal equivalent resistance when the system efficiency expression determines that the efficiency is maximum asR e-optWhen it is to be fully tunedR e-optIs marked as
。
Expressing the output efficiency at the time of full harmonic on two sides (7) to equivalent resistanceR eDerivation is carried out to obtain the optimal equivalent resistance when the full harmonic is obtained
：
Substituting into the approximate expression (6) of voltage gain at full harmonic to obtain the optimal voltage gain at full harmonic
：
It can be seen that under the full-harmonic design of the SS type resonator, the output efficiency is maximized when the equivalent resistance is the optimal resistance, the resistance is related to not only the intrinsic parameters of the resonator but also the variable-parameter mutual inductance, and the voltage gain at this time is called as the optimal voltage gain and is only related to the fixed parameters of the resonator.
Output efficiency expression (14) for equivalent resistance when designing secondary side detuningR eDerivation, analysing the resonator output efficiency by solving analytical expressions which are too complexηOptimum equivalent resistanceR e-optAnd detuning rateμThe basic parameters of the SS type MCR-WPT system shown in Table 1 are adopted.
TABLE 1 System parameters
When considering the secondary side detuning design (μ< 0), different detuning ratesμLower output efficiency-equivalent resistance curveμThe larger the greater the value of | is,
the larger the corresponding output efficiency maximumη maxThe smaller.
The essence of the detuning design is to sacrifice resonator efficiency to achieve soft switching of the inverter, so that the circuit count is reduced while achieving soft switching over the range of load variationμ| should be as close to 0 as possible so that the resonator is reachableη maxAs large as possible. Secondary side detuning arrangementTiming deviceμ<0, so as to be paired at intervals of 0.01μIn the range of [ -0.2,0 [)]Taking value between two adjacent resonators, and inspecting the optimal equivalent resistance when the output efficiency of the resonator is maximumR e-optAnd optimum voltage gainG v-optAnd detuning rateμThe relationship between them, Table 2 is differentμIs as followsR e-optThe value is obtained.
TABLE 2 different detuning ratesμOptimum resistance value of
As shown in fig. 3, a scatter plot was plotted, the fitted model was observed and determined,R e-optaboutμApproximately in a linear relationship, fitting with a linear model:
the linear fit mathematical model is as follows,
is composed ofR e-optThe fitting value of (c):
full harmonic time (μ= 0), fitted value of optimum equivalent resistance
Is 14.15 omega, the actual value
15.36 Ω (as can be seen from table 2), with an error of about 1.2 Ω between the two. But for simplicity of analysis, the actual value of the optimal equivalent resistance at full harmonic is assumed to be equal to the fitted value:
is different fromμFitting value of lower optimal equivalent resistance
With actual value at full resonance
The following relationships exist:
bonding of
Expression (19) of (a), obtaining a differenceμIs as follows
Expression:
substituting the formula (24) into the voltage gain approximate formula (13) in the secondary side detuning design to obtain the differenceμOptimum voltage gain of
Expression:
it can be seen that the same rule exists between the secondary side detuning design and the full-harmonic design of the SS type resonator when the output efficiency is maximum, and the corresponding optimal equivalent resistance not only has fixed parameters with the resonatorR 1、R 2、μ、ωRelated to, and mutual inductanceMThis variation parameter is related; the optimal voltage gain is only related to the intrinsic parameters of the resonatorR 1、R 2、μCorrelation, thereby obtaining a correlation without parameter identificationI.e. a method that can track with maximum efficiency.
b) Voltage stabilization and maximum efficiency tracking composite control analysis
The wireless electric energy transmission system often requires constant voltage output, the control of the system can realize both the constant voltage output and the maximum efficiency tracking, the embodiment of the invention keeps the inversion bridge phase-shifting complementary conduction, and utilizes the controllable direct current source at the transmitting endV DCAnd a Boost converter cascaded at the receiving end to provide two degrees of control freedom. As can be seen from the above section, the optimal voltage gain expression when the soft switching of the inverter and the maximum efficiency of the resonator are simultaneously achieved,
this electrical parameter law can be used to develop an indirect optimal load matching strategy to track maximum efficiency for a value that is related only to the intrinsic parameters of the resonator.
The control strategy utilizes a rear-stage Boost converter to output voltage stabilization and regulate a front-end controllable direct-current sourceV DCWhen the resonator voltage gain is made to be maximum in output efficiency
And the control process does not need load and mutual inductance identification. Because the primary inverter and the secondary rectifier are in 180-degree complementary conduction, the input voltage of the Boost converter can be controlledV inAnd a controllable DC sourceV DCIn a ratio of
：
When the inductive current of the Boost converter is continuous, the input voltageV inAnd an output voltageV oThe relationship of (1) is:
in the formula (I), the compound is shown in the specification,Dis the duty cycle of the Boost converter.
Input resistanceR inAnd a load resistorR LThe relationship of (1) is:
as can be seen from the transmission efficiency expression (14), the transmission efficiency of the resonator is only equal to the current mutual inductance valueMEquivalent resistance ofR eWith respect to the gain of the resonator voltage of
At that time, the output efficiency is maximized, at that timeR eMust be the optimum value becauseR eAndDandR Lhaving a relationship of formula (29) where there must be a unique correspondenceD ∞ See formula (30).
Output voltage at maximum efficiencyV oAndD ∞ andV DCthe relationship of (a) to (b) is as follows:
by the analysis, the maximum efficiency tracking and the output voltage stabilization control are mutually decoupled and operated.
c) Voltage stabilization and maximum efficiency tracking composite control process analysis
The system dynamics control loop is shown in fig. 4.
When the output resistanceR LWhen the change occurs, the detected output voltageV oSending the data to a back-end PI controller to adjust the duty ratio of a back-stage Boost converterDSo that the output voltage is stabilized as a reference valueV o-ref. Input equivalent resistance of rectifierR eWith followingDIs changed, the voltage gain is necessarily deviated from the optimal value because the SS type resonator is in an output constant current topology. Using wireless communication equipment to collect input voltage of Boost converterV inTo a front-end controller for regulating a controllable DC sourceV DCMake the voltage gain of the resonator to the optimum value
. WhileV DCCan in turn influenceV oThen, a new cycle of adjusting the duty cycle is continuedDRealizing voltage stabilization and adjusting controllable direct current sourceV DCThe voltage gain is optimized and repeated, and when the output resistance value meets the boundary condition of system control:
this reciprocating dynamic coordination control process eventually goes to steady state,DandV DCrespectively converge to steady state valuesD ∞、V DC∞。
The method is used for solving the problems that the system becomes complicated and the identification precision is difficult to ensure due to mutual inductance and load identification in the direct impedance matching method, and can realize the improvement of the system efficiency under the variable coupling working condition. The invention analyzes the realization method of the inverter soft switch under the unilateral detuning design, deduces the optimal voltage gain expression when realizing the inverter soft switch and the maximum efficiency of the resonator at the same time, and is only related to the intrinsic parameters of the resonator, thereby obtaining the indirect impedance matching method without parameter identification. The receiving end Boost converter is used for outputting voltage stabilization, the transmitting end adjustable direct current power supply is used for realizing optimal voltage gain in a coordinated control mode, the indirect impedance matching strategy does not need mutual inductance and load identification, and the indirect impedance matching strategy has the advantages of high operation reliability and good output dynamic response.
It must be noted that in any of the above embodiments, the methods are not necessarily executed in order of sequence number, and as long as it cannot be assumed from the execution logic that they are necessarily executed in a certain order, it means that they can be executed in any other possible order.
It will be understood by those skilled in the art that the foregoing is only a preferred embodiment of the present invention, and is not intended to limit the invention, and that any modification, equivalent replacement, or improvement made within the spirit and principle of the present invention should be included in the scope of the present invention.
Claims (6)
1. A magnetic coupling type wireless power transmission system without parameter identification is characterized by comprising a wireless power transmission module, a Boost converter, a first control module and a second control module;
the wireless power transmission module is used for outputting the input voltage to the Boost converter in a magnetic coupling type wireless power transmission mode;
the Boost converter is used for transforming the input voltage and then providing the transformed input voltage to a load;
the first control module is used for adjusting the input voltage of the wireless power transmission module according to the input voltage of the Boost converter, so that the voltage gain of the wireless power transmission module is an optimal value;
the second control module is used for adjusting a driving signal provided for the Boost converter according to the output voltage of the Boost converter so as to stabilize the output voltage of the Boost converter;
the calculation of the optimal value of the voltage gain of the wireless power transmission module comprises the following steps:
determining a system efficiency expression of the wireless power transmission module, wherein the system efficiency expression is a function of the detuning rate;
determining an optimal equivalent resistance expression corresponding to the maximum efficiency according to the system efficiency expression;
calculating the optimal equivalent resistance values under different detuning rates;
performing linear fitting on the detuning rate and the optimal equivalent resistance value, and determining a fitting expression of the optimal equivalent resistance;
determining a calculation formula of the optimal value of the voltage gain of the wireless power transmission module according to the fitting expression of the optimal equivalent resistance;
the calculation formula of the optimal value of the voltage gain of the wireless power transmission module is as follows:
2. The system of claim 1, wherein the wireless power transmission module comprises a resonant network, and an operating frequency of the wireless power transmission module is set such that a transmitting end of the resonant network is in a full-harmonic state and a receiving end of the resonant network is in a detuned state.
3. The system of claim 1, wherein the first control module and the second control module perform multiple rounds of adjustment until the outputs of the wireless power transmission module and the Boost converter are in a stable state.
4. The system of claim 2, wherein the optimal voltage gain of the wireless power transmission module is calculated as a function of the detuning rate at the receiving end of the resonant network, the internal resistance of the transmitting coil and the internal resistance of the receiving coil of the resonant network.
5. The system of claim 2, wherein the wireless power transmission module further comprises an inverter and a rectifier module, the inverter is configured to convert an input dc voltage into an ac voltage and provide the ac voltage to the resonant network, the rectifier module is configured to rectify an output of the resonant network, and the inverter and the rectifier module are both in 180-degree complementary conduction.
6. The system of claim 1, wherein the input voltage of the wireless power transmission module is a ratio of the input voltage of the Boost converter to an optimal value of a voltage gain of the wireless power transmission module.
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