CN115460509A

CN115460509A - Method and apparatus for widening bandwidth of transducer using nonlinear non-Foster system

Info

Publication number: CN115460509A
Application number: CN202211071042.5A
Authority: CN
Inventors: 杨鑫; 许梦伟; 张智贺; 李姝汛; 欧阳晓平
Original assignee: Changsha Semiconductor Technology And Application Innovation Research Institute; Hunan University
Current assignee: Changsha Semiconductor Technology And Application Innovation Research Institute; Hunan University
Priority date: 2022-09-02
Filing date: 2022-09-02
Publication date: 2022-12-09
Anticipated expiration: 2042-09-02
Also published as: CN115460509B

Abstract

The invention provides an electroacoustic transducer bandwidth widening method by utilizing a nonlinear non-Foster system, which comprises the following steps: calculating the modulation degree m' and the modulation phase angle theta of the single-phase full-bridge inverter; in a first lumped parameter model of the electroacoustic transducer with inductive input impedance, a first static equivalent resistance model, an equivalent inductance model and a first vibration system equivalent circuit structure are mutually connected in series between two driving ends of the electroacoustic transducer; the equivalent inductance model is an eddy current impedance model formed by mutually connecting a nonlinear inductance model and a nonlinear resistance model in series; and the structure of the reactance model and the resistance model which are connected in series with each other and the structure of the first static equivalent resistance, the equivalent inductance model and the equivalent circuit structure of the first vibration system which are connected in series with each other are equivalent structures.

Description

Transducer bandwidth widening method and device using nonlinear non-Foster system

Technical Field

The invention relates to the technical field of underwater acoustic communication, in particular to a method and a device for widening the bandwidth of a transducer by utilizing a nonlinear non-Foster system, which are used for broadband live acoustic emission.

Background

With the continuous development of underwater wireless communication technology and detection means, sound waves become a unique carrier capable of transmitting information underwater at a long distance due to the advantages of high wave speed, low attenuation frequency and the like of the sound waves in an aqueous medium. Therefore, an underwater sound technology capable of researching an underwater sound propagation rule and underwater signal processing is derived, and a key ring in the underwater sound technology is an electroacoustic transmitting device. The system can complete high-power energy conversion and realize long-distance information transmission, and mainly comprises an electromagnetic energy storage element and a mechanical vibration system: the energy storage element is responsible for completing the motor conversion under certain physical effect, and the mechanical vibration system is responsible for outputting the converted energy, namely sound energy.

In practical application of the electroacoustic transmitting device, a relatively serious impedance mismatch exists between a driving power supply and the device, reactive loss is generated, the driving power supply cannot obtain the maximum power from a power supply end, and the output power and the signal transmission efficiency of a system are influenced. In order to improve the transmission efficiency and quality of the high-power electroacoustic transmitting device, a broadband impedance matching network needs to be additionally arranged to counteract reactive loss caused by equivalent reactance of the high-power electroacoustic transmitting device, and expand the resonance bandwidth of a system. The traditional passive matching mode is limited by a gain bandwidth theory, can only realize matching at a single frequency and cannot meet the requirement of broadband matching.

The existing non-foster matching system is composed of transistors and operational amplifiers, the output capability of the transistors and the operational amplifiers is extremely limited, and the output characteristic presents linear characteristics due to the limitations of the working principle and the internal structure of the transistors and the operational amplifiers. The linear non-Foster matching system is only suitable for high-frequency-band micro-power sound sources, and is currently and intensively applied to miniature devices such as miniature antennas and the like with almost unchanged input reactance in the working frequency range. However, due to the existence of dynamic mechanical impedance, the impedance characteristic presented by the high-power electroacoustic transmitting device in the field of underwater acoustic communication is no longer completely consistent with the static impedance of an electric terminal in the broadband operation process, so that the input impedance characteristic of a system is not fixed and can be changed along with the change of frequency.

In addition, the high-power electroacoustic transmitting device is in a complex working environment, and a plurality of obvious nonlinear phenomena exist in the working process of the high-power electroacoustic transmitting device, so that the input impedance characteristic of the device presents nonlinear characteristics. This means that the transducer often operates in a nonlinear state, and energy loss and temperature disturbance are caused during operation, so that nonlinear harmonic distortion is generated in the output response of the transducer, and the output performance of the transducer is seriously affected. The input reactance of the electroacoustic transducer can generate nonlinear change along with the change of working conditions and external disturbance, so that a constant-output matching network cannot accurately match the high-power electroacoustic transducer, the power magnitude and the matching linearity (self-adaptability) of the conventional linear non-Foster matching system cannot meet the requirement of a large bandwidth of a high-power electroacoustic transmitting device, the applicability is extremely limited, the requirement of broadband real-time adjustment of the high-power electroacoustic transmitting device cannot be met, and the energy transmission efficiency of the system is reduced.

Disclosure of Invention

The invention provides a method and a device for widening the bandwidth of an electroacoustic transducer by using a switch type nonlinear non-Foster system, aiming at the problem that a linear non-Foster matching system consisting of a transistor and an operational amplifier in the prior art cannot meet the requirement of the high bandwidth of a high-power electroacoustic transducer.

In order to solve the technical problems, the invention adopts the technical scheme that: the bandwidth widening method of the electroacoustic transducer by utilizing the switch type nonlinear non-Foster system is characterized in that the input impedance of the electroacoustic transducer is inductive, a driving signal of the electroacoustic transducer is provided by a power amplifier, the electroacoustic transducer is provided with a first centralized parameter model, and the first centralized parameter model is composed of a first static equivalent resistor, an equivalent inductance model and a first vibration system equivalent circuit structure which are mutually connected in series between two driving ends of the electroacoustic transducer;

the switch type nonlinear non-Foster system is characterized by comprising a first capacitor, a first inductor and a single-phase full-bridge inverter connected to the driving end of the electroacoustic transducer; two switching tubes of the single-phase full-bridge inverter are provided with a first common connecting end, and the other two switching tubes are provided with a second common connecting end; two input ends of the single-phase full-bridge inverter are respectively and correspondingly electrically connected with two output ends of the direct-current power supply module; the first common connecting end is electrically connected with one end of the first inductor, and the second common connecting end, one end of the first capacitor and one driving end of the electroacoustic transducer are electrically connected with each other;

the other end of the first inductor, the other end of the first capacitor and one output end of the power amplifier are electrically connected with a first electric connection point, and the other output end of the power amplifier and the other driving end of the electroacoustic transducer are electrically connected with each other;

the bandwidth widening method of the electroacoustic transducer comprises the following steps:

the modulation degree m' and the modulation phase angle theta of the single-phase full-bridge inverter are calculated by the following formula, wherein

m’＝m+Δm；

Wherein the expression of angular frequency ω is ω =2 π F _s ，V _dc Is the output voltage of the DC supply module, L _s 、C _s The inductance value of the first inductor and the capacitance value of the first capacitor are respectively; e _m 、F _s 、θ _m Respectively output power to the power amplifierAmplitude, frequency, phase angle of the voltage signal;

in a first lumped parameter model of the electroacoustic transducer, a first static equivalent resistance model, an equivalent inductance model and a first vibration system equivalent circuit structure are mutually connected in series between two driving ends of the electroacoustic transducer; the equivalent inductance model is an eddy current impedance model formed by mutually connecting a nonlinear inductance model and a nonlinear resistance model in series;

U _eq (ω) is the ratio of the reactance value of the compensated reactance model to the angular frequency ω, Z _r (ω)、θ _r (ω) is the impedance mode and the impedance angle corresponding to the impedance expression of the residual impedance model, respectively;

the compensated reactance model and the residual impedance model are connected in series between two driving ends of an electroacoustic transducer (10), so as to form an equivalent structure of the first lumped parameter model;

Δ m is according to j ω U _eq (ω)-Z _{Equivalence of} Calculating the obtained modulation degree feedback quantity; z _Equivalence Is the measured value of the equivalent impedance of the switch type nonlinear non-Foster system between the first electric connection point and the second common connection point, j represents an imaginary number unit;

(A) The first lumped parameter model is equivalent to a structure formed by connecting an equivalent reactance model and an equivalent resistance model in series, and the compensated reactance model and the residual impedance model are respectively corresponding to the equivalent reactance model and the equivalent resistance model;

X _eq (ω) is the reactance value of the equivalent reactance model;

(B)U _eq (ω)＝L _em (ω), the compensated reactance model is a nonlinear inductance model, and the first static equivalent resistance, the nonlinear resistance model, and the first vibration system equivalent circuit structure constitute the residual impedance model.

Through the arrangement, the modulation degree m' of the single-phase full-bridge inverter can track the frequency of the voltage signal output by the power amplifier, so that the resonance bandwidth of the system can be expanded, the modulation degree can be adjusted along with the frequency of the voltage signal output by the power amplifier, the equivalent reactance can be prevented from being offset under single frequency, and the single-phase full-bridge inverter is suitable for broadband matching. The value matched with the target is subtracted from the actually measured value, so that the modulation degree feedback quantity is calculated, closed-loop control can be realized, and the control precision is higher. Through the arrangement, the non-Foster impedance matching network is arranged, negative impedance is generated to offset the equivalent reactance of the transducer, namely only an equivalent resistance model exists in a model of the electroacoustic transducer after the offset, and therefore the transmission efficiency and the quality of output signals of the electroacoustic transducer can be improved.

In the invention, the modulation degree m' and the modulation phase angle theta of the single-phase full-bridge inverter are adjusted through the formula, so that the reactance value corresponding to the equivalent reactance model in the lumped parameter model is offset by the equivalent impedance of a circuit structure consisting of the single-phase full-bridge inverter, the first capacitor and the first inductor between the first electric connection point and the second common connection end.

In the invention, the equivalent inductance model is an eddy current impedance model formed by mutually connecting the nonlinear inductance model and the nonlinear resistance model in series, so that the influence of eddy current factors is considered in the static inductance model, and the model is more accurate.

In the invention, the modulation degree m' and the modulation phase angle theta of the single-phase full-bridge inverter are adjusted and adjusted through the formula _s And the equivalent impedance of a circuit structure formed by the single-phase full-bridge inverter, the first capacitor and the first inductor between the first electric connection point and the second common connection end offsets the impedance value corresponding to the static equivalent inductor in the lumped parameter model. I.e. such that the voltage across the static equivalent inductance is cancelled from the voltage between the first electrical connection point and the second common connection terminal. Therefore, the output voltage of the power amplifier can be provided to the equivalent circuit structure of the vibration system as far as possible without considering other losses, and the performance of the vibration system of the transducer is prevented from being influenced as far as possible.

In the present invention, only the nonlinear inductance model may be used as the compensated reactance model, or the first lumped-parameter model may be equivalent to a structure in which an equivalent reactance model and an equivalent resistance model are connected in series with each other, and then the equivalent reactance model may be used as the compensated reactance model. When the equivalent reactance model is used as a compensated reactance model, when the reactance of the electric end and the mechanical end of the electroacoustic transducer changes due to change of working conditions or external disturbance, a good matching effect can be still realized on the nonlinear reactance of the electric end and the mechanical end of the electroacoustic transducer, so that the influence of nonlinear change of the reactance is reduced, and the requirement of a larger bandwidth of the electroacoustic transmitting device is met.

In the above technical scheme: an inductance value L of the nonlinear inductance model (51) _em (omega) resistance value R of nonlinear resistance model (52) _em The expression of (ω) is:

wherein θ (ω) is a first phase angle value, L _ex Is a proportionality coefficient, p is a first coefficient, q is a second coefficient, p is more than 0 and less than 1, q is more than 0 and less than 1; wherein p (ω) = a ₀ +a ₁ ω+a ₂ ω ² ，q(ω)＝a ₃ +a ₄ ω+a ₅ ω ² ，a ₀ 、a ₁ 、a ₂ 、a ₃ 、a ₄ 、a ₅ Are all coefficients.

In the above technical scheme: the method comprises the steps of applying an excitation signal to a driving end of the electroacoustic transducer, obtaining an electric input admittance curve/electric input impedance curve of the electroacoustic transducer, and fitting according to the electric input admittance curve/electric input impedance curve to obtain a first lumped parameter model of the electroacoustic transducer.

In the invention, the first lumped parameter model is obtained according to the actual electric input admittance curve/electric input impedance curve of the electroacoustic transducer, so that the established model is consistent with the actual transducer performance, the effect of generating negative impedance by the switch type nonlinear non-Foster system is better, and the equivalent reactance of the transducer is better counteracted.

In the above technical scheme:

the first lumped parameter model of the electroacoustic transducer is obtained by the following steps (M1) and (M2):

(M1) fitting according to the electric input admittance curve/electric input impedance curve to obtain a second lumped parameter model of the electroacoustic transducer, wherein the second lumped parameter model of the electroacoustic transducer is formed by a second static equivalent resistance, a static equivalent inductance and a second vibration system equivalent circuit structure which are mutually connected in series between two driving ends of the electroacoustic transducer; the first vibration system equivalent circuit structure and the second vibration system equivalent circuit structure have the same structure; the first vibration system equivalent circuit structure consists of a first vibration system equivalent resistor, a first vibration system equivalent inductor and a first vibration system equivalent capacitor which are connected in parallel; the second vibration system equivalent circuit structure consists of a second vibration system equivalent resistor, a second vibration system equivalent inductor and a second vibration system equivalent capacitor which are connected in parallel; respectively taking the resistance value of the equivalent resistor of the second vibration system, the inductance value of the equivalent inductor of the second vibration system, the capacitance value of the equivalent capacitor of the second vibration system and the resistance value of the equivalent resistor of the second static state as the resistance value of the equivalent resistor of the first vibration system, the inductance value of the equivalent inductor of the first vibration system, the capacitance value of the equivalent capacitor of the first vibration system and the resistance value of the equivalent resistor of the first static state;

(M2) for the first coefficient p, the second coefficient q, the scaling coefficient L _ex And (3) fitting:

constructing a virtual three-dimensional coordinate system oxyz, wherein the x axis, the y axis and the z axis of the virtual three-dimensional coordinate system respectively represent the fitting value of a first coefficient p, the fitting value of a second coefficient q and a proportionality coefficient L _ex The fitting value of (a); in an oxyz coordinate system, coordinate points around a first coordinate point are searched by taking the first coordinate point as a starting point, coordinate points closer to the first coordinate point to coordinate points farther from the first coordinate point are taken as a searching sequence, and the result is obtained until a fitting value of a first coefficient p, a fitting value of a second coefficient q and a proportionality coefficient L which correspond to the searched coordinate points are obtained _ex The fitting value of (c) and the model formed by the resistance value of the first vibration system equivalent resistor, the inductance value of the first vibration system equivalent inductor, the capacitance value of the first vibration system equivalent capacitor and the resistance value of the first static equivalent resistor obtained in the step (M1)The fitting error of the admittance curve/impedance curve corresponding to the model relative to the electric input admittance curve/electric input impedance curve of the electroacoustic transducer is not more than the fitting error set value, and the fitting value of the first coefficient p, the fitting value of the second coefficient q and the proportionality coefficient L corresponding to the searched coordinate points _ex Respectively corresponding to the first coefficient p, the second coefficient q, and the scaling coefficient L _ex Thereby obtaining a first lumped parameter model of the electroacoustic transducer;

wherein the first coordinate point is represented by p ₀ 、q ₀ 、L _ex0 Forming coordinate points; p is a radical of formula ₀ ＝1，q ₀ ＝0，L _ex0 Is the inductance value of the static equivalent inductance.

The applicant found in the study that if p = arbitrary value, q = arbitrary value, L _ex If the fitting is started with an arbitrary value, the fitting speed is slow and the convergence time is long. Due to p ₀ ＝1、q ₀ ＝0、L _ex0 The set of parameters and the result of step (M1) form a model (i.e. a second lumped parameter model) that is at least able to match the admittance/impedance values of the electrical input admittance/impedance curve of the electroacoustic transducer (10) at the resonance point. That is, p, q, L can be considered _ex Coordinate point and p corresponding to true value of (1) ₀ ＝1、q ₀ ＝0、L _ex0 The first coordinate points are formed relatively close. In practice, if the search is started from the first coordinate point and the order from the coordinate point closer to the first coordinate point to the coordinate point farther from the first coordinate point is used as the search order, the search efficiency can be greatly improved compared with arbitrarily taking p, q, and L _ex The values of (a) start to fit and the search time can be greatly reduced.

In the above technical scheme, Z _{Equivalence of} Is composed of

And

the ratio of (a) to (b),

is the difference between the voltage at the other end of the first capacitor and the voltage at one end of the first capacitor,

the measured power amplifier output current.

In the above technical solution,. DELTA.m is according to jX _eq (ω)-Z _{Equivalence of} Calculated by a proportional-integral regulation method.

In the invention, the PI controller is adopted to ensure that the response is quick and the adjustment without static error can be realized.

In the above technical solution, the inductance value L of the first inductor _s Capacitance value C of the first capacitor _s Satisfies the following formula:

wherein, F _n The resonance frequency of the LC filter is formed by the first inductor and the first capacitor; f _p The switching frequency of a switching tube of the single-phase full-bridge inverter is set; delta i _{ac_max} The rated output current of the power amplifier is 30% of the effective value of the rated output current of the power amplifier.

In the above technical solution, the dc power supply module is connected in parallel with a second capacitor.

The present invention also provides an electroacoustic transducer bandwidth widening apparatus using a switch-type nonlinear non-foster system, comprising a processor configured to perform the steps of the electroacoustic transducer bandwidth widening method according to any one of the above.

Compared with the prior art, the invention has the following advantages:

1. the invention uses the active matching network constructed by the nonlinear non-Foster control system to realize the broadband matching of the high-power electroacoustic emission impedance, so that the electroacoustic transducer can work in a larger bandwidth range, and meanwhile, the power application magnitude of the non-Foster control system is improved; the system carries out self-adaptive tracking on the parameters of the control system according to the mechanical dynamic impedance of the electroacoustic transmitting device, and realizes a nonlinear non-Foster matching circuit;

2. the system has good matching effect aiming at the change of impedance characteristics of the electroacoustic transmitting device, particularly the specific dynamic mechanical impedance characteristics of electroacoustic transmission, can realize the matching of impedance in a wide frequency band, and widens the application range of a non-Foster circuit matching network;

3. the system carries out real-time matching on high-power electroacoustic emission in a working state, realizes on-line automatic impedance matching, has high matching precision and good continuity, expands the output bandwidth while ensuring high-power output, greatly improves the output efficiency of an electroacoustic emission device, and has practical theoretical value and engineering significance;

4. the invention adopts a feedforward control method, can effectively reduce the detection and control quantity, improves the convergence speed and the response speed, improves the reliability of the system while reducing the cost, and has practical theoretical value and engineering significance.

Drawings

In order to more clearly illustrate the technical solutions in the embodiments of the present application, the drawings required to be used in the description of the embodiments are briefly introduced below, and it is obvious that the drawings in the description below are only some embodiments of the present application, and it is obvious for those skilled in the art to obtain other drawings based on these drawings without creative efforts.

FIG. 1 is a prior art lumped parameter model of an electroacoustic transducer with inductive input impedance and a simplified diagram thereof;

fig. 2 is a schematic diagram of the electric connection between the bandwidth widening apparatus of the electroacoustic transducer of the

embodiments

1, 2 of the present invention and the power amplifier and the electroacoustic transducer;

FIG. 3 is a schematic view of principle analysis of embodiment 1 of the present invention;

FIG. 4 is a control block diagram of embodiment 1 of the present invention;

FIG. 5 is a schematic diagram of the method steps in example 1 of the present invention;

fig. 6 is a schematic diagram of reactance curves before and after impedance matching of the electroacoustic transducer of embodiment 1 of the present invention;

FIG. 7 is a schematic view of principle analysis of embodiment 2 of the present invention;

FIG. 8 is a control block diagram of embodiment 2 of the present invention;

FIG. 9 is a schematic view of the method steps of example 2 of the present invention;

fig. 10 is a schematic diagram of reactance curves before and after impedance matching of the electroacoustic transducer according to embodiment 2 of the present invention.

Detailed Description

In order to make the objects, technical solutions and advantages of the embodiments of the present invention more apparent, the embodiments of the present invention will be described below by referring to the specific embodiments in conjunction with the accompanying drawings. It is to be understood that the embodiments described are only a few embodiments of the present invention, and not all embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention.

The existing impedance matching network can only complete matching on a single-point frequency generally, so that the broadband matching of the load cannot be realized, and the non-Foster network is used for completing the broadband matching of the load. At present, the existing non-Foster network consists of elements with limited output capacity, such as operational amplifiers and the like, and only can realize impedance matching of milliwatt-level low-power loads, such as a micro antenna and the like. In the application, the provided switch type nonlinear non-Foster system has stronger output capability, can complete impedance matching of a high-power sound source with the equal power of more than 100W of the energy transducer, and can track the output signal frequency of the power amplifier 20, thereby realizing the matching of the static impedance of the electroacoustic transducer in the whole output signal frequency range and expanding the bandwidth of the electroacoustic transducer.

When the input impedance of the electroacoustic transducer is inductive, a commonly used lumped parameter model of the electroacoustic transducer and a simplified diagram thereof are shown in fig. 1. Modeling based on motor analogyThe method is derived and comprises a driving system equivalent circuit and a vibration system equivalent circuit, wherein the two systems are connected through a transformer with a transformation ratio being an electromechanical conversion coefficient. Wherein the equivalent circuit of the driving system comprises a second static equivalent resistor 57' (with a resistance value of R) connected in series _e ) And a static equivalent inductor 56' (inductance value L) _ex0 ) The equivalent circuit of the vibration system is composed of a dynamic inductor L _m Dynamic capacitor C _m Dynamic resistance R ₁ And a load impedance Z _L Are connected in series. The lumped parameter model of the inductive transducer can also be reduced to a second lumped parameter model, i.e. formed by a second static equivalent resistor 57' (with a value of R) _e E.g., drive coil dc resistance), static equivalent inductance 56' (inductance value L) _ex0 ) And a second vibration system equivalent resistor 53' (with a resistance value of R) _mes ) And a second vibration system equivalent inductor 54' (with an inductance value of L) _mes ) And the equivalent capacitance (capacitance value is C) of the second vibration system _mes ) The five-parameter element equivalent circuit is formed. Namely, the second vibration system equivalent resistor 53', the second vibration system equivalent inductor 54' and the second vibration system equivalent capacitor 55' constitute a second vibration system equivalent circuit structure V2.

Without the addition of a matching network, the equivalent inductance of the transducer shares the power provided by the power amplifier 20, resulting in a lower output power of the transducer at multiple frequencies. The scheme of the invention is that under the condition of keeping the output voltage of the power amplifier 20 unchanged, the equivalent admittance (impedance) of the input position of the transducer is adjusted in a wider frequency range by using a non-Foster matching network, the output power of the transducer is improved, and the broadband output of the transducer is realized.

Example 1

The present embodiment 1 provides an electroacoustic transducer bandwidth widening apparatus using a switch type nonlinear non-foster system for high-power broadband electroacoustic transmission, which is disposed between a power amplifier 20 and high-power electroacoustic transmission, and includes:

the inversion module comprises a voltage-adjustable power supply, an inverter, a filter and the like, and the output end of the inversion module is connected with the power amplifier 20 and the high-power electroacoustic emission in series; the negative reactance voltage is output according to the PWM excitation pulse signal generated by the PWM modulation module, so that broadband impedance matching with high-power electroacoustic emission is realized;

a signal detection module, configured to detect a voltage and an output current at two ends of the power amplifier 20 in real time; outputting the detected signal frequency, amplitude and phase information to a phase locking module and a PWM (pulse-width modulation) calculation module for calculation processing;

the phase-locking module is used for performing real-time phase-locking calculation on the detected voltage signal, realizing angle tracking control and finally outputting the angle tracking control to the PWM calculation module;

the PWM calculation module is used for calculating the modulation degree and the modulation phase angle of the PWM and taking the modulation degree and the modulation phase angle as the input quantity of the self-adaptive correction module;

the self-adaptive correction module is used for calculating and logically judging the PWM modulation degree and the phase angle according to the mechanical dynamic impedance characteristic of the electroacoustic transmitting device, automatically adjusting the modulation degree and the phase angle in real time according to the judgment result, and using the self-adaptive correction module as a feed-forward target quantity to compensate the phase and the amplitude of the output of the inverter in real time by using a PI controller so as to achieve the optimal broadband matching effect of the high-power electroacoustic transmitting; the self-adaptive correction module finally outputs the self-adaptive correction module to the PWM modulation module;

the input end of the PWM module is connected with the self-adaptive correction module, and the output end of the PWM module is connected with the single-phase full-bridge inverter 30; and is used for outputting a proper PWM pulse to control the working state of the internal components of the single-phase full-bridge inverter 3030.

And the control panel is used for controlling the amplitude and the frequency of the output voltage of the power amplifier 20, realizing the monitoring of the waveform and the numerical value of key parameters such as current and voltage after matching and realizing the man-machine interaction.

The PI controller input is the difference between the theoretical compensation reactance and the actual compensation reactance. The modulation degree m' is controlled to change by the controller. The PI controller parameters are mainly divided into a proportional coefficient Kp and an integral coefficient Ki, and are determined by combining theoretical calculation and actual condition optimization. In practical process, the proportional coefficient and the integral coefficient can be obtained by debugging, for example, the proportional coefficient and the integral coefficient are accessed into a control system during experiment and determined according to the quality of an output waveform and the magnitude of a steady-state error. In the present invention, the proportional coefficient Kp =10 and the integral coefficient Ki =5 are preferably used in the proportional-integral adjustment. The PI controller of the invention has quick response and can carry out no-static-error adjustment.

In an ideal state, the PWM waveform obtained according to the target modulation degree and the target modulation phase angle can meet the requirement, but in practice, due to the error of the single-phase full-bridge inverter 30 control, the negative impedance may not completely cancel the static reactance of the transducer, so the alternating current output by the single-phase full-bridge inverter 30 is collected, fed back to perform compensation calculation and PI control, and then the required target modulation degree m' is calculated.

The bandwidth widening device for the electroacoustic transducer with the high-power broadband electric acoustic emission, which is provided by the scheme, can be used for switching on the power amplifier 20 by giving the amplitude and the frequency of the output voltage of the power amplifier 20 in the control panel, then detecting the voltage and the output current at two ends of the power amplifier 20 in real time by using the signal detection module, then carrying out real-time phase locking calculation on the detected voltage signal by using the phase locking module, then preliminarily calculating the modulation degree and the modulation phase angle of PWM by using the PWM calculation module according to the output data of the signal detection module and the phase locking module, then operating the self-adaptive correction module according to the mechanical dynamic impedance characteristic of the electroacoustic emitting device to automatically adjust the target modulation degree and the phase angle, and outputting corresponding negative reactance under the working condition by using the PWM modulation module and the inversion module to realize the broadband impedance matching with the high-power electroacoustic emitting device.

The PWM calculation module comprises the following calculation processes:

the nonlinear reactance of the high-power electroacoustic transducer needing to be matched is assumed to be X _eq (ω) the residual impedance amplitude phasor of the electroacoustic transmitting device after matching is complete can be expressed as

Then the power amplifier output current amplitude phasor is:

according to fig. 3, the equivalent inductance model 56 is a structure in which the nonlinear inductance model 51 and the nonlinear resistance model 52 are connected in series. The structure formed by connecting the first static equivalent resistor 57, the equivalent inductance model 56, and the first vibration system equivalent circuit structure V1 in series is equivalent to the structure formed by connecting the equivalent reactance model E1 and the equivalent resistance model E2 in series.

In the invention, the nonlinear impedance to be matched refers to a reactance value X of an equivalent reactance model E1 _eq (ω), and the residual impedance after matching is the resistance value R of the equivalent resistance model E2 _eq (ω)。

Single-phase full-bridge inverter 30 target negative reactance is-X _eq (ω), the output voltage of the single-phase full-bridge inverter 30 is:

assuming that the modulation degree of the PWM waveform is m, the AC side voltage amplitude phasor of the single-phase full-bridge inverter 30 is

The amplitude phasor of the current at the AC side is

The voltage of the DC side of the single-phase full-bridge inverter 30 is V _dc When the bridge circuit loss of the power switch tube is not counted, the voltage relationship of the alternating current side and the direct current side is as follows:

according to kirchhoff's voltage law, the ac-side loop of the single-phase full-bridge inverter 30 satisfies that the ac-side voltage is equal to the first inductor 2 (the inductance value is L) _s ) The voltage at both ends and the first capacitor 1 (the capacitance value is C) _s ) The sum of the voltages across, i.e.:

meanwhile, the voltage across the first capacitor 1 is equal to the target output voltage of the single-phase full-bridge inverter 30:

the united vertical type (1), (2), (3), (4) and (5) can be obtained:

the modulation degree (i.e., the modulation degree of the output PWM waveform) of the single-phase full-bridge inverter 30 is:

m’＝m+Δm；

wherein:

the modulation phase angle (i.e., the modulation phase angle of the output PWM waveform) of the single-phase full-bridge inverter 30 is:

referring to fig. 2, fig. 2 is a schematic structural diagram of a system where high-power electroacoustic transmission is located according to an embodiment of the present invention. Wherein the power amplifier 20 provides voltage and current signals of variable amplitude and frequency for electro-acoustic emissions. The bandwidth widening device for the high-power broadband electroacoustic transducer with the acoustic emission is connected in series between the electroacoustic emitting device and the power amplifier 20.

The electroacoustic transducer 10 input impedance is inductive, the electroacoustic transducer 10 driving signal is provided by a power amplifier 20, and the first lumped parameter model of the electroacoustic transducer 10 has a first vibration system equivalent circuit structure V1. The first lumped parameter model and the second lumped parameter model are one parametric model and the other parametric model of the electroacoustic transducer 10.

The switch type nonlinear non-Foster system comprises a first capacitor 1, a first inductor 2 and a single-phase full-bridge inverter 30 connected to the driving end of the electroacoustic transducer 10; two switching tubes of the single-phase full-bridge inverter 30 have a first common connection end PA, and the other two switching tubes have a second common connection end PB; two input ends of the single-phase full-bridge inverter 30 are respectively and correspondingly electrically connected with two output ends of the direct current power supply module 4; the first common connection end PA is electrically connected with one end of the first inductor 2, and the second common connection end PB, one end of the first capacitor 1 and one driving end of the electroacoustic transducer 10 are electrically connected with each other. The other end of the first inductor 2, the other end of the first capacitor 1 and one output end of the power amplifier 20 are electrically connected to the first electrical connection point P1, and the other output end of the power amplifier 20 and the other driving end of the electroacoustic transducer 10 are electrically connected to each other.

the modulation degree m' and the modulation phase angle theta of the single-phase full-bridge inverter (30) are calculated by the following formula, wherein

m’＝m+Δm；

Wherein the angular frequency ω has an expression ω =2 π F _s ，V _dc For the output voltage, L, of the DC supply module 4 _s 、C _s The inductance value of the first inductor 2 and the capacitance value of the first capacitor 1 are respectively; e _m 、F _s 、θ _m The amplitude, frequency, and phase angle of the output voltage signal of the power amplifier 20 are respectively.

In the first lumped parameter model of the electroacoustic transducer 10, a first static equivalent resistance 57, an equivalent inductance model 56 and a first vibration system equivalent circuit structure V1 are connected in series between two driving ends of the electroacoustic transducer 10; the equivalent inductance model 56 is an eddy current impedance model formed by connecting the nonlinear inductance model 51 and the nonlinear resistance model 52 in series.

The structure formed by connecting the first static equivalent resistor 57, the equivalent inductance model 56, and the first vibration system equivalent circuit structure V1 in series is equivalent to the structure formed by connecting the equivalent reactance model E1 and the equivalent resistance model E2 in series.

Z _r (ω)、θ _r (ω) is an impedance mode and an impedance angle corresponding to the impedance expression of the equivalent resistance model E2, respectively; x _eq (ω) is a reactance value of the equivalent reactance model E1. Δ m is according to jX _eq (ω)-Z _{Equivalence of} Calculating the obtained modulation degree feedback quantity; z _Equivalence Is a measured value of the equivalent impedance of the switch-type nonlinear foster system between the first electric connection point P1 and the second common connection point PB, and j represents an imaginary number unit.

Fig. 6 is a diagram showing the matching effect of the nonlinear non-foster system of the present invention, wherein after the matching is completed, the input reactance of the system is theoretically completely eliminated, and the control objective can be achieved. The vibration system includes a reactance that is also part of the input reactance. In the present application, the reactance of the mechanical end is also matched by a non-Foster system, so that a complete impedance match is also achieved near the transducer resonance point, which is not possible by matching only the static inductance (i.e., the static equivalent inductance 56'). By the matching method, the matching of a wider frequency band can be realized, and the maximum output power of the transducer is improved.

Referring to fig. 4, fig. 4 is a schematic structural diagram of a high-power broadband electroacoustic transducer bandwidth widening device with acoustic emission according to the present invention. The method comprises the steps of switching on the power amplifier 20 by giving the amplitude and the frequency of the output voltage of the power amplifier 20 in a control panel, detecting the voltage and the output current at two ends of the power amplifier 20 in real time by using a signal detection module, performing real-time phase-locking calculation on the detected voltage signal by using a phase-locking module, preliminarily calculating the modulation degree and the modulation phase angle of PWM by using a PWM calculation module according to the output data of the signal detection module and the phase-locking module, automatically adjusting the target modulation degree and the phase angle by operating an adaptive correction module according to the mechanical dynamic impedance characteristic of the electroacoustic transmitting device, and outputting corresponding negative reactance under the working condition by using the PWM modulation module and an inversion module to realize broadband impedance matching with the high-power electroacoustic transmitting device.

Fig. 5 is a schematic step diagram of the method for expanding the bandwidth of the electroacoustic transducer in the embodiment. When the active matching network constructed by the electroacoustic transducer bandwidth widening device based on the high-power broadband acoustic emission realizes the broadband matching of the high-power sound source impedance, the specific operation steps are as follows:

s1: before the high-power electroacoustic transmitting device is switched on to work, an impedance curve of the device under specific power needs to be measured so as to obtain impedance information under required working conditions, and theoretical calculation and parameter identification are carried out particularly on the specific dynamic mechanical impedance characteristic of electroacoustic transmission;

in the step S1, the lumped parameter model established according to the high-power electroacoustic transmission structure is also suitable for broadband electroacoustic transmission in multi-modal design, and is different from the single-mode electroacoustic transmission lumped parameter model in that a mechanical end of the lumped parameter model is formed by connecting a plurality of mechanical capacitors, mechanical inductors and mechanical resistors in parallel, and model parameters of the lumped parameter model can be obtained by theoretical calculation and fitting according to a multi-resonance-point experimental impedance curve.

The step S1 specifically includes:

s101: applying a voltage signal and a current signal to high-power electroacoustic emission according to specific power, and acquiring an electrical input impedance curve of the high-power sound source under the condition of large signal;

s102: deducing a high-power electroacoustic emission centralized parameter model based on the coupling of multiple physical fields such as electroacoustic emission electricity, mechanical electricity, sound and the like and the working principle of an internal system of the device;

in order to more accurately describe the electroacoustic emission impedance characteristics by using a lumped parameter model, a nonlinear impedance model can be used to replace elements in the model.

In the high-power emission lumped parameter model according to the embodiment of the present invention, the influence of the eddy current factor is considered, so that in the embodiment, the static equivalent inductor 56' (the inductance value is L) in the driving system _ex0 ) Using an eddy-current impedance model, i.e. a non-linear inductor model 51 (inductance value L) _em (ω)) and a nonlinear resistance model 52 (resistance value R) _em (ω)) where ω is the operating angular frequency.

When p =1,q =0, L _em (ω)＝L _ex0 。L _em And (ω) represents static inductance as a function of frequency. p =1, q =0, L _em (ω)＝L _ex0 And an algorithm is provided for searching an initial value for a set of initial value parameters extracted according to the impedance curve when the eddy current loss is not considered, and for a subsequent power-finger model for representing the eddy current loss. As can be seen from the power finger model expression, when p =1,q =0, L _em (ω)＝L _ex0 . I.e. when p =1,q =0, L _em (ω) is a frequency invariant value, which is consistent with the linear model conditions, and therefore the linear result can be taken as a special point of the non-linear model.

In the invention, because the working process of the electroacoustic transducer has a nonlinear phenomenon, but the transducer has a nonlinear phenomenon such as an eddy current phenomenon in the actual working condition, the analog adaptation degree of the linear equivalent circuit model to the transducer becomes poor. In the invention, the nonlinear inductance model 51 and the nonlinear resistance model 52 are used for replacing the static equivalent inductance 56' in the original model, thereby realizing nonlinear processing and leading the model to be closer to the working condition of the transducer.

The first lumped parameter model of the electroacoustic transducer 10 is obtained by the following steps (M1) and (M2):

(M1) fitting according to the electrical input admittance curve/electrical input impedance curve to obtain a second lumped parameter model of the electroacoustic transducer 10, where the second lumped parameter model of the electroacoustic transducer 10 is formed by a second static equivalent resistance 57', a static equivalent inductance 56', and a second vibration system equivalent circuit V2 structure, which are mutually connected in series between two driving ends of the electroacoustic transducer 10; the first vibration system equivalent circuit structure V1 and the second vibration system equivalent circuit structure V2 have the same structure; the first vibration system equivalent circuit structure V1 is composed of a first vibration system equivalent resistor 53, a first vibration system equivalent inductor 54, and a first vibration system equivalent capacitor 55, which are connected in parallel; the second vibration system equivalent circuit structure V2 is composed of a second vibration system equivalent resistor 53', a second vibration system equivalent inductor 54', and a second vibration system equivalent capacitor 55' which are connected in parallel; taking the resistance value of the second vibration system equivalent resistor 53', the inductance value of the second vibration system equivalent inductor 54', the capacitance value of the second vibration system equivalent capacitor 55', and the resistance value of the second static equivalent resistor 57' as the resistance value of the first vibration system equivalent resistor 53, the inductance value of the first vibration system equivalent inductor 54, the capacitance value of the first vibration system equivalent capacitor 55, and the resistance value of the first static equivalent resistor 57, respectively;

constructing a virtual three-dimensional coordinate system oxyz, wherein the x-axis, the y-axis and the z-axis of the virtual three-dimensional coordinate system respectively represent the fitting value of a first coefficient p, the fitting value of a second coefficient q and a proportionality coefficient L _ex The fitting value of (a); in an oxyz coordinate system, coordinate points around a first coordinate point are searched by taking the first coordinate point as a starting point, coordinate points closer to the first coordinate point to coordinate points farther from the first coordinate point are taken as a searching sequence, and the result is obtained until a fitting value of a first coefficient p, a fitting value of a second coefficient q and a proportionality coefficient L which correspond to the searched coordinate points are obtained _ex A model (i.e., a fitting value according to p, a fitting value according to q, L) is formed by the fitting value of (c) and the resistance value of the first vibration system equivalent resistor 53, the inductance value of the first vibration system equivalent inductor 54, the capacitance value of the first vibration system equivalent capacitor 55, and the resistance value of the first static equivalent resistor 57 obtained in the step (M1) _ex The fitting value of the first coefficient p, the fitting value of the second coefficient q, and the proportionality coefficient L corresponding to the searched coordinate point are used to obtain a nonlinear inductance model 51 and a nonlinear resistance model 52, and then the model formed by combining the resistance value of the first vibration system equivalent resistor 53, the inductance value of the first vibration system equivalent inductor 54, the capacitance value of the first vibration system equivalent capacitor 55, the resistance value of the first static equivalent resistor 57, which are obtained in the step (M1) is combined with the fitting error of the admittance curve/impedance curve corresponding to the electric input admittance curve/electric input impedance curve of the electroacoustic transducer 10, which is not greater than the fitting error set value, and the fitting value of the first coefficient p, the fitting value of the second coefficient q, and the proportionality coefficient L corresponding to the searched coordinate point are used to obtain the fitting value of the first coefficient p, the fitting value of the second coefficient q, and the proportionality coefficient L _ex Respectively corresponding to the first coefficient p, the second coefficient q, and the scaling coefficient L _ex Thereby obtaining a first lumped parameter model of said electro-acoustic transducer 10;

wherein the first coordinate point is represented by p ₀ 、q ₀ 、L _ex0 Forming coordinate points; p is a radical of formula ₀ ＝1，q ₀ ＝0，L _ex0 Is the inductance value of the static equivalent inductance 56'.

In this application, in step (M1), the resistance value of the first vibration system equivalent resistor 53, the inductance value of the first vibration system equivalent inductor 54, the capacitance value of the first vibration system equivalent capacitor 55, and the resistance value of the first static equivalent resistor 57 are obtained first, that is, each parameter value of the first vibration system equivalent circuit structure in the first centralized parameter model and the resistance value of the first static equivalent resistor are obtained according to each parameter value of the second vibration system equivalent circuit structure in the second centralized parameter model and the resistance value of the first static equivalent resistor. In the step (M2), a fitting value of the first coefficient p, a fitting value of the second coefficient q and a proportionality coefficient L are obtained _ex The fitting values of (a) and (b) are obtained by the step (M2), and the expressions of the nonlinear inductance model 51 and the nonlinear resistance model 52 are obtained. And (5) combining the step (M1) and the step (M2) to obtain a first lumped parameter model.

S103: firstly, preliminarily calculating the parameter values of the lumped parameter model impedance elements of the device according to the frequency of the electroacoustic emission resonance point and a theoretical calculation method of the corresponding impedance of the electroacoustic emission resonance point;

in step S103, the method for calculating the electroacoustic transmission resonant point frequency and the corresponding impedance theory mainly includes:

(1) When the operating frequency F _s When the frequency is not less than 0Hz, the starting point of the amplitude of the impedance curve is the high-power electroacoustic emission driving direct-current resistor R _e (i.e., the value of the first static equivalent resistor 57).

(2) Resonant frequency when high power electroacoustic emission occurs series resonance

Only the static equivalent inductor 56' (with inductance L) remains in the circuit _ex0 ) I.e. when F _s ＝f _s Corresponding reactance value of ω _s L _ex0 。

(3) Resonant frequency when parallel resonance of high power electroacoustic emission occurs

At this time, the reactance value of the transmission is theoreticallyZero, only the first static equivalent resistor 57 (with a resistance of R) remains in the circuit _e ) First vibration system equivalent resistor 53 (resistance R) _mes ) When F is _s ＝f _p Corresponding resistance value of R _e +R _mes 。

The initial values of the element parameters in the centralized parameter model of the transmitting device can be obtained by a theoretical calculation method as shown in the following table.

TABLE 1

Parameters of elements	R _e /Ω	R _mes /Ω	L _mes /mH	C _mes /mF	L _ex0 /mH
						Fitting parameters	7.4	34	32.6	0.318	4.2

S104: and taking the element impedance parameter values theoretically calculated in the step S103 as initial parameters of a lumped parameter model identification algorithm, and fitting the element parameter values in the model according to an electrical input impedance experiment curve.

In the process of fitting the parameters of the model elements, the eddy current impedance model has practical physical significance and can accurately simulate the variation trend of the eddy current impedance along with the frequency. Therefore, in the embodiment, the power-exponent model is selected to fit the eddy current impedance model in the high-power electroacoustic emission model, and the expression of the power-exponent model is as follows:

L _ex the fitting of p and q is also obtained by the above method, i.e., the method in which the operating frequency is different and the corresponding impedance is different (for example, if Fs =0Hz, the impedance is Re), i.e., the model parameters are obtained from predetermined mathematical models by using the impedance values at different frequencies.

In actual calculation, p, q, L _ex The number of combinations formed is very large. p =1, q =0, L _ex Where 4.2 is one, this set of parameters can fit the impedance value at the upper resonance point, but the area after the resonance point is very poor. Thus, in practice, this point can be brought into the range of forward and backward (p, q, L close to this set of values _ex ) Parameter trials were performed to finally obtain p =0.7483, q =0.0398 _ex Set of satisfactory parameters (fitting error less than set fitting error) 43.9. If from p = arbitrary, q = arbitrary, L _ex If the fitting is carried out by using the value of = arbitrary, the fitting speed is slow, and the convergence time is long. For example, the least squares method is used to solve for each parameter p, q, L _ex When the least square method solves that the error of the impedance curve fitted under the parameter is within the set error compared with the actual curve, the parameter is derived.

Preferably, in step S104, a fitting algorithm such as a least square algorithm, a particle swarm algorithm, or the like may be selected to fit the electrical input impedance curve, but the fitting algorithm is not limited to these two algorithms.

And fitting the parameter values of the model elements according to the electric input impedance curve by using a fitting algorithm, wherein when the fitting error is smaller than a set value, the fitting parameters of the elements are shown in the following table.

TABLE 2

S2: when the high-power electroacoustic transmission works, the power amplifier 20 can be switched on, and the amplitude and the frequency of the output voltage of the power amplifier 20 and the voltage of the voltage-adjustable power supply are set (for example, the voltage can be set through a control panel);

s3: the bandwidth widening device of the electroacoustic transducer starts to work, and the impedance matching among a power amplifier, a high-power electroacoustic transmitting and controlling system is realized by utilizing the transformation of output negative reactance, so that the maximum active power output of the device is ensured all the time;

referring to fig. 3, fig. 3 is a schematic diagram of an active matching network where the high-power broadband electroacoustic transducer bandwidth widening device with acoustic emission is located in the present embodiment. Inverter port output voltage

With equivalent reactance model E1 to be matched (reactance value X) _eq (ω)) voltage across terminals

The relationship between them is:

current flowing into inverter from inverter port

The currents flowing through the equivalent reactance model E1 to be matched are all

Due to inverter output impedance

The impedance value corresponding to the reactance to be matched is determined by the voltage V at two ends of the equivalent inductor _L And current I _L Is calculated to obtain

So that Z can be obtained _non ＝-Z _eq (ω) to achieve a perfect match to the high power electro-acoustic transmit impedance.

The step S3 specifically includes:

s301: according to the element parameter values in the electroacoustic emission lumped parameter model, a negative reactance value required to be simulated at an output port of the inverter is given;

s302: by controlling the PWM phase and the modulation degree, the output voltage of the inverter is kept as negative reactance voltage, and therefore broadband matching of the transmitting impedance is achieved.

S303: the signal detection module is used for detecting the voltage signal of the power amplifier 20 in real time to obtain the amplitude value E of the output voltage of the power amplifier _m Frequency of F _s Initial phase angle of theta _m I.e. the amplitude phasor of the output voltage of the power amplifier is

S304: performing phase-locking calculation on the detected voltage signal by using a phase-locking module, performing tracking control on the angle, and finally outputting the angle to the PWM calculation module;

s305: calculating the modulation degree and the modulation phase angle of the PWM by using a PWM calculation module according to the impedance characteristic of the electroacoustic transmitting device to obtain a modulation degree m and a modulation phase angle theta;

s306: and inputting the calculation result into a self-adaptive correction module, logically judging the input quantity according to the mechanical dynamic impedance characteristic of the electroacoustic transmitting device, and automatically adjusting the modulation degree and the phase angle in real time according to the judgment result.

When m is larger than 0, controlling the self-adaptive control module to output the modulation degree and the phase angle unchanged; when m is less than 0, the output modulation degree is controlled to be the absolute value of m, and the modulation phase angle is increased by pi.

Because the conventional controller generally cannot output a negative value, i.e., the modulation degree cannot output a negative value. In the invention, when m is greater than 0, the modulation degree and the phase angle of the output can be controlled to be unchanged. When m is less than 0, the modulation degree can be output as the absolute value (positive value) of m, and the modulation phase angle can be output as theta + pi, thereby facilitating the output of signals.

S307: the PWM modulation module outputs proper PWM pulses according to the calculation and control results in the steps S304, S305 and S306, controls the working state of the nonlinear Foster matching system, and keeps the output voltage of the inverter as negative reactance voltage, thereby realizing the online broadband matching of the high-power electroacoustic emission impedance.

Preferably, in order to filter out higher harmonics around the switching frequency, the inverter module adopts an LC filter (i.e., a filter formed by the first capacitor 1 and the first inductor 2) according to the principles shown in equations (6) and (7):

wherein, F _n Is the resonant frequency of the LC filter; f _p A carrier frequency of PWM; Δ i _{ac_max} In order to make the power amplifier output 30% of the effective value of the rated current, in this embodiment, the inductance value L of the first inductor 2 may be selected _s Is 2mH; capacitance C of the first capacitor 1 _s 10 μ F.

S4: when the working condition of the high-power electroacoustic transmission changes (for example, the working condition can be adjusted through the control panel), the step S3 is repeated, and the self-adaptive tracking is carried out according to the mechanical dynamic impedance of the electroacoustic transmission device, so that the control system always outputs the corresponding negative reactive voltage, and the device is ensured to always output the maximum active power.

Referring to fig. 6, fig. 6 is a schematic diagram of reactance curves before and after impedance matching of high-power electroacoustic transmission according to an embodiment of the present invention. Wherein, the solid line (—) represents the output reactance curve of the power amplifier before matching, and it can be seen from the figure that the output reactance of the power amplifier increases rapidly with the increase of the frequency, which severely limits the broadband output of high-power transmission; the dotted line (· -) represents the output reactance curve of the switch type nonlinear non-foster system, and it can be seen that the invention completely eliminates the influence of mechanical reactance on the basis of eliminating static reactance of the non-foster system, and realizes the output target of the system; when the switch-type nonlinear non-Foster system is introduced, an output reactance curve between a point P1 and the negative output end of the power amplifier 20 is shown as a dotted line (—) in fig. 6, and under the condition of completely eliminating the transmitting static reactance and the transmitting dynamic reactance, the output reactance of the power amplifier can be always kept to be zero, so that the transmitting output efficiency is improved, and the output bandwidth of high-power transmission is expanded.

In the present embodiment 1, the nonlinear non-foster system matches both the static reactance (i.e., the reactance component in the equivalent inductance model) and the mechanical reactance (i.e., the reactance component in the equivalent circuit configuration of the first vibration system), and corresponds to the result of fig. 6.

The invention provides a bandwidth widening device of an electroacoustic transducer for high-power broadband electro-acoustic emission and an implementation method thereof.

The nonlinear non-Foster system can theoretically realize infinite bandwidth, and breaks through the limitation of gain-bandwidth theory on high-power electroacoustic emission bandwidth; compared with a common non-Foster system based on an operational amplifier, the power capacity of the system can be improved by more than 1000 times, and the power application magnitude of a non-Foster circuit matching network is improved; the system has good matching effect aiming at the change of impedance characteristics of electroacoustic transmission under different frequencies, particularly the characteristic dynamic mechanical impedance characteristics of the electroacoustic transmission, can realize the complete matching of impedance in a wide frequency band, and widens the application range of a non-Foster circuit matching network; the system carries out real-time matching on the high-power electroacoustic emission in a working state, has high matching precision and good continuity, expands the output bandwidth while ensuring the high-power output, greatly improves the output efficiency of the electroacoustic emission device, and has practical theoretical value and engineering significance.

In example 1, the compensated reactance model and the residual impedance model correspond to an equivalent reactance model (E1) and an equivalent resistance model (E2), respectively

Example 2

In example 2, U _eq (ω)＝L _em (ω) the compensated reactance model is a nonlinear inductance model 51, and the first static equivalent resistance 57, the nonlinear resistance model 52, and the first vibration system equivalent circuit configuration V1 constitute the residual impedance model. The first lumped parameter model of this embodiment 2 is the same as that of embodiment 1, and is composed of a first static equivalent resistor 57, a nonlinear inductance model 51, a nonlinear resistance model 52, and a first vibration system equivalent circuit structure V1, which are connected in series between two driving terminals of the electroacoustic transducer 10.

The difference between the embodiment 2 and the embodiment 1 is mainly the calculation of m, the formula and L _em (ω)、R _em For the calculation of (ω), the rest can refer to example 1.

Fig. 8 is a control block diagram of embodiment 2 of the present invention. The bandwidth widening device of the transducer comprises an equivalent circuit model building module, a model identification module, an inversion module, a signal detection module, a phase locking module, a PWM calculation module and a PWM modulation module. The equivalent circuit model building module deduces a nonlinear equivalent circuit model of the high-power transducer based on the electro-mechanical-acoustic multi-physical field coupling of the electroacoustic transducer, the working principle of an internal system of the transducer and the nonlinear impedance model; the model identification module is based on the nonlinear model of the transducer, and according to high-power impedance test experimental data, the parameters of each element of the nonlinear model are determined by adopting methods such as theoretical calculation, parameter identification and the like; the input end of the inversion module is connected with a direct-current voltage source, and the output end of the inversion module is connected with a power amplifier and the high-power sound source in series; the signal detection module is used for detecting the voltage and the output current at two ends of the power amplifier; the phase-locking module is used for performing phase-locking calculation on the detected signal, performing angle tracking control and finally outputting the angle tracking control to the PWM calculation module; the PWM calculation module is used for calculating a target modulation degree and a phase angle of PWM according to the identification parameter result of the model identification module, taking the target modulation degree and the phase angle as a feedforward quantity, using a PI controller to compensate the phase and the amplitude of the output of the inverter, and finally outputting the phase and the amplitude to the PWM modulation module; the input end of the PWM module is connected with the PWM calculation module, and the output end of the PWM module is connected with the inverter module and used for outputting proper PWM pulses to control the working state of internal elements of the inverter.

Fig. 9 is a schematic diagram of specific operation steps of the active matching network constructed by the transducer bandwidth widening apparatus for high-power electroacoustic transducer to implement broadband matching on the nonlinear impedance of the high-power sound source, where the specific steps are as follows:

s1: applying high-power voltage signals and current signals to an electroacoustic transducer under different working conditions so as to obtain an electrical input impedance curve of the transducer;

s1, applying high-power excitation signals to the transducer under different frequencies, and obtaining impedance amplitude and phase angle of the transducer under each frequency to obtain a relation curve of the resistance and reactance of the transducer with respect to the frequency;

s2: building a high-power nonlinear equivalent circuit model of the electroacoustic transducer through the system equivalent circuit model building module in the technical scheme 1, obtaining a total impedance expression based on frequency-dependent mathematical models of various impedances in the nonlinear impedance model, and determining parameter values of various elements in the nonlinear model through the parameter identification module according to the impedance test data in the step S1;

s201: constructing a linear centralized parameter model of the high-power electroacoustic transducer based on an electroacoustic multi-physical field analogy method of the electroacoustic transducer and the working principle of an internal system of the transducer;

s202: according to the resonance point frequency of the electroacoustic transducer and a theoretical calculation method of the corresponding impedance of the resonance point frequency, the parameters of the linear lumped parameter model element of the electroacoustic transducer are preliminarily calculated;

the element parameters in the transducer linear model can be obtained by a theoretical calculation method as shown in the table below.

TABLE 3 second lumped parameter model element parameter fit values

Parameters of elements	R _e /Ω	R _mes /Ω	L _mes /mH	C _mes /mF	L _ex0 /mH
						Fitting value	7.4	34	32.6	0.318	4.2

S203: deducing a high-power nonlinear equivalent circuit model of the electroacoustic transducer by combining the influence of eddy nonlinearity generated by high-frequency work on the impedance characteristic of the transducer and a frequency-dependent mathematical model of various impedances in a nonlinear impedance model based on the coupling of the electroacoustic transducer with multi-physical fields such as electricity, machine, sound and the like and the working principle of an internal system of the transducer;

the effect of electroacoustic transducer nonlinearity on the transducer impedance characteristics includes: under the condition of large-signal excitation, various parameters and conversion coefficients of active components in the transducer are greatly changed under the condition of high frequency, so that the impedance parameters in a centralized parameter model of the transducer are changed.

Compared with the linear model in the step S201, the high-power nonlinear equivalent circuit model of the electroacoustic transducer has the advantage that the linear elements in the equivalent circuit of the driving system and the vibration system are replaced by the nonlinear impedance model.

The frequency-dependent mathematical model considering the nonlinearity of eddy currents generated at high frequencies selects an eddy current impedance model combining a power exponent model and a polynomial model. The model has practical physical significance, and can accurately simulate the variation trend of the eddy current impedance along with the frequency as follows:

wherein p (ω) = a ₀ +a ₁ ω+a ₂ ω ² ，q(ω)＝a ₃ +a ₄ ω+a ₅ ω ² 。

In calculating L _ex 、a ₀ 、a ₁ 、a ₂ 、a ₃ 、a ₄ 、a ₅ When the value of (1) is used, the same calculation as that of example 1 can be used to calculate L _ex P, q are similar methods, i.e. first find each L that fits the impedance at the upper resonance point _ex 、a ₀ 、a ₁ 、a ₂ 、a ₃ 、a ₄ 、a ₅ The numerical value (as the initial value of search), and then with the initial value of search as the starting point, gradually expand the search range around until the obtained impedance curve fitted by each parameter and the actual curve have the error within the set error, then derive each parameter L _ex 、a ₀ 、a ₁ 、a ₂ 、a ₃ 、a ₄ 、a ₅ The value of (a) is,

s204: and (2) taking the element parameter values in the step (S202) as initialization parameters of the nonlinear model of the transducer in the step (S203), and determining the element parameter values in the nonlinear model through an identification algorithm according to the frequency-dependent electrical input impedance experimental curve measured in the step (S1). The parameters associated with the nonlinear model obtained by least squares fitting are shown in the following table:

TABLE 4 first lumped parameter model element parameter fit values

Aiming at possible factors which can cause the working of the transducer to present nonlinearity in the working process, the specific method for determining the parameters of each element in the nonlinear impedance model through an identification algorithm according to the electric input impedance experimental curve comprises the following steps: when the transducer works under a high-frequency excitation condition, eddy current nonlinearity occurs, and the impedance parameter in the transducer lumped parameter model generates nonlinear change. And (2) obtaining a relation curve of the resistance and the reactance of the transducer with respect to the frequency according to the impedance amplitude and the phase angle of the transducer under different frequencies measured in the step (S1), obtaining a total impedance expression based on frequency-related mathematical models of various impedances in the nonlinear impedance model, and fitting the total impedance expression with the measured impedance curve to obtain various impedance parameters in the nonlinear impedance model when the transducer works at a high frequency.

The factor-dependent mathematical model may be in the form of a polynomial model, a power exponent model, a logarithmic function model, or the like, but is not limited to these models.

In the step S2, the lumped parameter model established according to the high-power electroacoustic transducer structure is also applicable to a multimode-designed broadband electroacoustic transducer, and is different from the single-mode electroacoustic transducer lumped parameter model in that the mechanical end of the lumped parameter model is formed by connecting a plurality of mechanical capacitors, mechanical inductors and mechanical resistors in parallel, and the model parameters can be obtained by theoretical calculation and fitting according to the experimental impedance curve of multiple resonance points.

S3: according to the nonlinear reactance in the nonlinear equivalent circuit model in the step S2, a negative reactance value required to be simulated by the output port of the non-Foster system is given;

s4: the PWM phase and the modulation degree under the corresponding working condition are obtained by operating each module of the non-Foster system in the technical scheme 1, and the output voltage of the system is kept as negative reactance voltage, so that the broadband matching of the nonlinear impedance of the transducer is realized.

S401: the signal detection module is used for detecting the voltage signal of the power amplifier to obtain the amplitude value E of the output voltage of the power amplifier _m Frequency of F _s Initial phase angle of theta _m I.e. the amplitude phasor of the output voltage of the power amplifier is

S402: performing phase-locking calculation on the detected voltage signal by using a phase-locking module, performing tracking control on the angle, and finally outputting the angle to the PWM calculation module;

s403: the PWM calculation module is used for calculating a target modulation degree and a phase angle of PWM according to the identification parameter result of the model identification module, using a PI controller to compensate the phase and amplitude of the output of the inverter by taking the target modulation degree and the phase angle as a feedforward quantity, and outputting the phase and amplitude to the PWM modulation module;

the PWM calculation module comprises the following calculation processes:

Then the power amplifier output current amplitude phasor is:

referring to fig. 7, the equivalent inductance model 56 is a structure in which the nonlinear inductance model 51 and the nonlinear resistance model 52 are connected in series. The structure formed by connecting the first static equivalent resistor 57, the equivalent inductance model 56, and the first vibration system equivalent circuit structure V1 in series is equivalent to the structure formed by connecting the equivalent reactance model E1 and the equivalent resistance model E2 in series.

In the present invention, the nonlinear impedance to be matched means a reactance value L of the equivalent reactance model E1 _em (ω), and the residual impedance amplitude phasor after matching can be expressed as

Single-phase full-bridge inverter 30 target negative reactance is-L _em (ω), the output voltage of the single-phase full-bridge inverter 30 is:

The amplitude phasor of the AC side current is

according to kirchhoff's voltage law, the ac-side loop of the single-phase full-bridge inverter 30 satisfies that the ac-side voltage is equal to the first inductor 2 (the inductance value is L) _s ) The voltage at both ends and the first capacitor 1 (the capacitance value is C) _s ) The sum of the voltages across them, i.e.:

the united vertical type (1), (2), (3), (4) and (5) can be obtained:

the modulation phase angle of the single-phase full-bridge inverter 30 (i.e., the modulation phase angle of the output PWM waveform) is:

namely:

PWM target modulation degree: m' = m + Δ m; wherein,

PWM target modulation phase angle:

s404: and the PWM modulation module outputs a proper PWM pulse according to the calculation and control results in the step S402 and the step S403, controls the working state of the switch type non-Foster feedforward control system, and keeps the output voltage of the inverter as negative reactance voltage, thereby realizing the matching of the nonlinear impedance of the high-power electroacoustic transducer.

Fig. 7 is a schematic diagram of an active matching network of the transducer bandwidth widening apparatus for the neutralization of nonlinearity of the high-power electroacoustic transducer in the present embodiment. Inverter port output voltage V _non And a nonlinear reactance L _em Voltage V across (omega) _L The relationship between them is: v _non ＝-V _L (ii) a Current I flowing into the inverter from the inverter port _non And a current through nonlinear reactance L _em Current I of (omega) _L The relationship between them is: I.C. A _non ＝I _L (ii) a Due to the inverter output impedance Z _non ＝V _non /I _non Nonlinear reactance Z _L ＝V _L /I _L So that Z can be obtained _non ＝-Z _L Thereby achieving broadband matching of the nonlinear impedance of the transducer.

The invention provides a device for widening the bandwidth of a transducer considering the nonlinearity of a high-power electroacoustic transducer and an implementation method thereof. By constructing a nonlinear equivalent circuit model of the electroacoustic transducer under the condition of high power, the system can track the nonlinear impedance of the transducer more accurately and quickly and provide negative reactance, thereby realizing the broadband matching of the high-power electroacoustic transducer; the system can theoretically realize infinite bandwidth, and breaks through the limitation of gain-bandwidth theory on the bandwidth of a high-power electroacoustic transducer; compared with a common non-Foster system based on an operational amplifier, the power capacity of the system can be improved by more than 1000 times, the power application magnitude of a non-Foster circuit matching network is improved, and the application range of the non-Foster system is widened; the invention ensures the high-power output of the energy converter, simultaneously expands the output bandwidth, greatly improves the output efficiency of the electroacoustic transducer, and has practical theoretical value, engineering significance and wide applicability.

Referring to fig. 10, fig. 10 is a control effect diagram of the transducer bandwidth widening method according to the present invention. Wherein, the solid line (—) represents the output reactance curve of the power amplifier before matching, and it can be seen from the figure that the output reactance of the power amplifier increases rapidly with the increase of the frequency, which severely limits the broadband output of the high-power transducer; the dotted line (· -) represents the power amplifier output reactance curve after the matching of the active matching network formed by the transducer bandwidth widening device based on the linear model of the high-power electroacoustic transducer, and as can be seen from fig. 10, although the matching effect of the linear system is obvious, the static inductance is not completely matched due to the influence of nonlinear factors such as the eddy current effect and the like; the dotted line (· -) represents the output reactance curve of the switching nonlinear non-foster system introduced in the electroacoustic transducer system, and it can be seen that the negative reactance of the system output is not linear with the frequency, and the output target of the system is realized; after the control system is introduced, namely the output reactance (namely the reactance between the point P1 and the negative output end of the power amplifier 20) curve of the power amplifier is shown as a dotted line (-) in the figure, under the condition of completely eliminating the nonlinear static reactance of the transducer, the output reactance of the power amplifier under the high-frequency condition can be kept close to zero, so that the output efficiency of the transducer is improved, and the output bandwidth of the high-power transducer is expanded.

In the present embodiment 2, the nonlinear non-foster system matches the static reactance in fig. 7 (i.e., the reactance portion in the equivalent inductance model of the element 51 shown in fig. 7) which is an equivalent circuit considering the nonlinear eddy current effect, corresponding to the result of fig. 10.

It should be noted that, in this specification, each embodiment is described in a progressive manner, and each embodiment focuses on differences from other embodiments, and portions that are the same as and similar to each other in each embodiment may be referred to.

The embodiments of the present invention have been described in detail, but the description is only for the preferred embodiments of the present invention and should not be construed as limiting the scope of the present invention. All equivalent changes and modifications made within the scope of the present invention should be covered by this patent. Various equivalent modifications of the invention, which fall within the scope of the appended claims of this application, will be suggested to those skilled in the art after reading this disclosure. The embodiments and features of the embodiments of the present invention may be combined with each other without conflict.

Claims

1. An electroacoustic transducer bandwidth widening method using a switch-type nonlinear non-foster system, wherein the input impedance of the electroacoustic transducer (10) is inductive, the driving signal of the electroacoustic transducer (10) is provided by a power amplifier (20), the electroacoustic transducer (10) is provided with a first lumped parameter model, and the first lumped parameter model is composed of a first static equivalent resistance (57), an equivalent inductance model (56) and a first vibration system equivalent circuit structure (V1) which are mutually connected in series between two driving ends of the electroacoustic transducer (10);

the switching nonlinear non-Foster system is characterized by comprising a first capacitor (1), a first inductor (2) and a single-phase full-bridge inverter (30) connected to the driving end of the electroacoustic transducer (10); two switching tubes of the single-phase full-bridge inverter (30) are provided with a first common connection end (PA), and the other two switching tubes are provided with a second common connection end (PB); two input ends of the single-phase full-bridge inverter (30) are respectively and correspondingly electrically connected with two output ends of the direct-current power supply module (4); the first common connecting end (PA) is electrically connected with one end of the first inductor (2), and the second common connecting end (PB), one end of the first capacitor (1) and one driving end of the electroacoustic transducer (10) are electrically connected with each other; the other end of the first inductor (2), the other end of the first capacitor (1) and one output end of the power amplifier (20) are electrically connected with a first electric connection point (P1), and the other output end of the power amplifier (20) and the other driving end of the electroacoustic transducer (10) are electrically connected with each other;

the modulation degree m' and the modulation phase angle theta of the single-phase full-bridge inverter (30) are calculated by the following formula

m’＝m+Δm；

Wherein the expression of angular frequency ω is ω =2 π F _s ，V _dc Is the output voltage, L, of the DC supply module (4) _s 、C _s The inductance value of the first inductor (2) and the capacitance value of the first capacitor (1) are respectively; e _m 、F _s 、θ _m The amplitude, frequency and phase angle of the output voltage signal of the power amplifier (20) are respectively;

the equivalent inductance model (56) is an eddy current impedance model formed by mutually connecting a nonlinear inductance model (51) and a nonlinear resistance model (52) in series;

U _eq (omega) being compensated reactanceRatio of reactance value of model to angular frequency ω, Z _r (ω)、θ _r (ω) is the impedance mode and the impedance angle corresponding to the impedance expression of the residual impedance model, respectively;

the compensated reactance model and a residual impedance model are connected in series between two driving ends of an electroacoustic transducer (10) so as to form an equivalent structure of the first lumped parameter model;

Δ m is according to j ω U _eq (ω)-Z _Equivalence Calculating the obtained modulation degree feedback quantity; z _{Equivalence of} Is the measured value of the equivalent impedance of the switch type nonlinear Foster system between a first electric connection point (P1) and a second public connection end (PB), wherein j represents an imaginary number unit;

(A) The first lumped parameter model is equivalent to a structure formed by connecting an equivalent reactance model (E1) and an equivalent resistance model (E2) in series, and the compensated reactance model and the residual impedance model respectively correspond to the equivalent reactance model (E1) and the equivalent resistance model (E2);

X _eq (ω) is a reactance value of the equivalent reactance model (E1);

(B)U _eq (ω)＝L _em (ω) the compensated reactance model is a nonlinear inductance model (51), and the first static equivalent resistance (57), the nonlinear resistance model (52), and a first vibration system equivalent circuit structure (V1) constitute the residual impedance model.

2. The method for broadening the bandwidth of an electroacoustic transducer of claim 1, wherein: an inductance value L of the nonlinear inductance model (51) _em (omega) resistance value R of nonlinear resistance model (52) _em The expression of (ω) is:

or

3. The method for broadening the bandwidth of an electroacoustic transducer as claimed in claim 2, wherein: an excitation signal is applied to the driving end of the electroacoustic transducer (10), an electric input admittance curve/electric input impedance curve of the electroacoustic transducer (10) is obtained, and a first lumped parameter model of the electroacoustic transducer (10) is obtained according to the electric input admittance curve/electric input impedance curve fitting.

4. The method for broadening the bandwidth of an electroacoustic transducer as claimed in claim 3, wherein:

the first lumped parameter model of the electroacoustic transducer (10) is obtained by the following steps (M1) and (M2):

(M1) fitting according to the electric input admittance curve/electric input impedance curve to obtain a second lumped parameter model of the electroacoustic transducer (10), wherein the second lumped parameter model of the electroacoustic transducer (10) is formed by a second static equivalent resistance (57 '), a static equivalent inductance (56') and a second vibration system equivalent circuit (V2) which are mutually connected in series between two driving ends of the electroacoustic transducer (10); the first vibration system equivalent circuit structure (V1) and the second vibration system equivalent circuit structure (V2) have the same structure; the first vibration system equivalent circuit structure (V1) consists of a first vibration system equivalent resistor (53), a first vibration system equivalent inductor (54) and a first vibration system equivalent capacitor (55) which are connected in parallel; the second vibration system equivalent circuit structure (V2) is composed of a second vibration system equivalent resistor (53 '), a second vibration system equivalent inductor (54 ') and a second vibration system equivalent capacitor (55 ') which are connected in parallel; respectively taking the resistance value of the second vibration system equivalent resistor (53 '), the inductance value of the second vibration system equivalent inductor (54'), the capacitance value of the second vibration system equivalent capacitor (55 '), and the resistance value of the second static equivalent resistor (57') as the resistance value of the first vibration system equivalent resistor (53), the inductance value of the first vibration system equivalent inductor (54), the capacitance value of the first vibration system equivalent capacitor (55), and the resistance value of the first static equivalent resistor (57);

constructing a virtual three-dimensional coordinate system oxyz, wherein the x axis, the y axis and the z axis of the virtual three-dimensional coordinate system respectively represent the fitting value of a first coefficient p, the fitting value of a second coefficient q and a proportionality coefficient L _ex The fitting value of (a); in an oxyz coordinate system, coordinate points around a first coordinate point are searched by taking the first coordinate point as a starting point, coordinate points closer to the first coordinate point to coordinate points farther from the first coordinate point are taken as a searching sequence, and the result is obtained until a fitting value of a first coefficient p, a fitting value of a second coefficient q and a proportionality coefficient L which correspond to the searched coordinate points are obtained _ex The fitting error of the admittance curve/impedance curve corresponding to the model consisting of the fitting value of (1), the resistance value of the first vibration system equivalent resistor (53), the inductance value of the first vibration system equivalent inductor (54), the capacitance value of the first vibration system equivalent capacitor (55) and the resistance value of the first static equivalent resistor (57) relative to the electric input admittance curve/electric input impedance curve of the electroacoustic transducer (10) is not more than the fitting error set value, and the fitting value of the first coefficient p, the fitting value of the second coefficient q and the proportionality coefficient L corresponding to the searched coordinate points are used _ex Respectively corresponding to the first coefficient p, the second coefficient q, and the scaling coefficient L _ex Thereby obtaining a first lumped parameter model of the electroacoustic transducer (10);

wherein the first coordinate point is represented by p ₀ 、q ₀ 、L _ex0 Forming coordinate points; p is a radical of ₀ ＝1，q ₀ ＝0，L _ex0 Is the inductance value of the static equivalent inductance (56').

5. According to the claimsThe method for broadening the bandwidth of the electroacoustic transducer is characterized in that: z is a linear or branched member _{Equivalence of} Is composed of

And with

The ratio of (a) to (b),

is the difference between the voltage measured at the other end of the first capacitor (1) and the voltage measured at one end of the first capacitor (1),

a current is output for the measured power amplifier (20).

6. The method for broadening the bandwidth of an electroacoustic transducer as claimed in any one of claims 1 to 4, wherein: Δ m is according to j ω U _eq (ω)-Z _{Equivalence of} Calculated by a proportional-integral regulation method.

7. The method for broadening the bandwidth of an electroacoustic transducer as claimed in any one of claims 1 to 4, wherein: inductance L of the first inductor (2) _s A capacitance value C of the first capacitor (1) _s Satisfies the following formula:

wherein, F _n The resonance frequency of an LC filter formed by a first inductor (2) and a first capacitor (1); f _p The switching frequency of a switching tube of the single-phase full-bridge inverter (30); delta i _{ac_max} Is that theThe power amplifier (20) is rated to output 30% of the effective value of the current.

8. The method for broadening the bandwidth of an electroacoustic transducer as claimed in any one of claims 1 to 4, wherein: and the direct current power supply module (4) is connected with a second capacitor (3) in parallel.

9. An electroacoustic transducer bandwidth widening apparatus using a switch-type nonlinear non-foster system, characterized by comprising a processor configured to perform the steps of the electroacoustic transducer bandwidth widening method of any one of claims 1 to 8.