CN115149817A - Hybrid control LLC series resonant converter variable dead time control method - Google Patents

Abstract

The invention provides a variable dead zone control method of a hybrid control LLC series resonant converter, which respectively calculates the current of leading and lagging phase positions of bridge arm switching moments and the shortest dead zone time required by realizing ZVS (zero voltage switching on), simplifies the calculated amount by using a parameter fitting method, and calculates and controls the dead zone time of two bridge arm switching tubes on an embedded controller in real time. The invention considers the difference of dead time required by zero voltage switching-on of the switching tubes of two bridge arms at the primary side of the hybrid control LLC series resonance converter, performs real-time control, prevents the switching tubes at the primary side from losing zero voltage switching-on due to working condition change, reduces reverse conduction time of the switching tubes as much as possible and reduces loss. The invention has better application effect in the high-frequency and wide-gain-range hybrid control LLC series resonant converter.

Description

Hybrid control LLC series resonant converter variable dead time control method

technical field

The invention relates to the technical field of power electronics, in particular to a hybrid control LLC series resonant converter variable dead zone control method.

Background technique

LLC series resonant DC/DC converters have received extensive attention due to their simple circuit structure and superior switching performance. It is easy to realize the zero-voltage turn-on (ZVS) of the primary side switch tube and the zero-current turn-off (ZCS) of the secondary side rectifier tube, which is convenient to increase the switching frequency and improve the power density.

The LLC series resonant converter generally adopts a frequency conversion control strategy, which can stabilize the output voltage by changing the switching frequency to obtain the corresponding gain. However, when the input and output voltage range is wide, the frequency variation range is large, which leads to difficulties in the design of magnetic components and EMI design. Therefore, a hybrid control strategy of frequency conversion + phase shift is generally used to reduce the frequency variation range. However, when the phase shift angle gradually increases, the turn-off current of the phase-leading bridge arm gradually increases, and the phase-lag bridge arm turn-off current gradually decreases. In order to ensure that the primary side switch tube is ZVS (zero voltage turn-on), it is necessary to give the primary side bridge arm a suitable dead time, so that the bridge arm turn-off current can be charged and discharged for the parasitic output capacitance of the switch tube during the dead time.

The traditional method uses the primary side switch to achieve the maximum dead time required for ZVS, and does not distinguish between the two bridge arms. However, in the high-frequency hybrid control LLC series resonant converter, when the phase shift angle is large, the dead time required by the switch of the leading bridge arm is small, and the reverse conduction time of the body diode will be prolonged, which will cause a large reverse conduction time. pass loss. The prior art analyzes the variable dead time control of LLC resonant converters under variable frequency control. This method can dynamically adjust the dead time of the primary side switch tube according to the switching frequency and load conditions during circuit operation, but this method does not work. It is not suitable for LLC resonant converters under phase-shift and frequency-conversion hybrid control.

SUMMARY OF THE INVENTION

Therefore, it is necessary to propose a dead time control method for the hybrid control LLC series resonant converter, which can dynamically adjust the dead time of the switches of the two bridge arms according to the circuit operation, and prevent the switches from losing zero due to the short dead time. When the voltage is turned on, or the dead time is too long, the reverse conduction loss of the switch tube is large, so as to improve the overall operating efficiency of the circuit.

The purpose of the present invention is to disclose a hybrid control LLC series resonant converter variable dead zone control method, which improves the problem of reducing the circuit operation efficiency due to the unreasonable setting of the phase leading and phase lag bridge arm switch dead zones. The invention has the advantages of accurate calculation result, small calculation amount, high operation efficiency, convenient implementation and the like.

It calculates the current at the switching time of the phase leading and lagging bridge arms and the shortest dead time required for zero-voltage turn-on respectively, and uses the method of parameter fitting to simplify the calculation amount, and calculates and controls the two bridge arms in real time on the embedded controller. Dead time of the switch. The present invention takes into account the difference in dead time required for zero-voltage turn-on of the two bridge arm switches on the primary side of the hybrid control LLC series resonant converter, and performs real-time control to prevent the primary-side switch from losing zero-voltage turn-on due to changes in operating conditions , and reduce the reverse conduction time of the switch tube as much as possible, reducing the loss. The invention has good application effect in the mixed control LLC series resonant converter with high frequency and wide gain range.

The present invention specifically adopts the following technical solutions:

A hybrid control LLC series resonant converter variable dead zone control method is characterized in that, comprises the following steps:

Step S1: Obtain circuit hardware parameters and the range of input and output parameters, including: resonant inductance L _r , resonant capacitor _Cr , primary excitation inductance L _m of transformer T _x , transformer turns ratio n, primary switch output capacitor C _oss , Input voltage V _i variation range, output voltage V _o variation range, and output current I _o variation range;

Step S2: using the improved time-domain analysis method to calculate the current i _off at the turn-off moment of the bridge arm switch tube of which the phase of the drive signal of the LLC resonant converter is mixed and controlled under different input and output parameters;

The improved time-domain analysis method includes: writing out the equations satisfied by the resonant current i _Lr , the excitation current i _Lm and the resonant capacitor voltage v _Cr in different time periods within half a switching cycle, and combining the full-bridge LLC resonant transformation in steady state According to the symmetry of the resonant waveform of the resonator, the resonant current i _Lr , the excitation current i _Lm and the resonant capacitor voltage v _Cr at any time are solved;

Step S3: According to the parasitic capacitance C _eq of the switch tube and the turn-off current i _off , calculate and obtain the shortest dead time t _{dead_min} required for the primary side switch tube to realize zero-voltage turn-on under different input and output parameters;

C_oss为开关管寄生输出电容；Among them, the calculation method of the shortest dead time is as follows

C _oss is the parasitic output capacitance of the switch;

Step S4: use a simple fitting function model f(f _s , θ) to fit the functional relationship between the shortest dead time t _{dead_min} and the switching frequency f _s and the phase shift angle θ, and solve the undetermined parameters in the fitting function model;

Step S5: Write the solved fitting function model f(f _s , θ) into the embedded controller, and calculate it in real time when the circuit is running to obtain the requirements for zero-voltage turn-on of each switch on the primary side under this working condition Minimum dead time;

Step S6: After adding an appropriate margin to the shortest dead time calculated in step S5, the final dead time signal is transmitted to the PWM generator, and the dead time of the primary switch is controlled by the drive circuit.

Further, in the improved time domain analysis method, in the time period _tsp during which the excitation inductance _Lm , the resonant inductance _Lr and the resonant capacitor _Cr resonate together, the resonant current is regarded as a linear change, and the rate of the linear change is The average value of the rate of change of the resonant current at the _tsp endpoint.

Further, the improved time domain analysis method specifically includes:

Step S21: Obtain the specific software and hardware parameters of the controlled LLC series resonant converter during operation, including:

Resonant inductance L _r , excitation inductance L _m , resonant capacitor C _r , transformer ratio n, switching frequency f _s , phase shift angle θ, input voltage V _i , output voltage V _o , output current I _o ; among them, V _i , At least two of V _o and I _o should be obtained;

Step S22: Calculate the initial value conditions of the excitation current and the resonant capacitor voltage:

Among them, i _Lr refers to the instantaneous value of the resonant inductor current; v _Cr refers to the instantaneous value of the resonant capacitor voltage; t _{0 ~ 2} and t _{2 ~ 3} refer to the time from t ₀ to t ₂ and the time from t ₂ to t ₃ respectively, and t ₀ is The initial moment of a switching cycle, t ₂ is the discontinuous moment of the secondary side current in the first half of the switching cycle, which is unknown, and t ₃ is the middle moment of the switching cycle; k ₂₃ refers to the change of the resonant inductor current during the time period from t ₂ to t ₃ . rate;

Solve the equation set (1) to obtain the initial value of the excitation current i _Lr (t ₀ ) and the initial value of the resonance capacitor voltage v _Cr (t ₀ );

Step S23: According to the operation of the circuit, list the differential equations that i _Lr and v _Cr satisfy in the time period from t ₀ to t ₃ , and combine the initial value conditions in equation (1) to solve the time domain to obtain i _Lr and v _Cr expression, the corresponding differential equation is:

Step S24: According to the satisfied boundary conditions of i _Lr and u _Cr , bring it into the time domain expression obtained in formula (3), eliminate the unknowns therein, and obtain the final expression, and the corresponding boundary conditions are:

Step S25 : Use the numerical solution to obtain the off currents of the two bridge arms under different working conditions, including: the lead off current i _Lr (t ₁ ) and the lagging off current i _Lr (t ₃ ).

Further, the simple fitting function model used in step S4 is:

Among them, f _s refers to the switching frequency; θ refers to the phase shift angle, which is greater than or equal to 0, and C ₀ -C ₅ are the parameters to be fitted; the phase shift angle θ; the dead time under different load resistances is fitted, Simplifies dead time as a simple function of switching frequency and phase shift angle.

Further, the shortest dead time required for zero-voltage turn-on of the bridge arm switch with the phase lag of the drive signal calculated in step S5 is greater than or equal to the shortest dead time required for zero-voltage turn-on of the phase-leading bridge arm switch.

Further, a voltage and current sampling circuit is used to connect the load of the LLC series resonant converter to collect the information of the output voltage V _o and the output current I _o ; a driving circuit is used to connect the four switching tubes of the LLC series resonant converter to provide a driving signal; The embedded controller is connected with the voltage and current sampling circuit and the driving circuit.

Compared with the prior art, the solution provided by the present invention and its preferred solution can dynamically adjust the dead time of the switch tubes of different bridge arms of the primary side according to the change of input and output without increasing the hardware circuit resources at all, so as to make it work. In the state where ZVS can be achieved, it can prevent some switches from losing ZVS conditions due to changes in working conditions, and greatly reduce the reverse conduction loss of the primary switch when working at high frequency and large phase angle. It has the advantages of less computation, easy implementation, and high circuit operation efficiency.

Description of drawings

The present invention will be described in further detail below in conjunction with the accompanying drawings and specific embodiments:

FIG. 1 is a topology diagram of a full-bridge LLC series resonant converter according to an embodiment of the present invention.

FIG. 2 is a time-domain waveform diagram of a phase-shifting and frequency-changing hybrid control LLC series resonant converter according to an embodiment of the present invention.

3 is an equivalent circuit diagram of a resonant cavity when the full-bridge LLC series resonant converter according to an embodiment of the present invention operates in different time periods.

FIG. 4 is an equivalent circuit diagram of a full-bridge LLC series resonant converter in accordance with an embodiment of the present invention when the phase leads the switch tube on the bridge arm when the switch is turned off.

FIG. 5 is an equivalent circuit diagram of a full-bridge LLC series resonant converter at the turn-off time of the lower switch of the bridge arm with a phase lag according to an embodiment of the present invention.

FIG. 6 is a schematic diagram of a method for controlling variable dead zone of a hybrid control LLC series resonant converter according to an embodiment of the present invention.

FIG. 7 is a PSIM simulation waveform comparison diagram of the existing dead zone control method according to the embodiment of the present invention and the variable dead zone control method of the present invention.

Detailed ways

In order to make the features and advantages of the present patent more obvious and easy to understand, the following examples are given and described in detail as follows.

It should be noted that the following detailed description is exemplary and intended to provide further explanation of the application. Unless otherwise defined, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this application belongs.

It should be noted that the terminology used herein is for the purpose of describing specific embodiments only, and is not intended to limit the exemplary embodiments according to the present application. As used herein, unless the context clearly dictates otherwise, the singular is intended to include the plural as well, furthermore, it is to be understood that when the terms "comprising" and/or "including" are used in this specification, it indicates that There are features, steps, operations, devices, components and/or combinations thereof.

In the scheme provided by the present invention:

In the first aspect, an improved time domain analysis method is provided, which is applied to a phase-shift and frequency-modulated hybrid control full-bridge LLC series resonant converter to obtain the turn-off currents of the phase leading and lagging bridge arms. The full-bridge LLC series resonant converter sequentially includes an input capacitor, an inverter bridge, a resonant cavity, and a rectifier filter circuit. The inverter bridge includes two half-bridge arms composed of four switch tubes. If the phase difference θ between the driving signals of the two bridge arms is greater than zero, the bridge arm with the driving signal phase leading is the leading bridge arm, otherwise it is the lagging bridge. arm; resonant cavity includes resonant inductor, transformer, resonant capacitor, and is composed of cascade; rectifier filter circuit includes full-wave, full-bridge or voltage doubler rectifier circuit and capacitor filter circuit, which is used to convert the AC power output by the transformer into DC power .

The specific implementation of this method is as follows:

Step S21: Obtain the specific software and hardware parameters of the controlled LLC series resonant converter during operation, including:

Resonant inductance L _r , excitation inductance L _m , resonant capacitor C _r , transformer ratio n, switching frequency f _s , phase shift angle θ, input voltage V _i , output voltage V _o , output current I _o . Among them, at least two of _Vi , V _o and I _o should be obtained.

Step S22: Calculate the initial value conditions of the excitation current and the resonant capacitor voltage:

Among them, i _Lr refers to the instantaneous value of the resonant inductor current; v _Cr refers to the instantaneous value of the resonant capacitor voltage; t _{0 ~ 2} and t _{2 ~ 3} refer to the time from t ₀ to t ₂ and the time from t ₂ to t ₃ respectively, and t ₀ is The initial moment of a switching cycle, t ₂ is the discontinuous moment of the secondary side current in the first half of the switching cycle, which is unknown, and t ₃ is the middle moment of the switching cycle; k ₂₃ refers to the change of the resonant inductor current during the time period from t ₂ to t ₃ . rate.

Solving equations (1) can obtain the initial value of the excitation current i _Lr (t ₀ ) and the initial value of the resonant capacitor voltage v _Cr (t ₀ ).

Step S23: According to the operation of the circuit, list the differential equations that i _Lr and v _Cr satisfy in the time period from t ₀ to t ₃ , and combine the initial value conditions in equation (1) to solve the time domain to obtain i _Lr and v _Cr expression. The corresponding differential equation is:

Note that the time domain expression obtained at this time includes the unknown t ₂ , and one of V _i , V _o , and I _o .

Step S24: According to the satisfied boundary conditions of i _Lr and u _Cr , bring it into the time domain expression obtained in formula (3), eliminate the unknowns therein, and obtain the final expression. The corresponding boundary conditions are:

Step S25 : Use the numerical solution to obtain the off currents of the two bridge arms under different working conditions, including: the lead off current i _Lr (t ₁ ) and the lagging off current i _Lr (t ₃ ).

Through the improved time domain analysis method in the first aspect, parameters such as the DC gain of the LLC series resonant converter and the resonant current at any time can be accurately calculated. Compared with the traditional method, the computational complexity is slightly increased and the calculation accuracy is greatly increased, ensuring the accuracy of the dead time control.

In the second aspect, a dead time calculation and numerical fitting method are provided, which are used to accurately calculate the dead time required for the two bridge arms of the LLC series resonant converter to achieve ZVS, and use a simpler fitting function to fit Obtain the functional relationship between the required dead time and the circuit software and hardware parameters.

The specific implementation of this method is as follows:

Step A1: Obtain the distribution parameters of the primary side switch tubes of the LLC series resonant converter controlled, including the output capacitance C _oss of the primary side switch tubes.

Step A2: Combining with the improved time domain analysis method proposed in the first aspect, solve and obtain the minimum dead time required by the switching tube.

Among them, i _off refers to the off current of the bridge arm, the off current of the leading bridge arm is i _Lr (t ₁ ), and the off current of the lagging bridge arm is i _Lr (t ₃ ).

Step A3: Use mathematical software (such as mathcad, etc.) to fit the dead time under different load resistances, and simplify the dead time to a simple function of switching frequency and phase shift angle. The fitting function is:

Among them, f _s refers to the switching frequency, θ refers to the phase shift angle, and C ₀ -C ₅ are the parameters to be fitted.

Through the dead-time calculation and fitting method of the second aspect, the dead-time required by the switch tube to realize ZVS in the hybrid control LLC series resonant converter can be accurately calculated. In addition, after using the function to perform parameter fitting, the complexity of the dead time calculation is greatly reduced, and it is convenient to use the method in the embedded controller.

In a third aspect, a hybrid control LLC series resonant converter bridge arm variable dead time control method is provided, which is used to precisely control the dead time of the two bridge arms and adapt to changes in circuit operating conditions. include:

Step B1: Sample the output voltage and output current of the circuit, and adjust the output voltage to stabilize through closed-loop control.

Step B2: According to the control parameters (switching frequency, phase shift angle) stored in the controller, the above-mentioned fitting function is brought into, and the optimal dead time of the leading and lagging bridge arms is calculated respectively.

Step B3: On the basis of the optimal dead time of the two bridge arms, after adding a small amount of margin respectively, output the corresponding driving signal to the driving circuit and control the switching of the switch tube.

The scheme of the present invention and the corresponding principle are further introduced below in conjunction with the accompanying drawings:

Figure 1 shows the topology of the full-bridge LLC series resonant converter, where V _i is the input voltage; Q1-Q4 are switching tubes, forming a full-bridge inverter circuit; C _q1 -C _q4 are the parasitic output capacitances of the switching tubes; L _r is the resonant inductance , C _r is the resonant capacitor, T _x is the transformer with center tap, the transformer excitation inductance is L _m , D1 and D2 are rectifier diodes, forming a full-wave rectifier circuit, C _o is the output filter capacitor, and _Ro is the load resistance. In this embodiment, GaN HEMT is preferably used as the switch tube, which has no parasitic body diode in its body, but it can still conduct reverse conduction due to the symmetry of the structure, and there is no reverse recovery process of ordinary diodes. The disadvantage is that it conducts reversely. The pressure drop is large, and the reverse conduction for a long time will cause a large loss.

FIG. 2 is a time-domain waveform diagram of the above-mentioned full-bridge LLC resonant converter operating in a phase-shift and frequency-modulation hybrid control mode. In this embodiment, the current of the secondary rectifier tube is intermittent. FIG. 3 is an equivalent circuit diagram of a resonant cavity at different time periods. In the following, the analysis method is described by time-domain waveform diagram and equivalent circuit diagram. For the convenience of analysis, time t ₀ indicates t=0.

[t ₀ -t ₁ ]: The switches Q1 and Q4 are turned on, the input voltage of the resonant cavity is the power supply voltage, the secondary diode is turned on, the excitation inductance is clamped by the output voltage, and only the resonant inductor and resonant capacitor participate in the resonance. The equivalent circuit diagram of this period is shown in Figure 3(a). The resonant current, the excitation current and the resonant capacitor voltage satisfy the equations as follows:

[t ₁ -t ₂ ]: Switch tube Q1 is turned off, and Q3 and Q4 are turned on. The input of the resonant cavity is short-circuited, the secondary side rectifier diode continues to flow freely, and the resonant current drops rapidly. The equivalent circuit diagram of this period is shown in Figure 3(b). The resonant current, the excitation current and the resonant capacitor voltage satisfy the equations respectively:

[t ₂ -t ₃ ]: At time t ₂ , the secondary side rectifier diode current drops to zero, and the diode is naturally turned off. The magnetizing inductance is no longer clamped by the output voltage and participates in resonance. The equivalent circuit diagram of this period is shown in Figure 3(c). The traditional analysis method considers that the excitation inductance is much larger than the resonant inductance, so the excitation current during this period is regarded as a constant value. However, when the phase shift angle is large or L _m /L _r is small, it will cause a large analysis error. There are also methods to perform calculations directly using differential equations, but the overall calculation is computationally expensive and unintuitive. The present invention improves the traditional analysis method, linearizes the excitation current in this period, and takes the average value of the resonant current change rate at _t2 and _t3 as the resonant current change rate in the whole period. In the period of t ₂ -t ₃ , according to the equivalent circuit shown in Figure 3(c), the resonant current and the resonant capacitor voltage satisfy the relationship:

Therefore, the rate of change of the resonant current k at any time is:

Then the average value k ₂₃ of the rate of change of the resonant current at times t ₂ and t ₃ is:

Assuming that during the period of t ₂ -t ₃ , the resonant current changes linearly at the rate of k ₂₃ , then the equation during this period is simplified as:

In addition, when the system is stable, the resonant waveforms of adjacent half switching cycles have the same shape and are symmetrical about the time axis, so there are:

In the above equations, L _m ; L _r ; C _r are circuit hardware parameters, which have been determined during design and are known parameters; t ₀ and t ₂ can be obtained by known parameters switching frequency f _s and phase shift angle θ .

Therefore, the numerical solution of the DC gain M=nV _o /V _i of the hybrid control LLC series resonant converter can be obtained through mathematical software. If one of V _o and V _i is set, the above equations can be further solved to obtain the values of resonant current, excitation current, and resonant capacitor voltage at any time.

Further, according to the waveform diagram shown in FIG. 2 , the resonant currents at t ₁ and t ₃ are the turn-off currents of the leading bridge arm and the lagging bridge arm. At t ₁ and t ₃ , the current flow in the main circuit is shown in Figure 4 and Figure 5, respectively. After the switch is turned off, the resonant current will charge and discharge the output capacitors of the two switches of the bridge arm. In order to ensure that the primary side full bridge of the LLC resonant converter can reliably realize zero-voltage turn-on, it is necessary to ensure that the output capacitor of the switch tube is fully charged and discharged within the dead time. Therefore, the shortest dead time of the leading and lagging bridge arms is:

Among them, C _oss is the output capacitance of the primary switch tube; i _off is the switch-off current, and the i _off of the leading and lagging bridge arms are the resonant currents i _Lr ( _{t 1 )} and i _Lr ( _t3 ).

Since the turn-off current i _Lr (t ₁ ) of the phase-leading bridge arm switches Q1 and Q3 is greater than the phase-lag bridge arm switch Q2 and Q4 turn-off current i _Lr (t ₃ ), the dead time required for the advance bridge arm switch It is shorter, and the dead time required for the switching of the lagging bridge arm is larger. Giving driving signals of different dead zones to the switch tubes of the two bridge arms can reduce the reverse conduction loss of the switch tubes, thereby reducing the loss.

The above method can more accurately calculate the dead time required by the switching tube, but the overall calculation amount is large. In practical applications, although it can be calculated again after many switching cycles, the general embedded controller is still unaffordable. In order to facilitate the use of the above variable dead zone control method in the embedded controller, the method through parameter fitting is further introduced below to reduce the amount of calculation.

When the switching frequency f _s , the phase shift angle θ and the load resistance _Ro remain unchanged, the DC gain M of the circuit remains unchanged, the shape of the resonator current waveform has nothing to do with the input and output voltages, and the current size is proportional to the input and output voltages. According to equation (13), the minimum dead time required by the switch is also proportional to the input voltage. Therefore, the minimum dead time required by the switch tube has nothing to do with the input and output voltage, and is only determined by three variables, the switching frequency f _s , the phase shift angle θ and the load resistance R _o .

When the output resistance does not change, the minimum dead time required by the switch is a function of the switching frequency f _s and the phase shift angle θ. Through the above improved time domain analysis method, the minimum dead time required by the switch tube under different switching frequency f _s and phase shift angle θ is calculated. Further, use the following fitting function to fit the function:

Among them, C ₀ -C ₅ are parameters to be fitted.

Further, for the change of the load resistance, a plurality of possible load resistance values can be taken for calculation to obtain different fitting functions. When the circuit is running, first detect the actual load resistance value, and use the linear interpolation method to estimate the minimum dead time required for the switch.

Based on the improved time-domain analysis method and the optimal dead-time calculation and parameter fitting method of the bridge arm, the variable dead-time control method of the bridge arm of the hybrid control LLC series resonant converter can be obtained. The principle is shown in Figure 6, including:

Step 1: According to the hardware parameters and indicators, calculate the optimal dead time under different working conditions offline.

Step 2: Use the above fitting function to fit the dead time function, and write the calculated fitting function into the embedded controller.

Step 3: During the initial operation of the circuit, the output voltage and output current are collected in real time, and the voltage and current closed-loop control is normally performed. At this time, the dead time of the circuit is relatively large to prevent the primary side switch from losing zero voltage turn-on.

Step 4: After the circuit is basically running stably, parameters such as switching frequency and phase shift angle are obtained from the closed-loop control module, and combined with parameters such as output voltage and output current, they are brought into the fitting function to calculate the leading bridge arm and the lagging bridge respectively. Dead time required for the arm.

Step 5: Increase the dead time data calculated in step 4 with an appropriate margin and transmit it to the PWM generator to output a drive signal with the target dead time, thereby reducing the reverse conduction loss of the switch.

Step 6: When the dead time calculator detects that the circuit output voltage, output current, switching frequency, phase shift angle and other parameters have changed greatly. Repeat steps 4 and 5 again to ensure that the primary side switch tube can just achieve zero-voltage turn-on.

Therefore, the dead-time calculator only needs to be recalculated when a large change in circuit operating conditions is detected, which further reduces the computational load of the embedded controller.

The waveform diagrams of the PSIM simulation circuits applying the existing dead zone control method and the new variable dead zone control method of the present invention are respectively shown in (a) and (b) of FIG. 7 . When the converter operates at a switching frequency of 700 kHz and a phase shift angle of 100°, in order to achieve zero-voltage turn-on of the two bridge arm switches, the dead time of the phase lead and lag bridge arm switches in the existing method is consistent, which is both. 20ns. Under the same conditions, after using the new variable dead zone control method of the present invention, the dead zone time of the leading bridge arm switch tube is reduced to about 3 ns, the dead zone time of the lagging bridge arm switch tube is reduced to about 17 ns, and the two bridge arm switches are both. Just to achieve zero voltage turn-on. The method ensures that the switch tubes can be turned on at zero voltage under different circuit operation conditions, greatly reduces the reverse conduction time of the switch tubes, and reduces the loss in the switching process.

The above are only preferred embodiments of the present invention, and are not intended to limit the present invention in other forms. Any person skilled in the art may use the technical content disclosed above to make changes or modifications to equivalent changes. Example. However, any simple modifications, equivalent changes and modifications made to the above embodiments according to the technical essence of the present invention without departing from the content of the technical solutions of the present invention still belong to the protection scope of the technical solutions of the present invention.

This patent is not limited to the above-mentioned best embodiment, anyone can come up with other various forms of hybrid control LLC series resonant converter variable dead zone control methods under the inspiration of this patent. The equivalent changes and modifications of the above shall fall within the scope of this patent.

Claims (6)

1. a hybrid control LLC series resonant converter becomes dead zone control method, is characterized in that, comprises the following steps: Step S1: Obtain circuit hardware parameters and the range of input and output parameters, including: resonant inductance L _r , resonant capacitor _Cr , primary excitation inductance L _m of transformer T _x , transformer turns ratio n, primary switch output capacitor C _oss , Input voltage V _i variation range, output voltage V _o variation range, and output current I _o variation range; Step S2: using the improved time-domain analysis method to calculate the current i _off at the turn-off moment of the bridge arm switch tube of which the phase of the drive signal of the LLC resonant converter is mixed and controlled under different input and output parameters; The improved time-domain analysis method includes: writing out the equations satisfied by the resonant current i _Lr , the excitation current i _Lm and the resonant capacitor voltage v _Cr in different time periods within half a switching cycle, and combining the full-bridge LLC resonant transformation in steady state According to the symmetry of the resonant waveform of the resonator, the resonant current i _Lr , the excitation current i _Lm and the resonant capacitor voltage v _Cr at any time are solved; Step S3: According to the parasitic capacitance C _eq of the switch tube and the turn-off current i _off , calculate and obtain the shortest dead time t _{dead_min} required for the primary side switch tube to realize zero-voltage turn-on under different input and output parameters; Among them, the calculation method of the shortest dead time is as follows

C _oss is the parasitic output capacitance of the switch; Step S4: use a simple fitting function model f(f _s , θ) to fit the functional relationship between the shortest dead time t _{dead_min} and the switching frequency f _s and the phase shift angle θ, and solve the undetermined parameters in the fitting function model; Step S5: Write the solved fitting function model f(f _s , θ) into the embedded controller, and calculate it in real time when the circuit is running to obtain the requirements for zero-voltage turn-on of each switch on the primary side under this working condition Minimum dead time; Step S6: After adding an appropriate margin to the shortest dead time calculated in step S5, the final dead time signal is transmitted to the PWM generator, and the dead time of the primary switch is controlled by the drive circuit. 2. The hybrid control LLC series resonant converter variable dead zone control method according to claim 1, characterized in that: in the improved time domain analysis method, in the excitation inductance L _m , the resonant inductance L _r and the resonant capacitor C _r During the common resonance time period _tsp , the resonant current is regarded as a linear change, and the rate of linear change is the average value of the rate of change of the resonant current at the end point of _tsp . 3. The hybrid control LLC series resonant converter variable dead zone control method according to claim 1 or 2, wherein the improved time domain analysis method specifically comprises: Step S21: Obtain the specific software and hardware parameters of the controlled LLC series resonant converter during operation, including: Resonant inductance L _r , excitation inductance L _m , resonant capacitor C _r , transformer ratio n, switching frequency f _s , phase shift angle θ, input voltage V _i , output voltage V _o , output current I _o ; among them, V _i , At least two of V _o and I _o should be obtained; Step S22: Calculate the initial value conditions of the excitation current and the resonant capacitor voltage:

Among them, i _Lr refers to the instantaneous value of the resonant inductor current; v _Cr refers to the instantaneous value of the resonant capacitor voltage; t _{0 ~ 2} and t _{2 ~ 3} refer to the time from t ₀ to t ₂ and the time from t ₂ to t ₃ respectively, and t ₀ is The initial moment of a switching cycle, t ₂ is the discontinuous moment of the secondary side current in the first half of the switching cycle, which is unknown, and t ₃ is the middle moment of the switching cycle; k ₂₃ refers to the change of the resonant inductor current during the time period from t ₂ to t ₃ . rate; Solve the equation set (1) to obtain the initial value of the excitation current i _Lr (t ₀ ) and the initial value of the resonance capacitor voltage v _Cr (t ₀ ); Step S23: According to the operation of the circuit, list the differential equations that i _Lr and v _Cr satisfy in the time period from t ₀ to t ₃ , and combine the initial value conditions in equation (1) to solve the time domain to obtain i _Lr and v _Cr expression, the corresponding differential equation is:

Step S24: According to the satisfied boundary conditions of i _Lr and u _Cr , bring it into the time domain expression obtained in formula (3), eliminate the unknowns therein, and obtain the final expression, and the corresponding boundary conditions are:

Step S25 : Use the numerical solution to obtain the off currents of the two bridge arms under different working conditions, including: the lead off current i _Lr (t ₁ ) and the lagging off current i _Lr (t ₃ ). 4. hybrid control LLC series resonant converter variable dead zone control method according to claim 1 is characterized in that: the simple fitting function model used in step S4 is:

Among them, f _s refers to the switching frequency; θ refers to the phase shift angle, which is greater than or equal to 0, and C ₀ -C ₅ are the parameters to be fitted; the phase shift angle θ; the dead time under different load resistances is fitted, Simplifies dead time as a simple function of switching frequency and phase shift angle. 5. hybrid control LLC series resonant converter variable dead zone control method according to claim 1, is characterized in that: in step S5, the bridge arm switch tube of the phase lag of the driving signal obtained by calculation realizes the shortest dead time required for zero voltage turn-on. The zone time is greater than or equal to the shortest dead zone time required for the phase-leading bridge arm switch to achieve zero-voltage turn-on. 6. hybrid control LLC series resonant converter variable dead zone control method according to claim 1, is characterized in that: adopt voltage current sampling circuit to connect the load of LLC series resonant converter to collect output voltage V _o and output current I _o The four switching tubes of the LLC series resonant converter are connected by a driving circuit to provide driving signals; the embedded controller is connected with the voltage and current sampling circuit and the driving circuit.

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