CN115149817A - Variable dead zone control method for hybrid control LLC series resonant converter - Google Patents

Variable dead zone control method for hybrid control LLC series resonant converter Download PDF

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CN115149817A
CN115149817A CN202210875991.2A CN202210875991A CN115149817A CN 115149817 A CN115149817 A CN 115149817A CN 202210875991 A CN202210875991 A CN 202210875991A CN 115149817 A CN115149817 A CN 115149817A
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current
time
switching
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resonant
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毛行奎
张彬意
郑嘉冬
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Xiamen Lijing New Energy Technology Co ltd
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Fuzhou University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33573—Full-bridge at primary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/083—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the ignition at the zero crossing of the voltage or the current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/01—Resonant DC/DC converters
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

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  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)

Abstract

The invention provides a variable dead zone control method of a hybrid control LLC series resonant converter, which respectively calculates the current of leading and lagging phase positions of bridge arm switching moments and the shortest dead zone time required by realizing ZVS (zero voltage switching on), simplifies the calculated amount by using a parameter fitting method, and calculates and controls the dead zone time of two bridge arm switching tubes on an embedded controller in real time. The invention considers the difference of dead time required by zero voltage switching-on of the switching tubes of two bridge arms at the primary side of the hybrid control LLC series resonance converter, performs real-time control, prevents the switching tubes at the primary side from losing zero voltage switching-on due to working condition change, reduces reverse conduction time of the switching tubes as much as possible and reduces loss. The invention has better application effect in the high-frequency and wide-gain-range hybrid control LLC series resonant converter.

Description

Variable dead zone control method for hybrid control LLC series resonant converter
Technical Field
The invention relates to the technical field of power electronics, in particular to a variable dead zone control method of a hybrid control LLC series resonant converter.
Background
LLC series resonant DC/DC converters are receiving much attention due to their advantages such as simple circuit structure and superior switching performance. The zero-voltage switching-on (ZVS) of the primary side switching tube and the zero-current switching-off (ZCS) of the secondary side rectifying tube are easy to realize, and the switching frequency and the power density are convenient to improve.
The LLC series resonant converter generally adopts a frequency conversion control strategy, and the switching frequency is changed to obtain a corresponding gain to stabilize the output voltage. However, when the input/output voltage range is wide, the frequency variation range is large, which leads to the problems of difficult design of magnetic elements and EMI design, and the like, so a hybrid control strategy of frequency conversion and phase shift is generally adopted to reduce the frequency variation range. However, when the phase shift angle gradually increases, the phase leading bridge arm turn-off current gradually increases, and the phase lagging bridge arm turn-off current gradually decreases. In order to ensure that the primary side switching tube realizes ZVS (zero voltage switching on), the primary side bridge arm needs to be given proper dead time, so that the bridge arm turn-off current can be completed for charging and discharging of the parasitic output capacitor of the switching tube in the dead time.
The traditional method fixedly uses a primary side switching tube to realize the maximum dead time required by ZVS, and two bridge arms are not distinguished. However, in the high-frequency hybrid control LLC series resonant converter, the dead time required by the leading bridge arm switching tube is small when the phase shift angle is large, and the reverse conduction time of the body diode thereof will be prolonged, which will cause large reverse conduction loss. In the prior art, dead zone control of an LLC resonant converter under variable frequency control is analyzed, the dead zone time of a primary side switching tube can be dynamically adjusted according to switching frequency, load condition and the like during circuit operation, but the method is not suitable for the LLC resonant converter under phase-shifting variable frequency hybrid control.
Disclosure of Invention
Therefore, there is a need to provide a dead time control method for a hybrid control LLC series resonant converter, which can dynamically adjust the dead time of the switching tubes of the two bridge arms according to the circuit operation condition, and prevent the switching tubes from losing a zero-voltage turn-on state due to a short dead time or from having a large reverse conduction loss due to an excessively long dead time, so as to improve the overall operation efficiency of the circuit.
The invention aims to disclose a variable dead zone control method of a hybrid control LLC series resonant converter, which solves the problem that the dead zone setting of a phase leading bridge arm switching tube and a phase lagging bridge arm switching tube is unreasonable, so that the operation efficiency of a circuit is reduced. The invention has the advantages of accurate calculation result, small calculation amount, high operation efficiency, convenient implementation and the like.
The method respectively calculates the current of leading and lagging phases of the switching moments of the bridge arms and the shortest dead time required by realizing zero voltage switching-on, simplifies the calculated amount by using a parameter fitting method, and calculates and controls the dead time of the two bridge arm switching tubes on an embedded controller in real time. The invention considers the difference of dead time required by zero voltage switching-on of the switching tubes of two bridge arms at the primary side of the hybrid control LLC series resonance converter, performs real-time control, prevents the switching tubes at the primary side from losing zero voltage switching-on due to working condition change, reduces reverse conduction time of the switching tubes as much as possible and reduces loss. The invention has better application effect in the high-frequency and wide-gain-range hybrid control LLC series resonant converter.
The invention specifically adopts the following technical scheme:
a variable dead zone control method for a hybrid control LLC series resonant converter is characterized by comprising the following steps:
step S1: acquiring circuit hardware parameters and input and output parameter ranges, including: resonant inductor L r Resonant capacitor C r Transformer T x Primary side excitation inductance L m Transformer turn ratio n, primary side switch tube output capacitance C oss Input voltage V i Variation range, output voltage V o Variation range, and output current I o A range of variation;
step S2: under the condition of respectively calculating different input and output parameters by using an improved time domain analysis method, the current i at the turn-off time of a bridge arm switch tube for hybrid control of the phase lead and lag of a drive signal of the LLC resonant converter off ；
The improved time domain analysis method comprises the following steps: in which the resonant current i is written in half a switching cycle at different time intervals Lr Exciting current i Lm And resonant capacitor voltage v Cr The satisfied equation is combined with the symmetry of the resonance waveform of the full-bridge LLC resonance converter in a steady state to solve the resonance current i at any moment Lr Exciting current i Lm And resonant capacitor voltage v Cr ；
And step S3: according to the parasitic capacitance C of the switch tube eq And turn-off current i off Calculating the shortest dead time t required by the primary side switching tube to realize zero voltage switching-on under the condition of obtaining different input and output parameters dead_min ；
Wherein, the shortest dead time is calculated by
Figure BDA0003762114320000021
C oss A parasitic output capacitor of the switching tube;
and step S4: using a simple fitting function model f (f) s θ) fitting the shortest dead time t dead_min And the switching frequency f s Solving to-be-determined parameters in the fitting function model according to the function relation of the phase shift angle theta;
step S5: solving a good fitting function model f (f) s Theta) writing the data into the embedded controller, and calculating in real time when the circuit runs to obtain the shortest dead time required by each switching tube on the primary side to realize zero voltage switching-on under the working condition;
step S6: and (5) adding a proper margin to the shortest dead time calculated in the step (S5), transmitting a final dead time signal to a PWM generator, and controlling the dead time of the primary side switching tube through a driving circuit.
Further, in the improved time domain analysis method, in the excitation inductance L m Resonant inductance L r And a resonance capacitor C r Time period t of common resonance sp The resonant current is regarded as a linear change, and the rate of the linear change is t sp Average of the rate of change of the resonant current at the endpoint time.
Further, the improved time domain analysis method specifically includes:
step S21: obtaining specific software and hardware parameters of the controlled LLC series resonant converter during operation, including:
resonant inductor L r Excitation inductance L m Resonant capacitor C r Transformer transformation ratio n, switching frequency f s Phase shift angle θ, input voltage V i Output voltage V o Output current I o (ii) a Wherein, V i 、V o 、I o At least two of which should be obtained;
step S22: calculating the initial value conditions of the exciting current and the resonant capacitor voltage:
Figure BDA0003762114320000031
wherein i Lr A resonant inductor current transient; v. of Cr The instantaneous value of the voltage of the resonance capacitor; t is t 0～2 And t 2～3 Respectively mean t 0 To t 2 Time sum t 2 To t 3 Time of (t) 0 Is an initial moment of a switching cycle, t 2 The interruption time of the secondary side current in the first half of the switching period is unknown number t 3 Is the middle time of the switching cycle; k is a radical of 23 Finger resonant inductor current at t 2 To t 3 A rate of change over a period of time;
solving the system of equations (1) to obtain an initial value i of the exciting current Lr (t 0 ) And initial value v of resonant capacitor voltage Cr (t 0 )；
Step S23: according to the circuit operation condition, list i Lr And v Cr At t 0 To t 3 Solving to obtain i by combining the differential equation satisfied in the time period and the initial value condition in the formula (1) Lr And v Cr The corresponding differential equation is:
Figure BDA0003762114320000032
step S24: according to i Lr And u Cr Substituting the satisfied boundary conditions into the time domain expression obtained in the formula (3), eliminating the unknown number therein to obtain the final expression, phaseThe boundary conditions should be:
Figure BDA0003762114320000033
step S25: solving the turn-off current of the two bridge arms under different working conditions by using a numerical solution method, comprising the following steps of: leading bridge arm turn-off current i Lr (t 1 ) Delayed bridge arm off current i Lr (t 3 )。
Further, the simple fitting function model used in step S4 is:
Figure BDA0003762114320000041
wherein f is s The switching frequency; theta is the phase shift angle of 0,C or greater 0 -C 5 Is a parameter to be fitted; the phase shift angle theta; and fitting the dead time under different load resistances, and simplifying the dead time into a simple function of the switching frequency and the phase shift angle.
Further, the shortest dead time required by the bridge arm switch tube with the lagging driving signal phase obtained in the step S5 to achieve zero voltage switching-on is greater than or equal to the shortest dead time required by the bridge arm switch tube with the leading phase to achieve zero voltage switching-on.
Further, a voltage and current sampling circuit is adopted to be connected with a load of the LLC series resonant converter to acquire an output voltage V o And an output current I o The information of (a); a driving circuit is adopted to be connected with four switching tubes of the LLC series resonant converter so as to provide driving signals; the embedded controller is connected with the voltage and current sampling circuit and the driving circuit.
Compared with the prior art, the scheme provided by the invention and the preferred scheme thereof can dynamically adjust the dead time of the switching tubes of different bridge arms on the primary side according to the change of input and output under the condition of not increasing hardware circuit resources completely, so that the switching tubes can work in a state of just realizing ZVS, the condition that part of the switching tubes lose ZVS due to the change of working conditions is prevented, and the reverse conduction loss of the switching tubes on the primary side is greatly reduced when the switching tubes work at high frequency and large phase shift angle. The method has the advantages of small operation amount, convenient implementation, high circuit operation efficiency and the like.
Drawings
The invention is described in further detail below with reference to the following figures and detailed description:
fig. 1 is a topological diagram of a full-bridge LLC series resonant converter in accordance with an embodiment of the present invention.
FIG. 2 is a time domain waveform diagram of an LLC series resonant converter with phase-shift frequency conversion hybrid control according to an embodiment of the invention.
Fig. 3 is a resonant cavity equivalent circuit diagram of a full-bridge LLC series resonant converter operating in different periods according to an embodiment of the present invention.
Fig. 4 is an equivalent circuit diagram of the switching tube on the phase lead bridge arm of the full-bridge LLC series-resonant converter at the turn-off time according to the embodiment of the present invention.
Fig. 5 is an equivalent circuit diagram of the full-bridge LLC series resonant converter according to the embodiment of the present invention at the turn-off time of the switching tube under the phase lag bridge arm.
Fig. 6 is a schematic diagram of a dead zone varying control method for a hybrid control LLC series resonant converter according to an embodiment of the present invention.
Fig. 7 is a comparison diagram of simulation waveforms of a PSIM in the dead zone control method according to the present invention and the variable dead zone control method according to the present invention.
Detailed Description
In order to make the features and advantages of the present disclosure comprehensible, embodiments accompanied with figures are described in detail below.
It should be noted that the following detailed description is exemplary and is intended to provide further explanation of the disclosure. Unless defined otherwise, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this application belongs.
It is noted that the terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting of exemplary embodiments according to the present application. As used herein, the singular forms "a", "an", and "the" are intended to include the plural forms as well, and it should be understood that when the terms "comprises" and/or "comprising" are used in this specification, they specify the presence of stated features, steps, operations, devices, components, and/or combinations thereof, unless the context clearly indicates otherwise.
In the scheme provided by the invention:
on the first hand, an improved time domain analysis method is provided, which is applied to a phase-shifting and frequency-modulating hybrid control full-bridge LLC series resonant converter to obtain the turn-off current of a phase leading bridge arm and a phase lagging bridge arm. The full-bridge LLC series resonant converter sequentially comprises an input capacitor, an inverter bridge, a resonant cavity and a rectifying and filtering circuit. The inverter bridge comprises two half-bridge arms consisting of four switching tubes, and if the phase difference theta between the driving signals of the two bridge arms is greater than zero, the bridge arm with the leading driving signal phase is a leading bridge arm, otherwise, the bridge arm with the lagging driving signal phase is a lagging bridge arm; the resonant cavity comprises a resonant inductor, a transformer and a resonant capacitor, and is formed by cascading the resonant inductor, the transformer and the resonant capacitor; the rectification filter circuit comprises a full-wave rectification circuit, a full-bridge rectification circuit or a voltage-multiplying rectification circuit and a capacitor filter circuit and is used for converting alternating current output by the transformer into direct current.
The method is concretely realized as follows:
step S21: obtaining specific software and hardware parameters of the controlled LLC series resonant converter during operation, including:
resonant inductor L r Excitation inductance L m Resonant capacitor C r Transformer transformation ratio n, switching frequency f s Angle of phase shift θ, input voltage V i Output voltage V o Output current I o . Wherein, V i 、V o 、I o At least two of which should be obtained.
Step S22: calculating the initial value conditions of the exciting current and the resonant capacitor voltage:
Figure BDA0003762114320000051
wherein i Lr A resonant inductor current transient; v. of Cr A voltage instantaneous value of a finger resonance capacitor; t is t 0～2 And t 2～3 Respectively mean t 0 To t 2 Time sum t 2 To t 3 Time of (t) 0 Is an initial moment of a switching cycle, t 2 The interruption time of the secondary current in the first half of the switching period is unknown, t 3 Is the middle time of the switching cycle; k is a radical of 23 Finger resonant inductor current at t 2 To t 3 The rate of change over a period of time.
Solving the equation set (1) can obtain the initial value i of the exciting current Lr (t 0 ) And initial value v of resonant capacitor voltage Cr (t 0 )。
Step S23: according to the circuit operation condition, list i Lr And v Cr At t 0 To t 3 Solving to obtain i by combining the differential equation satisfied in the time period and the initial value condition in the formula (1) Lr And v Cr The time domain expression of (1). The corresponding differential equation is:
Figure BDA0003762114320000061
note that the time domain expression found at this time includes an unknown number t 2 And V i 、V o 、I o One of them.
Step S24: according to i Lr And u Cr Substituting the satisfied boundary condition into the time domain expression obtained in the formula (3), and eliminating the unknown number therein to obtain a final expression. The corresponding boundary conditions are:
Figure BDA0003762114320000062
step S25: the method for solving the turn-off current of the two bridge arms under different working conditions by using a numerical solution comprises the following steps: leading bridge arm turn-off current i Lr (t 1 ) Delayed bridge arm off current i Lr (t 3 )。
By the improved time domain analysis method of the first aspect, parameters such as direct current gain of the LLC series resonant converter and resonant current at any moment can be accurately calculated. Compared with the traditional method, the method has the advantages that the calculation complexity is slightly increased, the calculation accuracy is greatly increased, and the accuracy of dead time control is ensured.
In a second aspect, a dead time calculation and numerical fitting method is provided, which is used for accurately calculating the dead time required by two bridge arms of an LLC series resonant converter to just realize ZVS, and obtaining the functional relation between the required dead time and the circuit software and hardware parameters by using a simpler fitting function fitting.
The method is concretely realized as follows:
step A1: obtaining the distribution parameters of the primary side switching tube of the LLC series resonant converter, including the output capacitance C of the primary side switching tube oss 。
Step A2: the improved time domain analysis method provided in connection with the first aspect solves for obtaining the minimum dead time required for the switching tube.
Figure BDA0003762114320000063
Wherein i off The turn-off current of the finger bridge arm is i Lr (t 1 ) With lagging bridge arm turned off as current i Lr (t 3 )。
Step A3: the dead time is simplified to a simple function of the switching frequency and the phase shift angle by fitting the dead time under different load resistances using mathematical software (e.g., mathcad, etc.). Wherein the fitting function is:
Figure BDA0003762114320000071
wherein, f s Denotes switching frequency, theta denotes phase shift angle, C 0 -C 5 Are parameters to be fitted.
Through the dead time calculation and fitting method in the second aspect, the dead time required by ZVS of the switching tube in the hybrid control LLC series resonant converter can be obtained through accurate calculation. And after the function is used for parameter fitting, the complexity of dead time calculation is greatly reduced, and the method is convenient to use in an embedded controller.
In a third aspect, a hybrid control LLC series resonant converter bridge arm dead-zone-variable control method is provided for accurately controlling dead-zone time of two bridge arms and adapting to changes of circuit working conditions. The method comprises the following steps:
step B1: the sampling circuit outputs voltage and current, and the output voltage is regulated and stabilized through closed-loop control.
And step B2: and (4) substituting the fitting function according to control parameters (switching frequency and phase shift angle) stored in the controller, and respectively calculating to obtain the optimal dead time of the leading bridge arm and the optimal dead time of the lagging bridge arm.
And step B3: on the basis of the optimal dead time of the two bridge arms, a small amount of margin is respectively added, and then corresponding driving signals are output to a driving circuit and control the switch of the switching tube.
The invention and the corresponding principles are further described in the following with reference to the drawings of the specification:
FIG. 1 is a full bridge LLC series resonant converter topology, wherein V i Is the input voltage; Q1-Q4 are switching tubes to form a full-bridge inverter circuit; c q1 -C q4 A parasitic output capacitor of the switching tube; l is r Is a resonant inductor, C r Being resonant capacitors, T x Is a transformer with a center tap and L excitation inductance m D1 and D2 are rectifier diodes constituting a full-wave rectifier circuit, C o To output filter capacitors, R o Is a load resistor. In this embodiment, the switching tube preferably uses gan hemt, which has no parasitic body diode in its body, but can still conduct in the reverse direction due to the symmetry of the structure, and has no reverse recovery process of a common diode, and the disadvantage is that its reverse conduction voltage drop is large, and long-time reverse conduction will cause large loss.
Fig. 2 is a time domain waveform diagram of the full-bridge LLC resonant converter operating in the phase-shift frequency-modulation hybrid control mode. In this embodiment, the secondary rectifier current is discontinuous. Fig. 3 is an equivalent circuit diagram of the resonant cavity at different time periods. The analysis method will be described in time-division with reference to the time domain waveform diagram and the equivalent circuit diagram. For convenience of analysis, t 0 The time instant indicates t =0.
[t 0 -t 1 ]: and the switching tubes Q1 and Q4 are switched on, the input voltage of the resonant cavity is the power supply voltage, the secondary side diode is conducted, the excitation inductor is clamped by the output voltage, and only the resonant inductor and the resonant capacitor participate in resonance. The equivalent circuit diagram at this time is shown in fig. 3 (a). The resonance current, the excitation current and the resonance capacitor voltage satisfy the following equation:
Figure BDA0003762114320000081
[t 1 -t 2 ]: the switch tube Q1 is turned off, and Q3 and Q4 are turned on. The input of the resonant cavity is short-circuited, the secondary side rectifier diode continues to flow current, and the resonant current is rapidly reduced. The equivalent circuit diagram of this period is shown in fig. 3 (b). The resonance current, the excitation current and the resonance capacitor voltage respectively satisfy the equation:
Figure BDA0003762114320000082
[t 2 -t 3 ]: at t 2 At the moment, the current of the secondary side rectifier diode is reduced to zero, and the diode is naturally turned off. The excitation inductor is no longer clamped by the output voltage and participates in resonance. The equivalent circuit diagram of this period is shown in fig. 3 (c). The conventional analysis method considers that the excitation inductance is much larger than the resonance inductance, and therefore the excitation current in this period is regarded as a constant value. However, this is at larger phase shift angles or L m /L r Smaller, will cause larger analysis errors. In the other method, a differential equation is directly used for calculation, but the overall calculation amount is large and is not intuitive. The invention improves the traditional analysis method, linearizes the excitation current in the time period by t 2 And t 3 And taking the average value of the change rate of the resonance current at the moment as the change rate of the resonance current in the whole time period. At t 2 -t 3 In the time period, as shown in fig. 3 (c) of the equivalent circuit, the resonance current and the resonance capacitor voltage satisfy the relationship:
Figure BDA0003762114320000083
therefore, the rate k at which the resonant current changes at any instant is:
Figure BDA0003762114320000084
then at t 2 And t 3 Average value k of the rate of change of the resonant current at a time 23 Comprises the following steps:
Figure BDA0003762114320000085
setting t 2 -t 3 In time period, the resonant current is at k 23 The rate of time, then the equation in this period reduces to:
Figure BDA0003762114320000091
in addition, when the system is stable, the resonance waveforms of adjacent half switching periods have the same shape and are symmetrical about the time axis, so that the following steps are provided:
Figure BDA0003762114320000092
in the above equation, L m ；L r ；C r The parameters are circuit hardware parameters which are determined at the time of design and are known parameters; t is t 0 And t 2 The switching frequency f can be controlled by a known parameter s And the phase shift angle theta.
Therefore, the direct current gain M = nV of the hybrid control LLC series resonant converter can be obtained by mathematical software in the joint type (6), (7), (10), (11) and (12) o /V i The numerical solution of (c). If V is set o And V i One of them, the above equation can be further solved to obtain the values of the resonant current, the exciting current and the resonant capacitor voltage at any time.
Further in accordance withFIG. 2 shows a waveform t 1 And t 3 The resonance current at the moment is the turn-off current of the leading bridge arm and the lagging bridge arm. At t 1 And t 3 At this time, the current flows in the main circuit as shown in fig. 4 and 5, and after the switching tubes are turned off, the resonant current charges and discharges the output capacitors of the two switching tubes of the bridge arm. In order to ensure that the primary side full bridge of the LLC resonant converter can reliably realize zero voltage switching-on, the output capacitor of the switching tube needs to be ensured to completely complete charging and discharging within the dead time. Therefore, the shortest dead time of the leading and lagging bridge arms is:
Figure BDA0003762114320000093
wherein, C oss Outputting a capacitor for the primary side switching tube; i.e. i off I leading and lagging bridge arms for switching off current of switching tube off Are each t 1 And t 3 Resonance current i at time Lr (t 1 )、i Lr (t 3 )。
Because the phase advance bridge arm switching tubes Q1 and Q3 turn off the current i Lr (t 1 ) Bridge arm switching tube Q2 and Q4 turn-off current i with more than phase lag Lr (t 3 ) Therefore, the dead time required by leading bridge arm switches is shorter, and the dead time required by lagging bridge arm switches is larger. The driving signals with different dead zones are given to the two bridge arm switching tubes, so that the reverse conduction loss of the switching tubes can be reduced, and further, the loss is reduced.
The method can accurately calculate the dead time required by the switching tube, but the total calculation amount is large. In practical applications, although the calculation can be performed again after a plurality of switching cycles, the general embedded controller still has difficulty in burden. To facilitate the use of the variable dead zone control method described above in an embedded controller, the method by parameter fitting is further described below to reduce the amount of computation.
When the switching frequency f s Phase shift angle theta and load resistance R o When the current is kept unchanged, the direct current gain M of the circuit is unchanged, the shape of the current waveform of the resonant cavity is irrelevant to the input and output voltage, and the current magnitude is irrelevant to the inputThe output voltage is proportional. According to equation (13), the minimum dead time required for the switching tube is also proportional to the input voltage. Therefore, the minimum dead time required by the switching tube is independent of the input and output voltage and is only controlled by the switching frequency f s Phase shift angle theta and load resistance R o Three variables are determined.
When the output resistance is not changed, the minimum dead time required by the switching tube is the switching frequency f s And phase shift angle theta. Calculating different switching frequencies f by the improved time domain analysis method s And the minimum dead time required for the switching tube in the case of the phase shift angle theta. Further, the function is fitted using a fitting function as follows:
Figure BDA0003762114320000101
wherein, C 0 -C 5 Are parameters to be fitted.
Further, for the load resistance variation, a plurality of possible load resistance values can be calculated to obtain different fitting functions. When the circuit runs, the actual load resistance value is detected firstly, and the minimum dead time required by the switching tube is estimated by using a linear interpolation method.
Based on the improved time domain analysis method and the bridge arm optimal dead zone time calculation and parameter fitting method, the bridge arm variable dead zone control method of the hybrid control LLC series resonant converter can be obtained. The principle is shown in fig. 6, and includes:
step 1: and according to hardware parameters and indexes, calculating the optimal dead time under different working conditions in an off-line manner.
Step 2: and fitting the dead time function by using the fitting function, and writing the calculated fitting function into the embedded controller.
And 3, step 3: when the circuit initially operates, the output voltage and the output current are acquired in real time, and the voltage and current closed-loop control is normally carried out. At the moment, the dead time of the circuit is longer, and the switching-on of the primary side switching tube due to the loss of zero voltage is prevented.
And 4, step 4: and after the circuit basically runs stably, obtaining parameters such as switching frequency, phase shift angle and the like from the closed-loop control module, substituting the parameters such as output voltage, output current and the like into a fitting function, and respectively calculating dead time required by the leading bridge arm and the lagging bridge arm.
And 5: and (4) increasing the dead time data calculated in the step (4) by a proper margin and transmitting the data to a PWM generator to output a driving signal containing the target dead time so as to reduce the reverse conduction loss of the switching tube.
Step 6: when the dead time calculator detects that parameters such as circuit output voltage, output current, switching frequency, phase shift angle and the like are greatly changed. And (5) repeating the step (4) and the step (5) again to ensure that the primary side switching tube can just realize zero voltage switching-on.
Therefore, the dead time calculator only needs to recalculate when a large change in the circuit operating conditions is detected, which further reduces the amount of computation of the embedded controller.
Waveforms of the PSIM simulation circuit to which the existing dead band control method and the new variable dead band control method of the present invention are applied are shown in (a) and (b) of fig. 7, respectively. When the converter works at the switching frequency of 700kHz and the phase shift angle of 100 degrees, in order to enable the two bridge arm switch tubes to realize zero-voltage switching-on, the dead time of the phase lead and the dead time of the lag bridge arm switch tubes in the existing method are consistent and are both 20ns. Under the same condition, after the novel variable dead zone control method is used, the dead zone time of the leading bridge arm switching tube is reduced to about 3ns, the dead zone time of the lagging bridge arm switching tube is reduced to about 17ns, and the two bridge arm switching tubes can be switched on at zero voltage just. The method ensures that the switching tube can realize zero voltage switching-on under different circuit operation conditions, greatly reduces the reverse conduction time of the switching tube and reduces the loss in the switching process.
While the invention has been described with reference to a preferred embodiment, it will be understood by those skilled in the art that various changes may be made and equivalents may be substituted for elements thereof without departing from the scope of the invention. However, any simple modification, equivalent change and modification of the above embodiments according to the technical essence of the present invention are within the protection scope of the technical solution of the present invention.
The present invention is not limited to the above-mentioned preferred embodiments, and various other methods for controlling the variable dead zone of the LLC series resonant converter in various forms can be found by anyone in the light of the present invention.

Claims (6)

1. A variable dead zone control method for a hybrid control LLC series resonant converter is characterized by comprising the following steps:
step S1: acquiring circuit hardware parameters and input and output parameter ranges, comprising: resonant inductor L r Resonant capacitance C r Transformer T x Primary side excitation inductance L m Transformer turn ratio n, primary side switch tube output capacitance C oss Input voltage V i Variation range, output voltage V o Range of variation, and output current I o A range of variation;
step S2: respectively calculating currents i at the turn-off time of a bridge arm switching tube for hybrid control of the LLC resonant converter to drive signals with leading and lagging phases under the condition of different input and output parameters by using an improved time domain analysis method off ；
The improved time domain analysis method comprises the following steps: in which the resonant current i is written in half a switching cycle at different time intervals Lr Exciting current i Lm And resonant capacitor voltage v Cr The satisfied equation is combined with the symmetry of the resonance waveform of the full-bridge LLC resonance converter in a steady state to solve the resonance current i at any moment Lr Exciting current i Lm And resonant capacitor voltage v Cr ；
And step S3: according to the parasitic capacitance C of the switch tube eq And turn-off current i off Calculating the shortest dead time t required by the primary side switching tube to realize zero voltage switching-on under the condition of obtaining different input and output parameters dead_min ；
Wherein, the shortest dead time is calculated by
Figure FDA0003762114310000011
C oss A parasitic output capacitor of the switching tube;
and step S4: using a simple fitting function model f (f) s θ) fitting the shortest dead time t dead_min And the switching frequency f s Solving to-be-determined parameters in the fitting function model according to the function relation of the phase shift angle theta;
step S5: solving a good fitting function model f (f) s Theta) writing the data into the embedded controller, and calculating in real time when the circuit runs to obtain the shortest dead time required by each switching tube on the primary side to realize zero voltage switching-on under the working condition;
step S6: and (5) adding a proper margin to the shortest dead time calculated in the step (S5), transmitting a final dead time signal to a PWM generator, and controlling the dead time of the primary side switching tube through a driving circuit.
2. The hybrid control LLC series resonant converter dead-zone control method according to claim 1, characterized in that: in the improved time domain analysis method, the excitation inductance L m Resonant inductance L r And a resonance capacitor C r Time period t of common resonance sp The resonant current is regarded as a linear change, and the rate of the linear change is t sp Average of the rate of change of the resonant current at the endpoint time.
3. The hybrid control LLC series resonant converter variable dead zone control method according to claim 1 or 2, characterized in that: the improved time domain analysis method specifically comprises the following steps:
step S21: obtaining specific software and hardware parameters of the controlled LLC series resonant converter during operation, including:
resonant inductor L r Excitation inductance L m Resonant capacitor C r Transformer transformation ratio n, switching frequency f s Phase shift angle θ, input voltage V i Output voltage V o Output current I o (ii) a Wherein, V i 、V o 、I o At least two of which should be obtained;
step S22: calculating the initial value conditions of the exciting current and the resonant capacitor voltage:
Figure FDA0003762114310000021
wherein i Lr A resonant inductor current transient; v. of Cr The instantaneous value of the voltage of the resonance capacitor; t is t 0～2 And t 2～3 Respectively mean t 0 To t 2 Time sum t 2 To t 3 Time of (t) 0 Is an initial moment of a switching cycle, t 2 The interruption time of the secondary side current in the first half of the switching period is unknown number t 3 Is the middle time of the switching cycle; k is a radical of 23 Finger resonant inductor current at t 2 To t 3 A rate of change over a period of time;
solving the equation set (1) to obtain an initial value i of the exciting current Lr (t 0 ) And initial value v of resonant capacitor voltage Cr (t 0 )；
Step S23: according to the circuit operation condition, list i Lr And v Cr At t 0 To t 3 Solving a differential equation satisfied in a time period by combining the initial value condition in the formula (1) to obtain i Lr And v Cr The corresponding differential equation is:
Figure FDA0003762114310000022
step S24: according to i Lr And u Cr Substituting the satisfied boundary conditions into the time domain expression obtained in the formula (3), eliminating the unknown number therein, and obtaining a final expression, wherein the corresponding boundary conditions are as follows:
Figure FDA0003762114310000023
step S25: solving the turn-off current of the two bridge arms under different working conditions by using a numerical solution method, comprising the following steps of: leading bridge arm turn-off current i Lr (t 1 ) Delayed bridge arm off current i Lr (t 3 )。
4. The hybrid control LLC series resonant converter dead-zone control method according to claim 1, characterized in that: the simple fitting function model used in step S4 is:
Figure FDA0003762114310000024
wherein f is s The switching frequency; theta indicates a phase shift angle of 0,C or greater 0 -C 5 Is a parameter to be fitted; the phase shift angle θ; and fitting the dead time under different load resistances, and simplifying the dead time into a simple function of the switching frequency and the phase shift angle.
5. The hybrid control LLC series resonant converter dead-zone control method according to claim 1, characterized in that: and step S5, calculating that the shortest dead time required by the bridge arm switching tube with the lagging phase of the obtained driving signal to realize zero voltage switching-on is greater than or equal to the shortest dead time required by the bridge arm switching tube with the leading phase to realize zero voltage switching-on.
6. The hybrid control LLC series resonant converter dead-zone control method according to claim 1, characterized in that: adopt voltage current sampling circuit to connect LLC series resonance converter's load in order to gather output voltage V o And an output current I o The information of (a); a driving circuit is adopted to be connected with four switching tubes of the LLC series resonant converter so as to provide driving signals; the embedded controller is connected with the voltage and current sampling circuit and the driving circuit.
CN202210875991.2A 2022-07-25 2022-07-25 Variable dead zone control method for hybrid control LLC series resonant converter Pending CN115149817A (en)

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Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN117728696A (en) * 2024-02-08 2024-03-19 浙江艾罗网络能源技术股份有限公司 Controller, double-active bridge converter and control method thereof

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN117728696A (en) * 2024-02-08 2024-03-19 浙江艾罗网络能源技术股份有限公司 Controller, double-active bridge converter and control method thereof
CN117728696B (en) * 2024-02-08 2024-05-28 浙江艾罗网络能源技术股份有限公司 Controller, double-active bridge converter and control method thereof

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