CN115149817A - Hybrid control LLC series resonant converter variable dead time control method - Google Patents

Hybrid control LLC series resonant converter variable dead time control method Download PDF

Info

Publication number
CN115149817A
CN115149817A CN202210875991.2A CN202210875991A CN115149817A CN 115149817 A CN115149817 A CN 115149817A CN 202210875991 A CN202210875991 A CN 202210875991A CN 115149817 A CN115149817 A CN 115149817A
Authority
CN
China
Prior art keywords
time
current
resonant
voltage
dead
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Pending
Application number
CN202210875991.2A
Other languages
Chinese (zh)
Inventor
毛行奎
张彬意
郑嘉冬
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Xiamen Lijing New Energy Technology Co ltd
Original Assignee
Fuzhou University
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Fuzhou University filed Critical Fuzhou University
Priority to CN202210875991.2A priority Critical patent/CN115149817A/en
Publication of CN115149817A publication Critical patent/CN115149817A/en
Pending legal-status Critical Current

Links

Images

Classifications

    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02MAPPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00Conversion of DC power input into DC power output
    • H02M3/22Conversion of DC power input into DC power output with intermediate conversion into AC
    • H02M3/24Conversion of DC power input into DC power output with intermediate conversion into AC by static converters
    • H02M3/28Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC
    • H02M3/325Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33573Full-bridge at primary side of an isolation transformer
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02MAPPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00Details of apparatus for conversion
    • H02M1/0048Circuits or arrangements for reducing losses
    • H02M1/0054Transistor switching losses
    • H02M1/0058Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02MAPPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00Details of apparatus for conversion
    • H02M1/08Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/083Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the ignition at the zero crossing of the voltage or the current
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02MAPPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00Details of apparatus for conversion
    • H02M1/08Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02MAPPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00Conversion of DC power input into DC power output
    • H02M3/01Resonant DC/DC converters
    • YGENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02BCLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)

Abstract

The invention provides a variable dead zone control method of a hybrid control LLC series resonant converter, which respectively calculates the current of leading and lagging phase positions of bridge arm switching moments and the shortest dead zone time required by realizing ZVS (zero voltage switching on), simplifies the calculated amount by using a parameter fitting method, and calculates and controls the dead zone time of two bridge arm switching tubes on an embedded controller in real time. The invention considers the difference of dead time required by zero voltage switching-on of the switching tubes of two bridge arms at the primary side of the hybrid control LLC series resonance converter, performs real-time control, prevents the switching tubes at the primary side from losing zero voltage switching-on due to working condition change, reduces reverse conduction time of the switching tubes as much as possible and reduces loss. The invention has better application effect in the high-frequency and wide-gain-range hybrid control LLC series resonant converter.

Description

混合控制LLC串联谐振变换器变死区控制方法Hybrid control LLC series resonant converter variable dead time control method

技术领域technical field

本发明涉及电力电子技术领域,特别涉及一种混合控制LLC串联谐振变换器变死区控制方法。The invention relates to the technical field of power electronics, in particular to a hybrid control LLC series resonant converter variable dead zone control method.

背景技术Background technique

LLC串联谐振DC/DC变换器,因其电路结构简单、开关性能优越等优点而受到广泛关注。其容易实现原边开关管零电压开通(ZVS)和副边整流管零电流关断(ZCS),便于提高开关频率并提高功率密度。LLC series resonant DC/DC converters have received extensive attention due to their simple circuit structure and superior switching performance. It is easy to realize the zero-voltage turn-on (ZVS) of the primary side switch tube and the zero-current turn-off (ZCS) of the secondary side rectifier tube, which is convenient to increase the switching frequency and improve the power density.

LLC串联谐振变换器一般采用变频控制策略,通过改变开关频率以获得相应的增益,稳定输出电压。但当输入输出电压范围较宽时,其频率变化范围大,导致磁性元件设计困难以及EMI设计困难等问题,故一般采用变频+移相的混合控制策略,减小频率变化范围。但当移相角逐渐增加,相位超前桥臂关断电流逐渐增加,相位滞后桥臂关断电流逐渐减小。为保证原边开关管是实现ZVS(零电压开通),需要给予原边桥臂合适的死区时间,使得在死区时间内桥臂关断电流能为开关管寄生输出电容充放电完成。The LLC series resonant converter generally adopts a frequency conversion control strategy, which can stabilize the output voltage by changing the switching frequency to obtain the corresponding gain. However, when the input and output voltage range is wide, the frequency variation range is large, which leads to difficulties in the design of magnetic components and EMI design. Therefore, a hybrid control strategy of frequency conversion + phase shift is generally used to reduce the frequency variation range. However, when the phase shift angle gradually increases, the turn-off current of the phase-leading bridge arm gradually increases, and the phase-lag bridge arm turn-off current gradually decreases. In order to ensure that the primary side switch tube is ZVS (zero voltage turn-on), it is necessary to give the primary side bridge arm a suitable dead time, so that the bridge arm turn-off current can be charged and discharged for the parasitic output capacitance of the switch tube during the dead time.

传统方法固定使用原边开关管实现ZVS所需最大的死区时间,并对两桥臂不加区分。但在高频混合控制LLC串联谐振变换器中,移相角较大时超前桥臂开关管所需的死区时间较小,其体二极管反向导通时间将延长,将造成较大的反向导通损耗。有现有技术对变频控制下LLC谐振变换器的变死区控制进行了分析,该方法能根据电路运行时的开关频率、负载情况等动态调整原边开关管的死区时间,但该方法并不适用于移相变频混合控制下的LLC谐振变换器。The traditional method uses the primary side switch to achieve the maximum dead time required for ZVS, and does not distinguish between the two bridge arms. However, in the high-frequency hybrid control LLC series resonant converter, when the phase shift angle is large, the dead time required by the switch of the leading bridge arm is small, and the reverse conduction time of the body diode will be prolonged, which will cause a large reverse conduction time. pass loss. The prior art analyzes the variable dead time control of LLC resonant converters under variable frequency control. This method can dynamically adjust the dead time of the primary side switch tube according to the switching frequency and load conditions during circuit operation, but this method does not work. It is not suitable for LLC resonant converters under phase-shift and frequency-conversion hybrid control.

发明内容SUMMARY OF THE INVENTION

因此,有必要提出一种针对混合控制LLC串联谐振变换器的死区时间控制方法,能针对电路运行情况动态调整两桥臂开关管的死区时间,防止死区时间较短导致开关管失去零电压开通状态,或死区时间过长导致开关管反向导通损耗大,以提升电路的整体运行效率。Therefore, it is necessary to propose a dead time control method for the hybrid control LLC series resonant converter, which can dynamically adjust the dead time of the switches of the two bridge arms according to the circuit operation, and prevent the switches from losing zero due to the short dead time. When the voltage is turned on, or the dead time is too long, the reverse conduction loss of the switch tube is large, so as to improve the overall operating efficiency of the circuit.

本发明的目的是公开一种混合控制LLC串联谐振变换器变死区控制方法,改善相位超前及相位滞后桥臂开关管死区设置不合理而导致电路运行效率降低的问题。本发明具有计算结果精确,计算量小,运行效率高,实施方便等优点。The purpose of the present invention is to disclose a hybrid control LLC series resonant converter variable dead zone control method, which improves the problem of reducing the circuit operation efficiency due to the unreasonable setting of the phase leading and phase lag bridge arm switch dead zones. The invention has the advantages of accurate calculation result, small calculation amount, high operation efficiency, convenient implementation and the like.

其分别计算相位超前与滞后桥臂开关时刻的电流以及实现零电压开通所需的最短死区时间,并使用参数拟合的方法简化计算量,在嵌入式控制器上实时计算并控制两桥臂开关管的死区时间。本发明考虑了混合控制LLC串联谐振变换器原边两个桥臂开关管实现零电压开通所需死区时间的不同,并进行了实时控制,防止原边开关管因工况变化失去零电压开通,并尽可能的减小了开关管反向导通时间,减小了损耗。本发明在高频、宽增益范围的混合控制LLC串联谐振变换器中具有较好的应用效果。It calculates the current at the switching time of the phase leading and lagging bridge arms and the shortest dead time required for zero-voltage turn-on respectively, and uses the method of parameter fitting to simplify the calculation amount, and calculates and controls the two bridge arms in real time on the embedded controller. Dead time of the switch. The present invention takes into account the difference in dead time required for zero-voltage turn-on of the two bridge arm switches on the primary side of the hybrid control LLC series resonant converter, and performs real-time control to prevent the primary-side switch from losing zero-voltage turn-on due to changes in operating conditions , and reduce the reverse conduction time of the switch tube as much as possible, reducing the loss. The invention has good application effect in the mixed control LLC series resonant converter with high frequency and wide gain range.

本发明具体采用以下技术方案:The present invention specifically adopts the following technical solutions:

一种混合控制LLC串联谐振变换器变死区控制方法,其特征在于,包括以下步骤:A hybrid control LLC series resonant converter variable dead zone control method is characterized in that, comprises the following steps:

步骤S1:获取电路硬件参数以及输入输出参数范围,包括:谐振电感Lr,谐振电容Cr,变压器Tx的原边励磁电感Lm,变压器匝比n,原边开关管输出电容Coss,输入电压Vi变化范围,输出电压Vo变化范围,以及输出电流Io变化范围;Step S1: Obtain circuit hardware parameters and the range of input and output parameters, including: resonant inductance L r , resonant capacitor Cr , primary excitation inductance L m of transformer T x , transformer turns ratio n, primary switch output capacitor C oss , Input voltage V i variation range, output voltage V o variation range, and output current I o variation range;

步骤S2:使用改进的时域分析方法分别计算不同输入输出参数情况下,混合控制LLC谐振变换器驱动信号相位超前和滞后的桥臂开关管的关断时刻的电流ioffStep S2: using the improved time-domain analysis method to calculate the current i off at the turn-off moment of the bridge arm switch tube of which the phase of the drive signal of the LLC resonant converter is mixed and controlled under different input and output parameters;

所述改进时域分析方法包括:列写出半个开关周期内,不同时段谐振电流iLr、励磁电流iLm与谐振电容电压vCr所满足的方程,并结合稳态时全桥LLC谐振变换器谐振波形的对称性,求解出任意时刻的谐振电流iLr、励磁电流iLm与谐振电容电压vCrThe improved time-domain analysis method includes: writing out the equations satisfied by the resonant current i Lr , the excitation current i Lm and the resonant capacitor voltage v Cr in different time periods within half a switching cycle, and combining the full-bridge LLC resonant transformation in steady state According to the symmetry of the resonant waveform of the resonator, the resonant current i Lr , the excitation current i Lm and the resonant capacitor voltage v Cr at any time are solved;

步骤S3:根据开关管寄生电容Ceq以及关断电流ioff,计算得到不同输入输出参数情况下原边开关管实现零电压开通所需的最短死区时间tdead_minStep S3: According to the parasitic capacitance C eq of the switch tube and the turn-off current i off , calculate and obtain the shortest dead time t dead_min required for the primary side switch tube to realize zero-voltage turn-on under different input and output parameters;

其中,最短死区时间计算方法为

Figure BDA0003762114320000021
Coss为开关管寄生输出电容;Among them, the calculation method of the shortest dead time is as follows
Figure BDA0003762114320000021
C oss is the parasitic output capacitance of the switch;

步骤S4:使用简单的拟合函数模型f(fs,θ)拟合最短死区时间tdead_min与开关频率fs和移相角θ的函数关系,求解出拟合函数模型中的待定参数;Step S4: use a simple fitting function model f(f s , θ) to fit the functional relationship between the shortest dead time t dead_min and the switching frequency f s and the phase shift angle θ, and solve the undetermined parameters in the fitting function model;

步骤S5:将求解好的拟合函数模型f(fs,θ)写入嵌入式控制器,并在电路运行时实时计算,得到此工况下原边各个的开关管实现零电压开通所需最短死区时间;Step S5: Write the solved fitting function model f(f s , θ) into the embedded controller, and calculate it in real time when the circuit is running to obtain the requirements for zero-voltage turn-on of each switch on the primary side under this working condition Minimum dead time;

步骤S6:给步骤S5计算获得的最短死区时间增加合适的裕度后,将最终死区时间信号传输至PWM发生器,并通过驱动电路控制原边开关管的死区时间。Step S6: After adding an appropriate margin to the shortest dead time calculated in step S5, the final dead time signal is transmitted to the PWM generator, and the dead time of the primary switch is controlled by the drive circuit.

进一步地,所述改进时域分析方法中,在励磁电感Lm,谐振电感Lr与谐振电容Cr共同谐振的时间段tsp内,将谐振电流视作线性变化,且线性变化的速率为tsp端点时刻谐振电流变化速率的平均值。Further, in the improved time domain analysis method, in the time period tsp during which the excitation inductance Lm , the resonant inductance Lr and the resonant capacitor Cr resonate together, the resonant current is regarded as a linear change, and the rate of the linear change is The average value of the rate of change of the resonant current at the tsp endpoint.

进一步地,所述改进时域分析方法具体包括:Further, the improved time domain analysis method specifically includes:

步骤S21:获得所控制的LLC串联谐振变换器运行时的具体软硬件参数,包括:Step S21: Obtain the specific software and hardware parameters of the controlled LLC series resonant converter during operation, including:

谐振电感Lr,励磁电感Lm,谐振电容Cr,变压器变比n,开关频率fs,移相角θ,输入电压Vi,输出电压Vo,输出电流Io;其中,Vi、Vo、Io至少应获得到其中之二;Resonant inductance L r , excitation inductance L m , resonant capacitor C r , transformer ratio n, switching frequency f s , phase shift angle θ, input voltage V i , output voltage V o , output current I o ; among them, V i , At least two of V o and I o should be obtained;

步骤S22:计算励磁电流与谐振电容电压初值条件:Step S22: Calculate the initial value conditions of the excitation current and the resonant capacitor voltage:

Figure BDA0003762114320000031
Figure BDA0003762114320000031

其中,iLr指谐振电感电流瞬时值;vCr指谐振电容电压瞬时值;t0~2和t2~3分别指t0至t2的时间和t2至t3的时间,t0为一个开关周期初始时刻,t2为前半个开关周期内副边电流断续时刻,为未知数,t3为开关周期的中间时刻;k23指谐振电感电流在t2至t3时间段内变化的速率;Among them, i Lr refers to the instantaneous value of the resonant inductor current; v Cr refers to the instantaneous value of the resonant capacitor voltage; t 0 ~ 2 and t 2 ~ 3 refer to the time from t 0 to t 2 and the time from t 2 to t 3 respectively, and t 0 is The initial moment of a switching cycle, t 2 is the discontinuous moment of the secondary side current in the first half of the switching cycle, which is unknown, and t 3 is the middle moment of the switching cycle; k 23 refers to the change of the resonant inductor current during the time period from t 2 to t 3 . rate;

求解方程组(1)以获得励磁电流初值iLr(t0)和谐振电容电压初值vCr(t0);Solve the equation set (1) to obtain the initial value of the excitation current i Lr (t 0 ) and the initial value of the resonance capacitor voltage v Cr (t 0 );

步骤S23:根据电路运行情况,列出iLr和vCr在t0至t3时间段内满足的微分方程,结合式(1)中的初值条件,求解得到iLr和vCr的时域表达式,相应微分方程为:Step S23: According to the operation of the circuit, list the differential equations that i Lr and v Cr satisfy in the time period from t 0 to t 3 , and combine the initial value conditions in equation (1) to solve the time domain to obtain i Lr and v Cr expression, the corresponding differential equation is:

Figure BDA0003762114320000032
Figure BDA0003762114320000032

步骤S24:根据iLr和uCr的满足的边界条件,带入式(3)中求得的时域表达式中,消除其中的未知数,得到最终的表达式,相应边界条件为:Step S24: According to the satisfied boundary conditions of i Lr and u Cr , bring it into the time domain expression obtained in formula (3), eliminate the unknowns therein, and obtain the final expression, and the corresponding boundary conditions are:

Figure BDA0003762114320000033
Figure BDA0003762114320000033

步骤S25:使用数值解法求出不同工况下两桥臂关断电流,包括:超前桥臂关断电流iLr(t1),滞后桥臂关断电流iLr(t3)。Step S25 : Use the numerical solution to obtain the off currents of the two bridge arms under different working conditions, including: the lead off current i Lr (t 1 ) and the lagging off current i Lr (t 3 ).

进一步地,在步骤S4中使用的简单拟合函数模型为:Further, the simple fitting function model used in step S4 is:

Figure BDA0003762114320000041
Figure BDA0003762114320000041

其中,fs指开关频率;θ指移相角,大于或等于0,C0-C5为待拟合参数;所述移相角θ;对不同负载电阻下的死区时间进行拟合,将死区时间简化为开关频率与移相角的简单函数。Among them, f s refers to the switching frequency; θ refers to the phase shift angle, which is greater than or equal to 0, and C 0 -C 5 are the parameters to be fitted; the phase shift angle θ; the dead time under different load resistances is fitted, Simplifies dead time as a simple function of switching frequency and phase shift angle.

进一步地,在步骤S5中计算获得的驱动信号相位滞后的桥臂开关管实现零电压开通所需最短死区时间大于或等于相位超前桥臂开关管实现零电压开通所需最短死区时间。Further, the shortest dead time required for zero-voltage turn-on of the bridge arm switch with the phase lag of the drive signal calculated in step S5 is greater than or equal to the shortest dead time required for zero-voltage turn-on of the phase-leading bridge arm switch.

进一步地,采用电压电流采样电路连接LLC串联谐振变换器的负载以采集输出电压Vo和输出电流Io的信息;采用驱动电路连接LLC串联谐振变换器的四个开关管以提供驱动信号;所述嵌入式控制器连接电压电流采样电路和驱动电路。Further, a voltage and current sampling circuit is used to connect the load of the LLC series resonant converter to collect the information of the output voltage V o and the output current I o ; a driving circuit is used to connect the four switching tubes of the LLC series resonant converter to provide a driving signal; The embedded controller is connected with the voltage and current sampling circuit and the driving circuit.

相比于现有技术,本发明及其优选方案提供的方案在完全不增加硬件电路资源的情况下,能根据输入输出的变化动态调整原边不同桥臂开关管的死区时间,使之工作在恰能实现ZVS的状态,防止工况变化导致部分开关管失去ZVS条件,并在高频及大移相角工作时极大的减小了原边开关管反向导通损耗。具有运算量少,便于实施,电路运行效率较高等优点。Compared with the prior art, the solution provided by the present invention and its preferred solution can dynamically adjust the dead time of the switch tubes of different bridge arms of the primary side according to the change of input and output without increasing the hardware circuit resources at all, so as to make it work. In the state where ZVS can be achieved, it can prevent some switches from losing ZVS conditions due to changes in working conditions, and greatly reduce the reverse conduction loss of the primary switch when working at high frequency and large phase angle. It has the advantages of less computation, easy implementation, and high circuit operation efficiency.

附图说明Description of drawings

下面结合附图和具体实施方式对本发明进一步详细的说明:The present invention will be described in further detail below in conjunction with the accompanying drawings and specific embodiments:

图1是本发明实施例全桥LLC串联谐振变换器拓扑图。FIG. 1 is a topology diagram of a full-bridge LLC series resonant converter according to an embodiment of the present invention.

图2是本发明实施例移相变频混合控制LLC串联谐振变换器时域波形图。FIG. 2 is a time-domain waveform diagram of a phase-shifting and frequency-changing hybrid control LLC series resonant converter according to an embodiment of the present invention.

图3是本发明实施例全桥LLC串联谐振变换器工作在不同时段的谐振腔等效电路图。3 is an equivalent circuit diagram of a resonant cavity when the full-bridge LLC series resonant converter according to an embodiment of the present invention operates in different time periods.

图4是本发明实施例全桥LLC串联谐振变换器相位超前桥臂上开关管关断时刻等效电路图。FIG. 4 is an equivalent circuit diagram of a full-bridge LLC series resonant converter in accordance with an embodiment of the present invention when the phase leads the switch tube on the bridge arm when the switch is turned off.

图5是本发明实施例全桥LLC串联谐振变换器相位滞后桥臂下开关管关断时刻等效电路图。FIG. 5 is an equivalent circuit diagram of a full-bridge LLC series resonant converter at the turn-off time of the lower switch of the bridge arm with a phase lag according to an embodiment of the present invention.

图6是本发明实施例混合控制LLC串联谐振变换器变死区控制方法原理图。FIG. 6 is a schematic diagram of a method for controlling variable dead zone of a hybrid control LLC series resonant converter according to an embodiment of the present invention.

图7是本发明实施例已有死区控制方法与本发明所述变死区控制方法PSIM仿真波形对比图。FIG. 7 is a PSIM simulation waveform comparison diagram of the existing dead zone control method according to the embodiment of the present invention and the variable dead zone control method of the present invention.

具体实施方式Detailed ways

为让本专利的特征和优点能更明显易懂,下文特举实施例,作详细说明如下。In order to make the features and advantages of the present patent more obvious and easy to understand, the following examples are given and described in detail as follows.

应该指出,以下详细说明都是示例性的,旨在对本申请提供进一步的说明。除非另有指明,本文使用的所有技术和科学术语具有与本申请所属技术领域的普通技术人员通常理解的相同含义。It should be noted that the following detailed description is exemplary and intended to provide further explanation of the application. Unless otherwise defined, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this application belongs.

需要注意的是,这里所使用的术语仅是为了描述具体实施方式,而非意图限制根据本申请的示例性实施方式。如在这里所使用的,除非上下文另外明确指出,否则单数形式也意图包括复数形式,此外,还应当理解的是,当在本说明书中使用术语“包含”和/或“包括”时,其指明存在特征、步骤、操作、器件、组件和/或它们的组合。It should be noted that the terminology used herein is for the purpose of describing specific embodiments only, and is not intended to limit the exemplary embodiments according to the present application. As used herein, unless the context clearly dictates otherwise, the singular is intended to include the plural as well, furthermore, it is to be understood that when the terms "comprising" and/or "including" are used in this specification, it indicates that There are features, steps, operations, devices, components and/or combinations thereof.

在本发明提供的方案中:In the scheme provided by the present invention:

第一方面,提供一种改进时域分析方法,应用于移相、调频混合控制全桥LLC串联谐振变换器,求得相位超前、滞后桥臂的关断电流。全桥LLC串联谐振变换器依次包括输入电容,逆变桥,谐振腔,整流滤波电路。其中逆变桥包括四个开关管组成的两个半桥桥臂,若两桥臂的驱动信号间的相位差θ大于零,则驱动信号相位超前的桥臂为超前桥臂,反之为滞后桥臂;谐振腔包括谐振电感、变压器、谐振电容,并由其级联构成;整流滤波电路包括全波、全桥或倍压等整流电路和电容滤波电路,用于将变压器输出的交流电转化为直流电。In the first aspect, an improved time domain analysis method is provided, which is applied to a phase-shift and frequency-modulated hybrid control full-bridge LLC series resonant converter to obtain the turn-off currents of the phase leading and lagging bridge arms. The full-bridge LLC series resonant converter sequentially includes an input capacitor, an inverter bridge, a resonant cavity, and a rectifier filter circuit. The inverter bridge includes two half-bridge arms composed of four switch tubes. If the phase difference θ between the driving signals of the two bridge arms is greater than zero, the bridge arm with the driving signal phase leading is the leading bridge arm, otherwise it is the lagging bridge. arm; resonant cavity includes resonant inductor, transformer, resonant capacitor, and is composed of cascade; rectifier filter circuit includes full-wave, full-bridge or voltage doubler rectifier circuit and capacitor filter circuit, which is used to convert the AC power output by the transformer into DC power .

该方法具体实现如下:The specific implementation of this method is as follows:

步骤S21:获得所控制的LLC串联谐振变换器运行时的具体软硬件参数,包括:Step S21: Obtain the specific software and hardware parameters of the controlled LLC series resonant converter during operation, including:

谐振电感Lr,励磁电感Lm,谐振电容Cr,变压器变比n,开关频率fs,移相角θ,输入电压Vi,输出电压Vo,输出电流Io。其中,Vi、Vo、Io至少应获得到其中之二。Resonant inductance L r , excitation inductance L m , resonant capacitor C r , transformer ratio n, switching frequency f s , phase shift angle θ, input voltage V i , output voltage V o , output current I o . Among them, at least two of Vi , V o and I o should be obtained.

步骤S22:计算励磁电流与谐振电容电压初值条件:Step S22: Calculate the initial value conditions of the excitation current and the resonant capacitor voltage:

Figure BDA0003762114320000051
Figure BDA0003762114320000051

其中,iLr指谐振电感电流瞬时值;vCr指谐振电容电压瞬时值;t0~2和t2~3分别指t0至t2的时间和t2至t3的时间,t0为一个开关周期初始时刻,t2为前半个开关周期内副边电流断续时刻,为未知数,t3为开关周期的中间时刻;k23指谐振电感电流在t2至t3时间段内变化的速率。Among them, i Lr refers to the instantaneous value of the resonant inductor current; v Cr refers to the instantaneous value of the resonant capacitor voltage; t 0 ~ 2 and t 2 ~ 3 refer to the time from t 0 to t 2 and the time from t 2 to t 3 respectively, and t 0 is The initial moment of a switching cycle, t 2 is the discontinuous moment of the secondary side current in the first half of the switching cycle, which is unknown, and t 3 is the middle moment of the switching cycle; k 23 refers to the change of the resonant inductor current during the time period from t 2 to t 3 . rate.

求解方程组(1)即可求得励磁电流初值iLr(t0)和谐振电容电压初值vCr(t0)。Solving equations (1) can obtain the initial value of the excitation current i Lr (t 0 ) and the initial value of the resonant capacitor voltage v Cr (t 0 ).

步骤S23:根据电路运行情况,列出iLr和vCr在t0至t3时间段内满足的微分方程,结合式(1)中的初值条件,求解得到iLr和vCr的时域表达式。相应微分方程为:Step S23: According to the operation of the circuit, list the differential equations that i Lr and v Cr satisfy in the time period from t 0 to t 3 , and combine the initial value conditions in equation (1) to solve the time domain to obtain i Lr and v Cr expression. The corresponding differential equation is:

Figure BDA0003762114320000061
Figure BDA0003762114320000061

注意到,此时求得的时域表达式包含未知数t2,以及Vi、Vo、Io其中之一。Note that the time domain expression obtained at this time includes the unknown t 2 , and one of V i , V o , and I o .

步骤S24:根据iLr和uCr的满足的边界条件,带入式(3)中求得的时域表达式中,消除其中的未知数,得到最终的表达式。相应边界条件为:Step S24: According to the satisfied boundary conditions of i Lr and u Cr , bring it into the time domain expression obtained in formula (3), eliminate the unknowns therein, and obtain the final expression. The corresponding boundary conditions are:

Figure BDA0003762114320000062
Figure BDA0003762114320000062

步骤S25:使用数值解法求出不同工况下两桥臂关断电流,包括:超前桥臂关断电流iLr(t1),滞后桥臂关断电流iLr(t3)。Step S25 : Use the numerical solution to obtain the off currents of the two bridge arms under different working conditions, including: the lead off current i Lr (t 1 ) and the lagging off current i Lr (t 3 ).

通过第一方面的改进时域分析方法,可以精确的计算得到LLC串联谐振变换器的直流增益,任意时刻的谐振电流等参数。相比与传统方法少量增加了计算复杂性而大幅度的增加了计算的精确性,确保了死区时间控制的精确性。Through the improved time domain analysis method in the first aspect, parameters such as the DC gain of the LLC series resonant converter and the resonant current at any time can be accurately calculated. Compared with the traditional method, the computational complexity is slightly increased and the calculation accuracy is greatly increased, ensuring the accuracy of the dead time control.

第二方面,提供一种死区时间计算以及数值拟合方法,用于精确计算LLC串联谐振变换器两桥臂恰能实现ZVS所需的死区时间,并使用较简单的拟合函数拟合得到所需的死区时间与电路软硬件参数之间的函数关系。In the second aspect, a dead time calculation and numerical fitting method are provided, which are used to accurately calculate the dead time required for the two bridge arms of the LLC series resonant converter to achieve ZVS, and use a simpler fitting function to fit Obtain the functional relationship between the required dead time and the circuit software and hardware parameters.

该方法具体实现如下:The specific implementation of this method is as follows:

步骤A1:获得所控制的LLC串联谐振变换器原边开关管的分布参数,包括原边开关管输出电容CossStep A1: Obtain the distribution parameters of the primary side switch tubes of the LLC series resonant converter controlled, including the output capacitance C oss of the primary side switch tubes.

步骤A2:结合第一方面所提的改进时域分析方法求解得到开关管所需的最小死区时间。Step A2: Combining with the improved time domain analysis method proposed in the first aspect, solve and obtain the minimum dead time required by the switching tube.

Figure BDA0003762114320000063
Figure BDA0003762114320000063

其中,ioff指桥臂关断电流,超前桥臂关断电流为iLr(t1),滞后桥臂关断为电流iLr(t3)。Among them, i off refers to the off current of the bridge arm, the off current of the leading bridge arm is i Lr (t 1 ), and the off current of the lagging bridge arm is i Lr (t 3 ).

步骤A3:使用数学软件(如mathcad等)对不同负载电阻下的死区时间进行拟合,将死区时间简化为开关频率与移相角的简单函数。其中拟合函数为:Step A3: Use mathematical software (such as mathcad, etc.) to fit the dead time under different load resistances, and simplify the dead time to a simple function of switching frequency and phase shift angle. The fitting function is:

Figure BDA0003762114320000071
Figure BDA0003762114320000071

其中,fs指开关频率,θ指移相角,C0-C5为待拟合参数。Among them, f s refers to the switching frequency, θ refers to the phase shift angle, and C 0 -C 5 are the parameters to be fitted.

通过第二方面的死区时间计算及拟合方法,可以精确计算得到混合控制LLC串联谐振变换器中,开关管恰能实现ZVS所需的死区时间。并且,使用函数进行参数拟合后,大大降低了死区时间计算的复杂度,便于在嵌入式控制器中使用本方法。Through the dead-time calculation and fitting method of the second aspect, the dead-time required by the switch tube to realize ZVS in the hybrid control LLC series resonant converter can be accurately calculated. In addition, after using the function to perform parameter fitting, the complexity of the dead time calculation is greatly reduced, and it is convenient to use the method in the embedded controller.

第三方面,提供一种混合控制LLC串联谐振变换器桥臂变死区控制方法,用于精确控制两桥臂的死区时间,适应电路工况的变化。包括:In a third aspect, a hybrid control LLC series resonant converter bridge arm variable dead time control method is provided, which is used to precisely control the dead time of the two bridge arms and adapt to changes in circuit operating conditions. include:

步骤B1:采样电路输出电压和输出电流,并通过闭环控制调节输出电压稳定。Step B1: Sample the output voltage and output current of the circuit, and adjust the output voltage to stabilize through closed-loop control.

步骤B2:根据控制器内部存储的控制参数(开关频率,移相角),带入上述拟合函数,分别计算得到超前和滞后桥臂的最优死区时间。Step B2: According to the control parameters (switching frequency, phase shift angle) stored in the controller, the above-mentioned fitting function is brought into, and the optimal dead time of the leading and lagging bridge arms is calculated respectively.

步骤B3:在两桥臂最优死区时间的基础上,分别增加少量裕度后,输出相应的驱动信号给驱动电路并控制开关管的开关。Step B3: On the basis of the optimal dead time of the two bridge arms, after adding a small amount of margin respectively, output the corresponding driving signal to the driving circuit and control the switching of the switch tube.

以下结合说明书附图对本发明方案及对应的原理做进一步介绍:The scheme of the present invention and the corresponding principle are further introduced below in conjunction with the accompanying drawings:

图1为全桥LLC串联谐振变换器拓扑,其中Vi为输入电压;Q1-Q4为开关管,构成全桥逆变电路;Cq1-Cq4为开关管寄生输出电容;Lr为谐振电感,Cr为谐振电容,Tx为带中心抽头的变压器,变压器励磁电感为Lm,D1、D2为整流二极管,构成全波整流电路,Co为输出滤波电容,Ro为负载电阻。在本实施例中,作为优选,开关管使用GaNHEMT,其体内无寄生体二极管,但由于结构的对称性,仍能反向导通,且无普通二极管的反向恢复过程,缺点是其反向导通压降较大,长时间反向导通将造成较大损耗。Figure 1 shows the topology of the full-bridge LLC series resonant converter, where V i is the input voltage; Q1-Q4 are switching tubes, forming a full-bridge inverter circuit; C q1 -C q4 are the parasitic output capacitances of the switching tubes; L r is the resonant inductance , C r is the resonant capacitor, T x is the transformer with center tap, the transformer excitation inductance is L m , D1 and D2 are rectifier diodes, forming a full-wave rectifier circuit, C o is the output filter capacitor, and Ro is the load resistance. In this embodiment, GaN HEMT is preferably used as the switch tube, which has no parasitic body diode in its body, but it can still conduct reverse conduction due to the symmetry of the structure, and there is no reverse recovery process of ordinary diodes. The disadvantage is that it conducts reversely. The pressure drop is large, and the reverse conduction for a long time will cause a large loss.

图2为上述全桥LLC谐振变换器工作在移相调频混合控制模式下的时域波形图。在本实施例中,副边整流管电流断续。图3为不同时间段下的谐振腔等效电路图。下面结合时域波形图和等效电路图,分时段对分析方法进行叙述。为方便分析,t0时刻表明t=0。FIG. 2 is a time-domain waveform diagram of the above-mentioned full-bridge LLC resonant converter operating in a phase-shift and frequency-modulation hybrid control mode. In this embodiment, the current of the secondary rectifier tube is intermittent. FIG. 3 is an equivalent circuit diagram of a resonant cavity at different time periods. In the following, the analysis method is described by time-domain waveform diagram and equivalent circuit diagram. For the convenience of analysis, time t 0 indicates t=0.

[t0-t1]:开关管Q1,Q4开通,谐振腔输入电压为电源电压,副边二极管导通,励磁电感被输出电压钳位,仅谐振电感和谐振电容参与谐振。此时段等效电路图如图3(a)所示。谐振电流,励磁电流和谐振电容电压满足方程如下:[t 0 -t 1 ]: The switches Q1 and Q4 are turned on, the input voltage of the resonant cavity is the power supply voltage, the secondary diode is turned on, the excitation inductance is clamped by the output voltage, and only the resonant inductor and resonant capacitor participate in the resonance. The equivalent circuit diagram of this period is shown in Figure 3(a). The resonant current, the excitation current and the resonant capacitor voltage satisfy the equations as follows:

Figure BDA0003762114320000081
Figure BDA0003762114320000081

[t1-t2]:开关管Q1关断,Q3、Q4开通。谐振腔输入短路,副边整流二极管继续续流,谐振电流迅速下降。此时段等效电路图如图3(b)所示。谐振电流,励磁电流和谐振电容电压分别满足方程:[t 1 -t 2 ]: Switch tube Q1 is turned off, and Q3 and Q4 are turned on. The input of the resonant cavity is short-circuited, the secondary side rectifier diode continues to flow freely, and the resonant current drops rapidly. The equivalent circuit diagram of this period is shown in Figure 3(b). The resonant current, the excitation current and the resonant capacitor voltage satisfy the equations respectively:

Figure BDA0003762114320000082
Figure BDA0003762114320000082

[t2-t3]:在t2时刻,副边整流二极管电流下降为零,二极管自然关断。励磁电感不再被输出电压钳位,参与谐振。其此时段等效电路图如图3(c)所示。传统分析方法认为励磁电感远远大于谐振电感,因此将此时段内的励磁电流视作恒定值。然而这在移相角较大时或Lm/Lr较小时,将造成较大的分析误差。还有方法直接使用微分方程进行计算,但整体计算量大且不直观。本发明改进传统分析方法,将此时段内的励磁电流线性化,以t2和t3时刻谐振电流变化速率的平均值作为整时段内谐振电流变化的速率。在t2-t3时段内,根据等效电路图3(c)所示,谐振电流与谐振电容电压满足关系:[t 2 -t 3 ]: At time t 2 , the secondary side rectifier diode current drops to zero, and the diode is naturally turned off. The magnetizing inductance is no longer clamped by the output voltage and participates in resonance. The equivalent circuit diagram of this period is shown in Figure 3(c). The traditional analysis method considers that the excitation inductance is much larger than the resonant inductance, so the excitation current during this period is regarded as a constant value. However, when the phase shift angle is large or L m /L r is small, it will cause a large analysis error. There are also methods to perform calculations directly using differential equations, but the overall calculation is computationally expensive and unintuitive. The present invention improves the traditional analysis method, linearizes the excitation current in this period, and takes the average value of the resonant current change rate at t2 and t3 as the resonant current change rate in the whole period. In the period of t 2 -t 3 , according to the equivalent circuit shown in Figure 3(c), the resonant current and the resonant capacitor voltage satisfy the relationship:

Figure BDA0003762114320000083
Figure BDA0003762114320000083

因此,任意时刻谐振电流变化的速率k为:Therefore, the rate of change of the resonant current k at any time is:

Figure BDA0003762114320000084
Figure BDA0003762114320000084

则在t2和t3时刻的谐振电流变化速率的平均值k23为:Then the average value k 23 of the rate of change of the resonant current at times t 2 and t 3 is:

Figure BDA0003762114320000085
Figure BDA0003762114320000085

设定t2-t3时段内,谐振电流以k23的速率线性变化,则此时段内方程简化为:Assuming that during the period of t 2 -t 3 , the resonant current changes linearly at the rate of k 23 , then the equation during this period is simplified as:

Figure BDA0003762114320000091
Figure BDA0003762114320000091

此外,由于系统在稳定时,相邻半个开关周期的谐振波形形状相同,并关于时间轴对称,因此有:In addition, when the system is stable, the resonant waveforms of adjacent half switching cycles have the same shape and are symmetrical about the time axis, so there are:

Figure BDA0003762114320000092
Figure BDA0003762114320000092

在上述方程中,Lm;Lr;Cr为电路硬件参数,在设计时已经确定,为已知参数;t0和t2可以通过已知参数开关频率fs和移相角θ求得。In the above equations, L m ; L r ; C r are circuit hardware parameters, which have been determined during design and are known parameters; t 0 and t 2 can be obtained by known parameters switching frequency f s and phase shift angle θ .

因此联立式(6)(7)(10)(11)(12),通过数学软件,即可求得混合控制LLC串联谐振变换器的直流增益M=nVo/Vi的数值解。若设定Vo与Vi其中之一,则可进一步解上述方程求得任意时刻谐振电流、励磁电流、谐振电容电压的值。Therefore, the numerical solution of the DC gain M=nV o /V i of the hybrid control LLC series resonant converter can be obtained through mathematical software. If one of V o and V i is set, the above equations can be further solved to obtain the values of resonant current, excitation current, and resonant capacitor voltage at any time.

进一步的,根据图2所示波形图,t1和t3时刻的谐振电流即为超前桥臂和滞后桥臂的关断电流。在t1和t3时刻,主电路中电流流向分别如图4和图5所示,在开关管关断后,谐振电流将为此桥臂的两个开关管的输出电容充放电。为保证LLC谐振变换器的原边全桥能可靠的实现零电压开通,需保证开关管的输出电容在死区时间内完全完成充放电。因此超前与滞后桥臂最短死区时间为:Further, according to the waveform diagram shown in FIG. 2 , the resonant currents at t 1 and t 3 are the turn-off currents of the leading bridge arm and the lagging bridge arm. At t 1 and t 3 , the current flow in the main circuit is shown in Figure 4 and Figure 5, respectively. After the switch is turned off, the resonant current will charge and discharge the output capacitors of the two switches of the bridge arm. In order to ensure that the primary side full bridge of the LLC resonant converter can reliably realize zero-voltage turn-on, it is necessary to ensure that the output capacitor of the switch tube is fully charged and discharged within the dead time. Therefore, the shortest dead time of the leading and lagging bridge arms is:

Figure BDA0003762114320000093
Figure BDA0003762114320000093

其中,Coss为原边开关管输出电容;ioff为开关管关断电流,超前和滞后桥臂的ioff分别为t1和t3时刻的谐振电流iLr(t1)、iLr(t3)。Among them, C oss is the output capacitance of the primary switch tube; i off is the switch-off current, and the i off of the leading and lagging bridge arms are the resonant currents i Lr ( t 1 ) and i Lr ( t3 ).

由于相位超前桥臂开关管Q1、Q3关断电流iLr(t1)大于相位滞后桥臂开关管Q2、Q4关断电流iLr(t3),因此,超前桥臂开关所需死区时间较短,滞后桥臂开关所需死区时间较大。对两桥臂开关管给予不同死区的驱动信号,能减小开关管反向导通损耗,进而减小损耗。Since the turn-off current i Lr (t 1 ) of the phase-leading bridge arm switches Q1 and Q3 is greater than the phase-lag bridge arm switch Q2 and Q4 turn-off current i Lr (t 3 ), the dead time required for the advance bridge arm switch It is shorter, and the dead time required for the switching of the lagging bridge arm is larger. Giving driving signals of different dead zones to the switch tubes of the two bridge arms can reduce the reverse conduction loss of the switch tubes, thereby reducing the loss.

上述方法能较为精确的计算开关管所需的死区时间,但总体计算量较大。实际应用时,虽然可以在多次开关周期后再计算一次,但一般嵌入式控制器仍然难以负担。为便于在嵌入式控制器中使用上述变死区控制方法,下面进一步介绍通过参数拟合的方法,以减小计算量。The above method can more accurately calculate the dead time required by the switching tube, but the overall calculation amount is large. In practical applications, although it can be calculated again after many switching cycles, the general embedded controller is still unaffordable. In order to facilitate the use of the above variable dead zone control method in the embedded controller, the method through parameter fitting is further introduced below to reduce the amount of calculation.

当开关频率fs、移相角θ和负载电阻Ro保持不变时,电路直流增益M不变,谐振腔电流波形的形状与输入输出电压无关,电流大小与输入输出电压成正比。根据式(13),开关管所需最小死区时间同样与输入电压成正比。因此开关管所需最小死区时间与输入输出电压无关,仅由开关频率fs、移相角θ和负载电阻Ro三个变量确定。When the switching frequency f s , the phase shift angle θ and the load resistance Ro remain unchanged, the DC gain M of the circuit remains unchanged, the shape of the resonator current waveform has nothing to do with the input and output voltages, and the current size is proportional to the input and output voltages. According to equation (13), the minimum dead time required by the switch is also proportional to the input voltage. Therefore, the minimum dead time required by the switch tube has nothing to do with the input and output voltage, and is only determined by three variables, the switching frequency f s , the phase shift angle θ and the load resistance R o .

在输出电阻不发生变化时,开关管所需最小死区时间是开关频率fs和移相角θ的函数。通过上述改进时域分析方法,计算出不同开关频率fs和移相角θ的情况下开关管所需最小死区时间。进一步的,使用如下拟合函数对该函数进行拟合:When the output resistance does not change, the minimum dead time required by the switch is a function of the switching frequency f s and the phase shift angle θ. Through the above improved time domain analysis method, the minimum dead time required by the switch tube under different switching frequency f s and phase shift angle θ is calculated. Further, use the following fitting function to fit the function:

Figure BDA0003762114320000101
Figure BDA0003762114320000101

其中,C0-C5为待拟合参数。Among them, C 0 -C 5 are parameters to be fitted.

进一步的,对于负载电阻变化的情况,可以取多个可能的负载电阻值的进行计算,得到不同的拟合函数。电路运行时,先检测出实际的负载电阻值,使用线性插值法估算得到开关管所需最小死区时间。Further, for the change of the load resistance, a plurality of possible load resistance values can be taken for calculation to obtain different fitting functions. When the circuit is running, first detect the actual load resistance value, and use the linear interpolation method to estimate the minimum dead time required for the switch.

基于上述改进时域分析方法和桥臂最佳死区时间计算和参数拟合方法,可以得到混合控制LLC串联谐振变换器桥臂变死区控制方法。其原理如图6所示,包括:Based on the improved time-domain analysis method and the optimal dead-time calculation and parameter fitting method of the bridge arm, the variable dead-time control method of the bridge arm of the hybrid control LLC series resonant converter can be obtained. The principle is shown in Figure 6, including:

步骤1:根据硬件参数及指标,离线计算出不同工况下的最优死区时间。Step 1: According to the hardware parameters and indicators, calculate the optimal dead time under different working conditions offline.

步骤2:使用上述拟合函数对死区时间函数进行拟合,并将计算好的拟合函数写入嵌入式控制器中。Step 2: Use the above fitting function to fit the dead time function, and write the calculated fitting function into the embedded controller.

步骤3:电路初始运行时,实时采集输出电压,输出电流,正常进行电压电流闭环控制。此时电路死区时间较大,防止原边开关管失去零电压开通。Step 3: During the initial operation of the circuit, the output voltage and output current are collected in real time, and the voltage and current closed-loop control is normally performed. At this time, the dead time of the circuit is relatively large to prevent the primary side switch from losing zero voltage turn-on.

步骤4:待电路基本稳定运行后,从闭环控制模块中获得开关频率,移相角等参数,并结合输出电压,输出电流等参数,带入拟合函数并分别计算出超前桥臂与滞后桥臂所需的死区时间。Step 4: After the circuit is basically running stably, parameters such as switching frequency and phase shift angle are obtained from the closed-loop control module, and combined with parameters such as output voltage and output current, they are brought into the fitting function to calculate the leading bridge arm and the lagging bridge respectively. Dead time required for the arm.

步骤5:将步骤4计算出的死区时间数据增加以合适的裕度后传输至PWM发生器,以输出含有目标死区时间的驱动信号,进而减小开关管反向导通损耗。Step 5: Increase the dead time data calculated in step 4 with an appropriate margin and transmit it to the PWM generator to output a drive signal with the target dead time, thereby reducing the reverse conduction loss of the switch.

步骤6:当死区时间计算器检测到电路输出电压,输出电流,开关频率、移相角等参数发生较大变化时。再次重复步骤4和步骤5,以保证原边开关管恰好能实现零电压开通。Step 6: When the dead time calculator detects that the circuit output voltage, output current, switching frequency, phase shift angle and other parameters have changed greatly. Repeat steps 4 and 5 again to ensure that the primary side switch tube can just achieve zero-voltage turn-on.

因此,死区时间计算器仅需在检测到电路工作条件发生较大变化时才重新计算,这进一步的减小了嵌入式控制器的计算量。Therefore, the dead-time calculator only needs to be recalculated when a large change in circuit operating conditions is detected, which further reduces the computational load of the embedded controller.

应用已有的死区控制方法和本发明的变死区控制新方法的PSIM仿真电路波形图分别如图7的(a)、(b)所示。变换器工作在开关频率为700kHz,移相角为100°时,为使得两个桥臂开关管均实现零电压开通,已有方法中相位超前与滞后桥臂开关管死区时间一致,均为20ns。同等条件下,使用本发明变死区控制新方法后,超前桥臂开关管死区时间减小为约3ns,滞后桥臂开关管死区时间减小为约17ns,且两桥臂开关管均恰好实现零电压开通。本方法保证了在不同电路运行情况下,开关管能均实现零电压开通,并极大的减小了开关管反向导通时间,减小了开关过程中的损耗。The waveform diagrams of the PSIM simulation circuits applying the existing dead zone control method and the new variable dead zone control method of the present invention are respectively shown in (a) and (b) of FIG. 7 . When the converter operates at a switching frequency of 700 kHz and a phase shift angle of 100°, in order to achieve zero-voltage turn-on of the two bridge arm switches, the dead time of the phase lead and lag bridge arm switches in the existing method is consistent, which is both. 20ns. Under the same conditions, after using the new variable dead zone control method of the present invention, the dead zone time of the leading bridge arm switch tube is reduced to about 3 ns, the dead zone time of the lagging bridge arm switch tube is reduced to about 17 ns, and the two bridge arm switches are both. Just to achieve zero voltage turn-on. The method ensures that the switch tubes can be turned on at zero voltage under different circuit operation conditions, greatly reduces the reverse conduction time of the switch tubes, and reduces the loss in the switching process.

以上所述,仅是本发明的较佳实施例而已,并非是对本发明作其它形式的限制,任何熟悉本专业的技术人员可能利用上述揭示的技术内容加以变更或改型为等同变化的等效实施例。但是凡是未脱离本发明技术方案内容,依据本发明的技术实质对以上实施例所作的任何简单修改、等同变化与改型,仍属于本发明技术方案的保护范围。The above are only preferred embodiments of the present invention, and are not intended to limit the present invention in other forms. Any person skilled in the art may use the technical content disclosed above to make changes or modifications to equivalent changes. Example. However, any simple modifications, equivalent changes and modifications made to the above embodiments according to the technical essence of the present invention without departing from the content of the technical solutions of the present invention still belong to the protection scope of the technical solutions of the present invention.

本专利不局限于上述最佳实施方式,任何人在本专利的启示下都可以得出其它各种形式的混合控制LLC串联谐振变换器变死区控制方法,凡依本发明申请专利范围所做的均等变化与修饰,皆应属本专利的涵盖范围。This patent is not limited to the above-mentioned best embodiment, anyone can come up with other various forms of hybrid control LLC series resonant converter variable dead zone control methods under the inspiration of this patent. The equivalent changes and modifications of the above shall fall within the scope of this patent.

Claims (6)

1.一种混合控制LLC串联谐振变换器变死区控制方法,其特征在于,包括以下步骤:1. a hybrid control LLC series resonant converter becomes dead zone control method, is characterized in that, comprises the following steps: 步骤S1:获取电路硬件参数以及输入输出参数范围,包括:谐振电感Lr,谐振电容Cr,变压器Tx的原边励磁电感Lm,变压器匝比n,原边开关管输出电容Coss,输入电压Vi变化范围,输出电压Vo变化范围,以及输出电流Io变化范围;Step S1: Obtain circuit hardware parameters and the range of input and output parameters, including: resonant inductance L r , resonant capacitor Cr , primary excitation inductance L m of transformer T x , transformer turns ratio n, primary switch output capacitor C oss , Input voltage V i variation range, output voltage V o variation range, and output current I o variation range; 步骤S2:使用改进的时域分析方法分别计算不同输入输出参数情况下,混合控制LLC谐振变换器驱动信号相位超前和滞后的桥臂开关管的关断时刻的电流ioffStep S2: using the improved time-domain analysis method to calculate the current i off at the turn-off moment of the bridge arm switch tube of which the phase of the drive signal of the LLC resonant converter is mixed and controlled under different input and output parameters; 所述改进时域分析方法包括:列写出半个开关周期内,不同时段谐振电流iLr、励磁电流iLm与谐振电容电压vCr所满足的方程,并结合稳态时全桥LLC谐振变换器谐振波形的对称性,求解出任意时刻的谐振电流iLr、励磁电流iLm与谐振电容电压vCrThe improved time-domain analysis method includes: writing out the equations satisfied by the resonant current i Lr , the excitation current i Lm and the resonant capacitor voltage v Cr in different time periods within half a switching cycle, and combining the full-bridge LLC resonant transformation in steady state According to the symmetry of the resonant waveform of the resonator, the resonant current i Lr , the excitation current i Lm and the resonant capacitor voltage v Cr at any time are solved; 步骤S3:根据开关管寄生电容Ceq以及关断电流ioff,计算得到不同输入输出参数情况下原边开关管实现零电压开通所需的最短死区时间tdead_minStep S3: According to the parasitic capacitance C eq of the switch tube and the turn-off current i off , calculate and obtain the shortest dead time t dead_min required for the primary side switch tube to realize zero-voltage turn-on under different input and output parameters; 其中,最短死区时间计算方法为
Figure FDA0003762114310000011
Coss为开关管寄生输出电容;
Among them, the calculation method of the shortest dead time is as follows
Figure FDA0003762114310000011
C oss is the parasitic output capacitance of the switch;
步骤S4:使用简单的拟合函数模型f(fs,θ)拟合最短死区时间tdead_min与开关频率fs和移相角θ的函数关系,求解出拟合函数模型中的待定参数;Step S4: use a simple fitting function model f(f s , θ) to fit the functional relationship between the shortest dead time t dead_min and the switching frequency f s and the phase shift angle θ, and solve the undetermined parameters in the fitting function model; 步骤S5:将求解好的拟合函数模型f(fs,θ)写入嵌入式控制器,并在电路运行时实时计算,得到此工况下原边各个的开关管实现零电压开通所需最短死区时间;Step S5: Write the solved fitting function model f(f s , θ) into the embedded controller, and calculate it in real time when the circuit is running to obtain the requirements for zero-voltage turn-on of each switch on the primary side under this working condition Minimum dead time; 步骤S6:给步骤S5计算获得的最短死区时间增加合适的裕度后,将最终死区时间信号传输至PWM发生器,并通过驱动电路控制原边开关管的死区时间。Step S6: After adding an appropriate margin to the shortest dead time calculated in step S5, the final dead time signal is transmitted to the PWM generator, and the dead time of the primary switch is controlled by the drive circuit.
2.根据权利要求1所述的混合控制LLC串联谐振变换器变死区控制方法,其特征在于:所述改进时域分析方法中,在励磁电感Lm,谐振电感Lr与谐振电容Cr共同谐振的时间段tsp内,将谐振电流视作线性变化,且线性变化的速率为tsp端点时刻谐振电流变化速率的平均值。2. The hybrid control LLC series resonant converter variable dead zone control method according to claim 1, characterized in that: in the improved time domain analysis method, in the excitation inductance L m , the resonant inductance L r and the resonant capacitor C r During the common resonance time period tsp , the resonant current is regarded as a linear change, and the rate of linear change is the average value of the rate of change of the resonant current at the end point of tsp . 3.根据权利要求1或2所述的混合控制LLC串联谐振变换器变死区控制方法,其特征在于:所述改进时域分析方法具体包括:3. The hybrid control LLC series resonant converter variable dead zone control method according to claim 1 or 2, wherein the improved time domain analysis method specifically comprises: 步骤S21:获得所控制的LLC串联谐振变换器运行时的具体软硬件参数,包括:Step S21: Obtain the specific software and hardware parameters of the controlled LLC series resonant converter during operation, including: 谐振电感Lr,励磁电感Lm,谐振电容Cr,变压器变比n,开关频率fs,移相角θ,输入电压Vi,输出电压Vo,输出电流Io;其中,Vi、Vo、Io至少应获得到其中之二;Resonant inductance L r , excitation inductance L m , resonant capacitor C r , transformer ratio n, switching frequency f s , phase shift angle θ, input voltage V i , output voltage V o , output current I o ; among them, V i , At least two of V o and I o should be obtained; 步骤S22:计算励磁电流与谐振电容电压初值条件:Step S22: Calculate the initial value conditions of the excitation current and the resonant capacitor voltage:
Figure FDA0003762114310000021
Figure FDA0003762114310000021
其中,iLr指谐振电感电流瞬时值;vCr指谐振电容电压瞬时值;t0~2和t2~3分别指t0至t2的时间和t2至t3的时间,t0为一个开关周期初始时刻,t2为前半个开关周期内副边电流断续时刻,为未知数,t3为开关周期的中间时刻;k23指谐振电感电流在t2至t3时间段内变化的速率;Among them, i Lr refers to the instantaneous value of the resonant inductor current; v Cr refers to the instantaneous value of the resonant capacitor voltage; t 0 ~ 2 and t 2 ~ 3 refer to the time from t 0 to t 2 and the time from t 2 to t 3 respectively, and t 0 is The initial moment of a switching cycle, t 2 is the discontinuous moment of the secondary side current in the first half of the switching cycle, which is unknown, and t 3 is the middle moment of the switching cycle; k 23 refers to the change of the resonant inductor current during the time period from t 2 to t 3 . rate; 求解方程组(1)以获得励磁电流初值iLr(t0)和谐振电容电压初值vCr(t0);Solve the equation set (1) to obtain the initial value of the excitation current i Lr (t 0 ) and the initial value of the resonance capacitor voltage v Cr (t 0 ); 步骤S23:根据电路运行情况,列出iLr和vCr在t0至t3时间段内满足的微分方程,结合式(1)中的初值条件,求解得到iLr和vCr的时域表达式,相应微分方程为:Step S23: According to the operation of the circuit, list the differential equations that i Lr and v Cr satisfy in the time period from t 0 to t 3 , and combine the initial value conditions in equation (1) to solve the time domain to obtain i Lr and v Cr expression, the corresponding differential equation is:
Figure FDA0003762114310000022
Figure FDA0003762114310000022
步骤S24:根据iLr和uCr的满足的边界条件,带入式(3)中求得的时域表达式中,消除其中的未知数,得到最终的表达式,相应边界条件为:Step S24: According to the satisfied boundary conditions of i Lr and u Cr , bring it into the time domain expression obtained in formula (3), eliminate the unknowns therein, and obtain the final expression, and the corresponding boundary conditions are:
Figure FDA0003762114310000023
Figure FDA0003762114310000023
步骤S25:使用数值解法求出不同工况下两桥臂关断电流,包括:超前桥臂关断电流iLr(t1),滞后桥臂关断电流iLr(t3)。Step S25 : Use the numerical solution to obtain the off currents of the two bridge arms under different working conditions, including: the lead off current i Lr (t 1 ) and the lagging off current i Lr (t 3 ).
4.根据权利要求1所述的混合控制LLC串联谐振变换器变死区控制方法,其特征在于:在步骤S4中使用的简单拟合函数模型为:4. hybrid control LLC series resonant converter variable dead zone control method according to claim 1 is characterized in that: the simple fitting function model used in step S4 is:
Figure FDA0003762114310000024
Figure FDA0003762114310000024
其中,fs指开关频率;θ指移相角,大于或等于0,C0-C5为待拟合参数;所述移相角θ;对不同负载电阻下的死区时间进行拟合,将死区时间简化为开关频率与移相角的简单函数。Among them, f s refers to the switching frequency; θ refers to the phase shift angle, which is greater than or equal to 0, and C 0 -C 5 are the parameters to be fitted; the phase shift angle θ; the dead time under different load resistances is fitted, Simplifies dead time as a simple function of switching frequency and phase shift angle.
5.根据权利要求1所述的混合控制LLC串联谐振变换器变死区控制方法,其特征在于:在步骤S5中计算获得的驱动信号相位滞后的桥臂开关管实现零电压开通所需最短死区时间大于或等于相位超前桥臂开关管实现零电压开通所需最短死区时间。5. hybrid control LLC series resonant converter variable dead zone control method according to claim 1, is characterized in that: in step S5, the bridge arm switch tube of the phase lag of the driving signal obtained by calculation realizes the shortest dead time required for zero voltage turn-on. The zone time is greater than or equal to the shortest dead zone time required for the phase-leading bridge arm switch to achieve zero-voltage turn-on. 6.根据权利要求1所述的混合控制LLC串联谐振变换器变死区控制方法,其特征在于:采用电压电流采样电路连接LLC串联谐振变换器的负载以采集输出电压Vo和输出电流Io的信息;采用驱动电路连接LLC串联谐振变换器的四个开关管以提供驱动信号;所述嵌入式控制器连接电压电流采样电路和驱动电路。6. hybrid control LLC series resonant converter variable dead zone control method according to claim 1, is characterized in that: adopt voltage current sampling circuit to connect the load of LLC series resonant converter to collect output voltage V o and output current I o The four switching tubes of the LLC series resonant converter are connected by a driving circuit to provide driving signals; the embedded controller is connected with the voltage and current sampling circuit and the driving circuit.
CN202210875991.2A 2022-07-25 2022-07-25 Hybrid control LLC series resonant converter variable dead time control method Pending CN115149817A (en)

Priority Applications (1)

Application Number Priority Date Filing Date Title
CN202210875991.2A CN115149817A (en) 2022-07-25 2022-07-25 Hybrid control LLC series resonant converter variable dead time control method

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
CN202210875991.2A CN115149817A (en) 2022-07-25 2022-07-25 Hybrid control LLC series resonant converter variable dead time control method

Publications (1)

Publication Number Publication Date
CN115149817A true CN115149817A (en) 2022-10-04

Family

ID=83414843

Family Applications (1)

Application Number Title Priority Date Filing Date
CN202210875991.2A Pending CN115149817A (en) 2022-07-25 2022-07-25 Hybrid control LLC series resonant converter variable dead time control method

Country Status (1)

Country Link
CN (1) CN115149817A (en)

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN117728696A (en) * 2024-02-08 2024-03-19 浙江艾罗网络能源技术股份有限公司 Controller, dual active bridge converter and control method thereof

Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN109004827A (en) * 2018-07-20 2018-12-14 东南大学 A kind of control system of the adaptive asymmetric dead time of cascade converter
CN112994472A (en) * 2021-04-12 2021-06-18 华中科技大学 Optimal dead zone calculation method and variable dead zone control method for high-frequency LLC resonant converter
WO2022088744A1 (en) * 2020-10-28 2022-05-05 广州金升阳科技有限公司 Llc resonant converter, and wide gain control method

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN109004827A (en) * 2018-07-20 2018-12-14 东南大学 A kind of control system of the adaptive asymmetric dead time of cascade converter
WO2022088744A1 (en) * 2020-10-28 2022-05-05 广州金升阳科技有限公司 Llc resonant converter, and wide gain control method
CN112994472A (en) * 2021-04-12 2021-06-18 华中科技大学 Optimal dead zone calculation method and variable dead zone control method for high-frequency LLC resonant converter

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN117728696A (en) * 2024-02-08 2024-03-19 浙江艾罗网络能源技术股份有限公司 Controller, dual active bridge converter and control method thereof
CN117728696B (en) * 2024-02-08 2024-05-28 浙江艾罗网络能源技术股份有限公司 Controller, dual active bridge converter and control method thereof

Similar Documents

Publication Publication Date Title
WO2021077757A1 (en) Wide gain control method for variable topology llc resonant converter
CN110943606B (en) Control method based on double-active-bridge rectifier no-current sampling power factor correction
CN110365205B (en) High-efficiency totem-pole bridgeless PFC rectifier control method
CN111490683A (en) Trajectory control method for double-transformer series resonance double-active bridge DC-DC converter topology
CN105226929B (en) One kind becomes mode cascade converter
CN103457475B (en) A kind of fuzzy control method to high-voltage capacitor charging and device
CN107968571A (en) A kind of double active three phase-shifting control methods of bridging parallel operation
CN109067190B (en) LLC resonant converter of wide gain
CN112436728B (en) High-efficiency control method of bidirectional resonant converter
CN113489342B (en) Dual Phase Shift Control Method of Dual Active Bridge Converter Based on Transformer Inductance
CN108494256A (en) A kind of the LLC resonant converter underloading voltage modulation system and modulation strategy of energy feedback
CN105515366B (en) A kind of mixing control method for LCC resonance DC DC converters
CN111478572A (en) Mode Smooth Switching and Power Factor Correction Control Method for Unipolar AC-DC Converter
CN112311222A (en) An improved bridgeless DBPFC converter and control method based on composite predictive current control
CN117175968A (en) Single-stage CLLC bidirectional converter and control method thereof
CN109194135A (en) A kind of adaptive efficiency optimization method of resonant state adjustable type power inverter
CN110518818B (en) CRM (customer relationship management) buck-flyback PFC (Power factor correction) converter controlled in fixed frequency
CN115149817A (en) Hybrid control LLC series resonant converter variable dead time control method
CN209375466U (en) A Wide Gain LLC Resonant Converter
CN115242095A (en) An isolated CLLC converter bidirectional synchronous rectification control device and method
CN213846539U (en) A High Frequency Intermittent Control System for Bidirectional Series Resonant Converter
CN210867516U (en) LLC converter switching over wide voltage range based on alternating current switch
CN110995011B (en) A bidirectional DC-DC converter based on AC switch switching
CN118074492A (en) Soft switching control method for variable frequency of bidirectional DC-DC power supply
CN115473441B (en) Optimized control strategy for isolated three-port soft switching converter

Legal Events

Date Code Title Description
PB01 Publication
PB01 Publication
SE01 Entry into force of request for substantive examination
SE01 Entry into force of request for substantive examination
TA01 Transfer of patent application right
TA01 Transfer of patent application right

Effective date of registration: 20230327

Address after: No. 406, Anbian Road, Min'an Street, Xiang'an District, Xiamen City, Fujian Province, 361000

Applicant after: Xiamen Lijing New Energy Technology Co.,Ltd.

Address before: Fuzhou University, No.2, wulongjiang North Avenue, Fuzhou University Town, Minhou County, Fuzhou City, Fujian Province

Applicant before: FUZHOU University