CN112311222A - Improved bridgeless DBPFC converter based on composite prediction current control and control method - Google Patents

Improved bridgeless DBPFC converter based on composite prediction current control and control method Download PDF

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CN112311222A
CN112311222A CN202010980074.1A CN202010980074A CN112311222A CN 112311222 A CN112311222 A CN 112311222A CN 202010980074 A CN202010980074 A CN 202010980074A CN 112311222 A CN112311222 A CN 112311222A
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diode
current
inductor
time
mode
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郑宏
郭其金
沈建杨
袁雪凯
严序文
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Jiangsu University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H02M1/4258—Arrangements for improving power factor of AC input using a single converter stage both for correction of AC input power factor and generation of a regulated and galvanically isolated DC output voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/14—Arrangements for reducing ripples from dc input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/44—Circuits or arrangements for compensating for electromagnetic interference in converters or inverters
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Abstract

The invention discloses an improved bridgeless DBPFC converter based on composite prediction current control and a control method thereof, wherein the circuit reduces the switching loss of the converter by introducing a controllable auxiliary circuit, the loss generated by the auxiliary circuit is smaller, and a passive inductor is added at the output end of the converter, so that the output voltage is more stable. The invention has the following advantages and characteristics: 1. the improved circuit reduces the switching loss of the converter, reduces the output voltage ripple by 2, and combines prediction current control to enable the input current to well follow the input voltage, thereby obtaining higher power factor and smaller total harmonic distortion.

Description

Improved bridgeless DBPFC converter based on composite prediction current control and control method
Technical Field
The invention belongs to the technical field of power electronic application, and particularly designs an improved bridgeless DBPFC converter based on composite prediction current control and a control method.
Background
With the rapid development of power electronic technology, the national sustainable development strategy puts forward a definite requirement on the greening of power supplies, on one hand, power is saved, and on the other hand, the harmonic pollution of a power grid is greatly reduced, so that the power supply quality and the reliability of the power grid are ensured. In order to meet the requirements of related standards, such as IEC61000-3-2, on Power Factor and harmonic, more and more Power electronic devices are provided with a Power Factor Correction (PFC) link, and the PFC technology has the advantages of high Power Factor, stable output voltage, small input harmonic current, and the like, and is therefore widely applied to solving the harmonic pollution of the Power grid. The Boost PFC converter is relatively simple in control strategy, and the inductor exists on the side close to a power grid, so that the input current is easy to be continuous, and the Boost PFC converter is widely applied to the field of improving the quality of electric energy.
Compared with the defects of more switching devices, large switching loss and the like of the traditional Boost PFC converter, the bridgeless Dual _ Boost _ PFC converter (DBPFC) has the advantages that a rectifier bridge is omitted, a set of semiconductor devices and capacitors are added, and at any moment, the number of power devices conducted on a current circulation path is less, so that the switching loss of a circuit is reduced. However, the Dual-Boost PFC converter still has the problems of reverse recovery loss of the Boost diode and switching loss of the switching tube. For the problem of switching loss, it is desirable to realize Zero Voltage (ZVS) switching, Zero Current (ZCS) switching, or soft switching of the switching tube, and the ripple on the output side also meets the industrial requirements as much as possible. In the aspect of a control strategy, the traditional average current control parameter selection is complex, the operation amount is large, the gain of a current amplifier is limited at the switching frequency, and the dynamic response performance is poor. The peak current control is greatly affected by noise, and the THD value is large because an error exists between the average value and the current peak value. The frequency of the current hysteresis loop control is not fixed, and the controllability of the switching frequency is poor. Based on the above analysis, how to reduce the circuit loss problem of the DBPFC converter and achieve good control of the converter becomes the focus of the invention herein.
Disclosure of Invention
The invention aims to provide an improved bridgeless DBPFC converter based on composite prediction current control and a control method thereof, wherein the problem of loss of the converter is solved through a controllable auxiliary circuit, output ripples are reduced through adding a passive inductor on an output side, and the improvement of a circuit structure is realized.
In order to achieve the purpose, the invention adopts the technical scheme that: an improved bridgeless DBPFC converter based on composite prediction current control comprises a basic bridgeless DBPFC circuit, a controllable auxiliary circuit, a passive inductor, a second-order generalized integrator SOGI + frequency-locked loop FLL control module and a prediction current control module; the basic bridgeless DBPFC circuit consists of a traditional bridgeless Boost-PFC circuit and a slow recovery diode Da、DbWherein the conventional bridgeless Boost-PFC circuit is composed of an AC input source VinInput side boost inductor L1And L2Diode D1And D2Switch tube S1And S2And an output side capacitor C0And a load R0Composition of the slow recovery diode DaAnode connection D ofbAnd is connected to ground, a slow recovery diode DaThe cathode of the diode is connected with the anode of an alternating current input source, and a slow recovery diode DbThe cathode of the power supply is connected with the negative electrode of the alternating current input source; the AC input source is also respectively connected with the input side inductor L1、L2One end is connected; inductor L1The other end is connected to a diode D1Anode and switching tube S1Between the sources, an inductance L2The other end is connected to a diode D2Anode and switching tube S2Between the sources, diode D1And D2Cathode and capacitor C0Passive inductor L3One end connected to passive inductor L3The other end is connected with a load R0One terminal, capacitor C0And a load R0The other end of the switch tube S is uniformly connected with the switch tube1And S2The drain electrode is connected with the ground;
the controllable auxiliary circuit is a resonant capacitor CrResonant inductor LrSwitch tube S3Capacitor C1、C2、C3Diode D3、D4、D5、D6、D7Composition of, the diode D3Anode and input side inductor L1Connected to the diode D4Is connected to the input side inductor L2, the resonant capacitor CrAnd diode D3The resonant inductance LrAnd diode D4Is connected to the cathode of a diode D5And D7Is connected to LrOne end, D7Cathode and switching tube S3Is connected to the collector electrode of D5And D6Is connected to CrOne end, D6The cathode of the capacitor is connected with a load end and a capacitor C1、C2、C3Are respectively connected in parallel to the switch tube S1、S2、S3Two ends;
the control ends of the prediction current control module are respectively connected with a switch tube S1、S2PWM1 and PWM2 signal driving regulation are carried out, and meanwhile, the control end of the prediction current control module is connected with a switch tube S through a time delay module3PWM3 signal drive adjustment is carried out, a second-order generalized integrator SOGI + frequency locking ring FLL control module is composed of a second-order generalized integrator SOGI and a frequency locking ring FLL, the second-order generalized integrator SOGI can track sine signals with specific frequency without static error, alternating current input source voltage is collected through a voltage sensor and used for extracting power grid voltage fundamental waves, and the frequency locking ring FLL is used for extracting power grid frequency.
Further, the input of the frequency locking loop FLL is determined by the initial given angular frequency and the second-order generalized integral link, and the output is good grid frequency.
Further, the second order generalized integrator has a transfer function of
Figure BDA0002687205190000021
kiIs the gain factor, ωrIs the resonant angular frequency.
Furthermore, the prediction current control module analyzes the inductive current in a CCM mode, and deduces a relational expression of the inductive current and the input voltage in the nth switching period to obtain a corresponding control equation of the inductive current and the input voltage
Figure BDA0002687205190000022
Wherein d (n) characterizes the control duty cycle, TsCharacterizing a time parameter, iL(n) characterizing the inductor current, VinCharacterizing input voltage, iref(n +1) characterisation of the reference current, VrefThe output reference voltage is characterized.
The technical scheme of the method is as follows: an improved bridgeless DBPFC converter control method based on composite prediction current control comprises the following steps:
working mode 1: before mode 1, the main switching tube S1Auxiliary switch tube S3All are turned off, the controllable auxiliary circuit does not work and flows through L1Through a boost diode D1Into the load at t0Starting mode 1 at the moment, and then, for the switching tube S3Applying a drive signal due to the resonant inductance LrPresence of resonant inductor current iLrIncrease linearly from zero, S3Soft on, in this mode, diode D1Current iD1Linear decrease, when it decreases to 0, boost diode D1Naturally shutting off;
and (3) working mode 2: at t1Time of day Modal 2, at which time iD1Reduced to zero, diode D1Naturally shut off, and S1Not yet turned on, diode D1Voltage VD1Is still zero, D1The reverse recovery loss is close to zero, in this mode, LrAnd S1Parallel capacitor C1Resonance occurs, C1Starting discharging, the switch tube S1Voltage VS1Decrease of LrCharging iLrContinuing to increase;
working mode 3: at t2Time of day Modal 3, at which time iLrTo the maximum value of the current of the switching period, VS1Reduced to zero, S1Parasitic diode DS1Opening and maintaining S1Reverse current i ofS1In this mode, LrTerminal voltage V ofLr＝0，iLrFollow current, and maintain iLr＝ILr，ILrIs LrCurrent steady state value, at this time S1Reach ZVS turn-on (zero voltage turn-on) condition;
the working mode 4 is as follows: at t3Time of day onset modality 4, this time for S1Applying a drive signal while S3Removing the drive signal due to the resonant capacitor CrPresence of LrAnd CrStart of series resonance, LrStarting discharge, iLrFrom ILrStart to decrease, CrCharging, CrVoltage VCrIncreasing from zero, diode D5Naturally on due to diode D7Presence of, capacitance C3Cannot be connected with the inductor LrThe electric energy transmission is carried out, the unnecessary resonance is effectively restrained, and in the mode, S1ZVS turn-on is realized, and turn-on loss is zero; s3Soft switching is realized, the turn-off loss approaches to zero, D3Naturally shutting off;
working mode 5: at t4Time of day Modal 5, at which time LrAnd CrEnd of resonance, iLr＝0，D5Naturally turning off, wherein in the mode, the controllable auxiliary circuit does not work, and the ZVT bridgeless Boost PFC main circuit continues to work;
the working mode 6 is as follows: at t5Time of day Start Modal 6, this time for S1Removing the driving signal, due to C1Limit VS1Rate of rise of S1Is a soft turn-off, the turn-off loss is much less than a hard turn-off, I in this modeL1To C1Charging, VS1Increase from zero;
the working mode 7 is as follows: at t6Time of day starting modality 7, V at this timeS1+VCr＝V0Wherein V is0To output electricityPressure, in this mode, C1Continue charging while CrStarting discharge into the load, D3、D6Naturally opening;
the working mode 8 is as follows: at t7Time start mode 8, end of charge, VS1＝V0While C isrEnd of discharge, VCr＝0，D3、D6Naturally turn-off, boost diode D1Naturally on and the converter goes back to the next switching cycle.
Further, the passive inductor L3The filter capacitor is added to the output end of the load under the condition of not changing the filter capacitor, the filter capacitor has the characteristic of direct current resistance and alternating current, and the current flowing through the inductor cannot change suddenly.
The SOGI + FLL and prediction current algorithm combined composite control strategy is embodied in a reference current i in the algorithmrefAnd (n +1) extracting and tracking a power grid fundamental wave signal by the SOGI + FLL, compounding the power grid fundamental wave signal with an output generated by voltage loop control to obtain a corresponding reference current value, and processing the reference current value by a prediction current control algorithm to obtain a final PWM (pulse width modulation) switch control signal so as to realize the control of the circuit.
The controllable auxiliary circuit realizes the switch tube S1、S2Zero voltage turn-on (ZVS) and soft turn-off of the switch tube S are realized3The soft turn-off greatly reduces the switching loss of the circuit.
According to the SOGI + FLL and prediction current algorithm combined composite control strategy, the SOGI does not need phase signals of a power grid, only needs power grid angular frequency, the frequency-locked loop is simple in structure, a trigonometric function does not need to be calculated, calculation time can be saved, and extraction of the power grid frequency is achieved. The predictive current control obtains excellent tracking capability in a mode of directly controlling duty ratio, and after the predictive current control is combined with an SOGI + FLL link, the obtained composite control strategy can realize good following of input current to input voltage, so that a high power factor value (PF), small Total Harmonic Distortion (THD) and good dynamic response capability are obtained.
The passive inductor has the characteristic of direct current resistance and alternating current resistance, the influence of high-frequency common mode interference can be further reduced, and the current flowing through the inductor cannot change suddenly, so that the voltage of the load on the output side can be more stable.
Compared with the traditional control methods such as average current control and the like, the prediction current control has better tracking performance and faster response speed, but the superior tracking capability is obtained by directly controlling the duty ratio, so that the inductive current is extremely easily influenced by the voltage distortion of the power grid, and the influence of the voltage distortion of the power grid on the power factor correction is reduced by adopting the method of SOGI + FLL.
The passive inductor L3 has the characteristic of direct current resistance and alternating current resistance, the influence of high-frequency common mode interference can be further reduced, and the current flowing through the inductor cannot change suddenly, so that the voltage of the load on the output side can be more stable.
Drawings
FIG. 1 is a circuit topology structure diagram of an improved bridgeless DBPFC converter based on composite prediction current control according to the invention;
fig. 2 is an equivalent circuit of each switch working mode of the improved bridgeless DBPFC converter based on composite prediction current control in the positive half period according to the present invention. (a) Is mode 1; (b) is mode 2; (c) is modality 3; (d) is modality 4; (e) is modality 5; (f) is modality 6; (g) is modality 7; (h) is modal 8;
FIG. 3 is a SOGI + FLL control diagram of an improved bridgeless DBPFC converter based on composite prediction current control according to the invention;
FIG. 4 is a general control diagram of an improved bridgeless DBPFC converter based on composite predictive current control according to the present invention;
FIG. 5 is a waveform diagram of the soft switching of the auxiliary switch tube of the improved bridgeless DBPFC converter based on the composite predicted current control according to the present invention
FIG. 6 is a graph of input current-following input voltage waveforms for an improved bridgeless DBPFC converter based on composite predicted current control according to the present invention;
FIG. 7 is a graph of the output voltage waveform of an improved bridgeless DBPFC converter based on composite predicted current control according to the present invention;
fig. 8 is a complete diagram of the circuit topology.
Detailed Description
The following detailed description of embodiments of the invention refers to the accompanying drawings in which:
as shown in fig. 1 and 8, a circuit topology structure diagram of an improved bridgeless DBPFC converter based on composite prediction current control includes a basic bridgeless Boost-PFC circuit and a slow recovery diode Da、DbAnd a controllable auxiliary circuit.
As shown in fig. 2, an improved bridgeless DBPFC converter based on composite predicted current control includes 8 equivalent circuits of switching operation modes in a positive half cycle, and the specific steps are as follows:
working mode 1[ t ]0～t1]：
Before mode 1, the main switching tube S1Auxiliary switch tube S3All are turned off, the controllable auxiliary circuit does not work, and the inductive current passes through the boost diode D1Into the load. At t0Time of day Modal 1, this time for S3Applying a drive signal due to the resonant inductance LrPresence of (i)LrIncrease linearly from zero, S3Soft switching on. In this mode, iD1Linear decrease, when it decreases to 0, boost diode D1And naturally shutting down.
Working mode 2[ t ]1～t2]：
As shown in fig. 2(b), at t1Time of day Modal 2, at which time iD1Reduced to zero, D1And naturally shutting down. And S1Has not been opened yet, VD1Is still zero, D1The reverse recovery loss is nearly zero. In this mode, LrAnd S1Parallel capacitor C1Onset of resonance, C1Starting discharge, VS1Decrease of LrCharging iLrThe increase continues.
Working mode 3[ t ]2～t3]：
As shown in fig. 2(c), at t2Time of day Modal 3, at which time iLrTo the maximum value of the current of the switching period, VS1Reduced to zero, S1Parasitic diode DS1Opening and maintaining S1Reverse current i ofS1. In this mode, LrTerminal voltage V ofLr＝0，iLrFollow current, and maintain iLr＝ILr，S1The ZVS turn-on condition is reached.
Working mode 4[ t ]3～t4]：
At t3Time of day onset modality 4, this time for S1Applying a drive signal while S3The drive signal is removed. Due to resonant capacitance CrPresence of LrAnd CrStart of series resonance, LrStarting discharge, iLrFrom ILrStart to decrease, CrCharging, VCrIncrease from zero, D5Naturally open the circuit. Due to D7Presence of, capacitance C3Cannot be connected with the inductor LrAnd the electric energy is transmitted, and unnecessary resonance is effectively inhibited. In this mode, S1ZVS turn-on is realized, and turn-on loss is zero; s3Soft switching is realized, the turn-off loss approaches to zero, D3And naturally shutting down.
Working mode 5[ t ]4～t5]：
As shown in fig. 2(e), at t4Time of day Modal 5, at which time LrAnd CrEnd of resonance, iLr＝0，D5And naturally shutting down. In this mode, the controllable auxiliary circuit does not work, and the ZVT bridgeless Boost PFC main circuit continues to work.
Working mode 6[ t ]5～t6]：
At t, as shown in FIG. 2(f)5Time of day Start Modal 6, this time for S1Removing the driving signal, due to C1Limit VS1Rate of rise of S1Is a soft turn-off, the turn-off loss is much less than a hard turn-off. In this mode, IL1To C1Charging, VS1Starting from zero.
Working mode 7[ t ]6～t7]：
As shown in FIG. 2(g), at t6Time of day starting modality 7, V at this timeS1+VCr＝V0Wherein V is0For output voltage, in this mode, C1Continue charging while CrStarting discharge into the load, D3、D6Naturally open the circuit.
Working mode 8[ t ]7～t8]：
As shown in FIG. 2(h), at t7Time start mode 8, end of charge, VS1＝V0. At the same time CrEnd of discharge, VCr＝0，D3、D6Naturally turn-off, boost diode D1Naturally on and the converter goes back to the next switching cycle.
When the passive inductor added on the output side takes values, the values of the filter capacitor and the load need to be considered comprehensively, resonance of the inductor and the capacitor is avoided, and meanwhile, the influence on the load is reduced to the maximum extent.
As shown in fig. 3, the outline of the SOGI + FLL control diagram of an improved bridgeless DBPFC converter based on composite predicted current control is represented by a SOGI structure diagram, an FLL structure diagram, and a transfer function of a second-order generalized integrator
Figure BDA0002687205190000061
kiIs the gain factor, ωrFor the resonance angular frequency, the sinusoidal signal V passes through the SOGI to generate two orthogonal signals, whose transfer functions are:
Figure BDA0002687205190000062
can be mixed with Hα(s) is used as a second-order band-pass filter, and can accurately track the frequency omeganOf the signal of (1). The performance such as bandwidth, response speed and the like of the SOGI are adjusted by controlling the k value, and the alternating current signal is well controlled. Omega1For a given initial angular frequency, the value is 100 pi for an alternating current of 50 HZ. Gamma is a coefficient, the negative sign of the former indicates that the negative feedback is negative, and the parameters of the negative feedback can be obtained by gradually debugging in practical application. It can be seen from the block diagram that the SOGI does not need a phase signal of the grid, but only needs a grid angular frequency. And from the structure of the frequency-locked loopThe structure is simple, a trigonometric function does not need to be calculated, the calculation time can be saved, and the extraction of the power grid frequency is realized.
As shown in fig. 4, an overall control diagram of an improved bridgeless DBPFC converter based on composite predictive current control includes a predictive current control strategy and the aforementioned SOGI + FLL control module. Derivation premise of predictive current control: the power switch tube and the recovery diode in the circuit are set as ideal elements; the inductor current works in a CCM mode; the output dc voltage is regarded as a certain value. In the nth switching period, the starting time is t (n), the ending time is t (n +1), the duty ratio is d (n), and the relation between the inductive current and the input voltage in the nth switching period is as follows: when the switch tube is conducted, the switch tube is connected with the switch,
Figure BDA0002687205190000063
(t(n)≤t≤t(n)+d(n).Ts) (ii) a When the switch tube is turned off, the switch tube is turned on,
Figure BDA0002687205190000064
according to the differential definition, and considering that the frequency of the input voltage is negligible with respect to the switching tube operating frequency, the above relation can be expressed as:
Figure BDA0002687205190000065
wherein iL(t(n))、iL(t (n +1)) is the value of the inductor current at the beginning and end of the nth switching cycle. By further derivation
Figure BDA0002687205190000066
Discretizing the obtained product to obtain
Figure BDA0002687205190000067
At steady state, the output voltage V0Is equal to the output voltage reference voltage VrefSo that I is at the end of the nth switching cyclemaxFollowing reference iref(n +1) may be represented by iref(n +1) instead of iL(n +1) is obtained through corresponding transformation
Figure BDA0002687205190000068
By applying a current i to the inductorL(n) input voltage VinReference current iref(n +1), output reference voltage VrefThe control can realize predictive current control. Although the duty ratio d (n) is directly controlled without the need of current loop PI regulation, the control state quantity depends on the stability of the grid voltage, and in order to reduce the influence of grid voltage distortion on the control, the grid frequency is extracted and tracked by using a mode of combining a second-order generalized integrator (SOGI) and a frequency-locked loop (FLL). Reference current i embodied in an algorithmrefAnd (n +1) is obtained by extracting a power grid fundamental wave signal by the SOGI + FLL and then compounding the power grid fundamental wave signal with an output generated by voltage loop control.
An improved bridgeless DBPFC converter based on composite prediction current control is simulated by utilizing PSIM software, wherein input voltage Vin220V, and outputting a reference voltage V0Rated output power P of 400V0600W, switching frequency fs100kHz, output capacitance C01120 μ F boost inductor L1/L20.5mH, output side resistance R0180 Ω, and 50 HZ. The simulation result is shown in FIG. 5, when the auxiliary switch tube starts to conduct, due to the resonant inductance LrThe current rising rate is limited, the current rises slowly, and soft switching is realized.
As shown in fig. 6, a graph of an input current-following input voltage waveform of an improved bridgeless DBPFC converter based on composite prediction current control is obvious, and it is obvious that an input side current can well follow a voltage to fluctuate in a sine shape, so that a circuit can obtain a high power factor and a small total harmonic distortion;
as shown in fig. 7, a graph of an output voltage waveform of an improved bridgeless DBPFC converter based on composite predictive current control is shown, and it is obvious from the graph that the output voltage ripple of the converter is smaller, and the converter has no overshoot and has better output performance.
In summary, the improved bridgeless DBPFC converter based on composite prediction current control and the control method thereof reduce the switching loss of the converter by introducing a controllable auxiliary circuit, the loss generated by the auxiliary circuit is smaller, the output end of the converter is added with a passive inductor, so that the output voltage is more stable, and meanwhile, a composite prediction current control strategy is invented to realize the control of the converter, and the strategy adopts a second-order generalized integrator (SOGI) and a Frequency Locking Loop (FLL) to be combined to extract the fundamental wave of a power grid, and is combined with a prediction current control algorithm to inhibit the influence of the voltage distortion of the power grid on the inductive current. The invention has the following advantages and characteristics: 1. the improved circuit reduces the switching loss of the converter, reduces the output voltage ripple by 2, and combines prediction current control to enable the input current to well follow the input voltage, thereby obtaining higher power factor and smaller total harmonic distortion.
In the description herein, references to the description of the term "one embodiment," "some embodiments," "an illustrative embodiment," "an example," "a specific example," or "some examples" or the like mean that a particular feature, structure, material, or characteristic described in connection with the embodiment or example is included in at least one embodiment or example of the invention. In this specification, the schematic representations of the terms used above do not necessarily refer to the same embodiment or example. Furthermore, the particular features, structures, materials, or characteristics described may be combined in any suitable manner in any one or more embodiments or examples.
While embodiments of the invention have been shown and described, it will be understood by those of ordinary skill in the art that: various changes, modifications, substitutions and alterations can be made to the embodiments without departing from the principles and spirit of the invention, the scope of which is defined by the claims and their equivalents.

Claims (6)

1. An improved bridgeless DBPFC converter based on composite prediction current control is characterized by comprising a basic bridgeless DBPFC circuit, a controllable auxiliary circuit, a passive inductor, a second-order generalized integrator SOGI + frequency locking loop FLL control module and a prediction current control module;
the basic bridgeless DBPFC circuit consists of a traditional bridgeless Boost-PFC circuit and a slow recovery diode Da、DbWherein the conventional bridgeless Boost-PFC circuit is composed of an AC input source VinInput side boost inductor L1And L2Diode D1And D2Switch tube S1And S2And an output side capacitor C0And a load R0Composition of the slow recovery diode DaAnode connection D ofbAnd is connected to ground, a slow recovery diode DaThe cathode of the diode is connected with the anode of an alternating current input source, and a slow recovery diode DbThe cathode of the power supply is connected with the negative electrode of the alternating current input source; the AC input source is also respectively connected with the input side inductor L1、L2One end is connected; inductor L1The other end is connected to a diode D1Anode and switching tube S1Between the sources, an inductance L2The other end is connected to a diode D2Anode and switching tube S2Between the sources, diode D1And D2Cathode and capacitor C0Passive inductor L3One end connected to passive inductor L3The other end is connected with a load R0One terminal, capacitor C0And a load R0The other end of the switch tube S is uniformly connected with the switch tube1And S2The drain electrode is connected with the ground;
the controllable auxiliary circuit is a resonant capacitor CrResonant inductor LrSwitch tube S3Capacitor C1、C2、C3Diode D3、D4、D5、D6、D7Composition of, the diode D3Anode and input side inductor L1Connected to the diode D4Is connected to the input side inductor L2, the resonant capacitor CrAnd diode D3The resonant inductance LrAnd diode D4Is connected to the cathode of a diode D5And D7Is connected to LrOne end, D7Cathode and switching tube S3Is connected to the collector electrode of D5And D6Is connected to CrOne end, D6The cathode of the capacitor is connected with a load end and a capacitor C1、C2、C3Are respectively connected in parallel to the switch tube S1、S2、S3Two ends;
the control ends of the prediction current control module are respectively connected with a switch tube S1、S2PWM1 and PWM2 signal driving regulation are carried out, and meanwhile, the control end of the prediction current control module is connected with a switch tube S through a time delay module3PWM3 signal drive adjustment is carried out, a second-order generalized integrator SOGI + frequency locking ring FLL control module is composed of a second-order generalized integrator SOGI and a frequency locking ring FLL, the second-order generalized integrator SOGI collects alternating current input source voltage through a voltage sensor and is used for extracting power grid voltage fundamental waves, and the frequency locking ring FLL is used for extracting power grid frequency.
2. The improved bridgeless DBPFC converter based on the composite prediction current control as claimed in claim 1, wherein an input of the frequency-locked loop FLL is determined by an initial given angular frequency and a second-order generalized integral element, and an output is a good grid frequency.
3. The improved bridgeless DBPFC converter based on composite predicted current control as claimed in claim 1, wherein a transfer function of the second-order generalized integrator is
Figure FDA0002687205180000011
kiIs the gain factor, ωrIs the resonant angular frequency.
4. The improved bridgeless DBPFC converter based on composite predictive current control of claim 1, wherein the predictive current control module analyzes an inductor current in a CCM mode, and derives a control equation corresponding to the inductor current and an input voltage relation from an n-th switching period
Figure FDA0002687205180000012
Wherein d (n) characterizes the control duty cycle, TsCharacterizing a time parameter, iL(n) characterizing the inductor current, VinCharacterizing input voltage, iref(n +1) characterisation of the reference current, VrefThe output reference voltage is characterized.
5. The method as claimed in claim 1, wherein the method comprises the following steps:
working mode 1: before mode 1, the main switching tube S1Auxiliary switch tube S3All are turned off, the controllable auxiliary circuit does not work and flows through L1Through a boost diode D1Into the load at t0Starting mode 1 at the moment, and then, for the switching tube S3Applying a drive signal due to the resonant inductance LrPresence of resonant inductor current iLrIncrease linearly from zero, S3Soft on, in this mode, diode D1Current iD1Linear decrease, when it decreases to 0, boost diode D1Naturally shutting off;
and (3) working mode 2: at t1Time of day Modal 2, at which time iD1Reduced to zero, diode D1Naturally shut off, and S1Not yet turned on, diode D1Voltage VD1Is still zero, D1The reverse recovery loss is close to zero, in this mode, LrAnd S1Parallel capacitor C1Resonance occurs, C1Starting discharging, the switch tube S1Voltage VS1Decrease of LrCharging iLrContinuing to increase;
working mode 3: at t2Time of day Modal 3, at which time iLrTo the maximum value of the current of the switching period, VS1Reduced to zero, S1Parasitic diode DS1Opening and maintaining S1Reverse current i ofS1In this mode, LrTerminal voltage V ofLr＝0，iLrFollow current, and maintain iLr＝ILr，ILrIs LrCurrent steady state value, at this time S1Reach ZVS turn-on (zero voltage turn-on) condition;
the working mode 4 is as follows: at t3Time of day onset modality 4, this time for S1Applying a drive signal while S3Removing the drive signal due to the resonant capacitor CrPresence of LrAnd CrStart of series resonance, LrStarting discharge, iLrFrom ILrStart to decrease, CrCharging, CrVoltage VCrIncreasing from zero, diode D5Naturally on due to diode D7Presence of, capacitance C3Cannot be connected with the inductor LrThe electric energy transmission is carried out, the unnecessary resonance is effectively restrained, and in the mode, S1ZVS turn-on is realized, and turn-on loss is zero; s3Soft switching is realized, the turn-off loss approaches to zero, D3Naturally shutting off;
working mode 5: at t4Time of day Modal 5, at which time LrAnd CrEnd of resonance, iLr＝0，D5Naturally turning off, wherein in the mode, the controllable auxiliary circuit does not work, and the ZVT bridgeless Boost PFC main circuit continues to work;
the working mode 6 is as follows: at t5Time of day Start Modal 6, this time for S1Removing the driving signal, due to C1Limit VS1Rate of rise of S1Is a soft turn-off, the turn-off loss is much less than a hard turn-off, I in this modeL1To C1Charging, VS1Increase from zero;
the working mode 7 is as follows: at t6Time of day starting modality 7, V at this timeS1+VCr＝V0Wherein V is0For output voltage, in this mode, C1Continue charging while CrStarting discharge into the load, D3、D6Naturally opening;
the working mode 8 is as follows: at t7Time start mode 8, end of charge, VS1＝V0While C isrEnd of discharge, VCr＝0，D3、D6Naturally turn-off, boost diode D1Naturally open, the converter returns to the nextA switching cycle process.
6. The method as claimed in claim 5, wherein the passive inductor L is the passive inductor L3The filter capacitor is added to the output end of the load under the condition of not changing the filter capacitor, the filter capacitor has the characteristic of direct current resistance and alternating current, and the current flowing through the inductor cannot change suddenly.
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