CN110611444B - Bridgeless integrated AC-DC (alternating current-direct current) rectifying circuit and rectifying method - Google Patents

Bridgeless integrated AC-DC (alternating current-direct current) rectifying circuit and rectifying method Download PDF

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CN110611444B
CN110611444B CN201910871624.3A CN201910871624A CN110611444B CN 110611444 B CN110611444 B CN 110611444B CN 201910871624 A CN201910871624 A CN 201910871624A CN 110611444 B CN110611444 B CN 110611444B
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circuit
capacitor
stage
switch tube
bridge
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CN110611444A (en
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潘尚智
王明龙
宫金武
查晓明
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Wuhan University WHU
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Wuhan University WHU
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/14—Arrangements for reducing ripples from dc input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/219—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only in a bridge configuration
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

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  • Power Engineering (AREA)
  • Rectifiers (AREA)

Abstract

The invention provides a bridgeless integrated AC-DC rectification circuit and a rectification method, comprising the following steps: single-phase ac power, first inductor, etc. The post-stage DC-DC circuit comprises but is not limited to a half-bridge LLC resonant circuit, a full-bridge LLC resonant circuit, a double-active full-bridge conversion circuit, a double-active half-bridge conversion circuit and the like. The invention possesses two degrees of freedom of control, including: pre-stage control: the control object is a first bridge arm (low frequency), and the duty ratio of a switch tube of the bridge arm is adjusted in real time by a sine pulse width modulation method so as to realize the function of power factor correction and the adjustment of voltage on the direct current side output by the PFC; and (3) post-stage control: the control object is a second bridge arm (high frequency), and the output voltage of the rear-stage DC-DC circuit is adjusted in real time through frequency conversion control or phase-shift control; the control method enables the front and rear stages to be controlled separately. Meanwhile, the bridge arm can be shared with a post-stage DC-DC conversion circuit, so that the aim of improving the efficiency is fulfilled.

Description

Bridgeless integrated AC-DC (alternating current-direct current) rectifying circuit and rectifying method
Technical Field
The invention relates to the technical field of power electronic conversion, in particular to a bridgeless integrated AC-DC rectifying circuit and a method.
Background
At present, a large amount of bridge type uncontrolled rectification is used, so that not only is serious harmonic pollution caused to a power grid, but also waste of electric energy is caused by low power factor of an alternating current side. The power factor correction technology can realize that the alternating current side current tracks the alternating current side voltage, and can improve the power factor of the alternating current side.
Power factor correction techniques must be able to achieve high power factor and low input current harmonics to meet IEC 61000-3-2 harmonic standards. Therefore, a conventional power supply generally includes a two-stage power conversion structure, a Boost circuit used as a Power Factor Correction (PFC) circuit for realizing a high power factor and a low input current harmonic, and a DC-DC conversion circuit for outputting a required DC voltage. The two-stage circuit topology can enable the circuit to achieve optimal performance, such as high power factor, stable PFC output DC side voltage, and stable DC-DC output voltage. However, the two-stage structure has too many components, so that the power consumption is high, the efficiency is relatively low, the circuit control is complex, and most of the system loss is consumed in the rectifier diode.
Aiming at the problem of efficiency reduction caused by excessive number of components, two solutions are mainly provided, one is to improve the topological structure of the first-stage power factor correction circuit to form a bridgeless PFC structure so as to reduce the number of diodes and switching tubes as much as possible. At present, various bridgeless PFC circuits are available, such as a double-Boost PFC circuit, a double-inductor PFC circuit, a totem-pole PFC circuit and the like; the other method is to integrate the first-stage PFC circuit and the second-stage DC-DC circuit by sharing one bridge arm.
Some students combine the two solutions together, and propose a topology structure in which the totem-pole PFC and the post-stage DC-DC conversion circuit are integrated by sharing one bridge arm, although the number of switching devices can be further reduced, the circuit topology only has one degree of control freedom in this way, so that the voltage on the output direct current side of the PFC is uncontrollable, and when the input voltage is increased to a certain degree, the voltage stress on the output direct current side of the PFC and the voltage stress on the switching devices are too high, so that the devices are easily damaged; in addition, because the voltage of the output direct current side of the PFC is uncontrollable, the voltage change is too large, and the parameter design of a post-stage DC-DC circuit is poor.
In order to solve the problem that the output dc voltage of the PFC is not controllable, some researchers have proposed a solution to achieve the purpose of controlling the output dc voltage of the PFC by combining pulse frequency modulation and pulse width modulation. When the voltage of the direct current side output by the PFC does not exceed a specified limit value, pulse frequency modulation is adopted; when the voltage of the output direct current side of the PFC approaches or exceeds a specified limit value, the purpose of reducing the voltage of the output direct current side of the PFC is achieved by changing the duty ratio in a mode of combining pulse frequency modulation and pulse width modulation, however, input harmonic current is increased, the power factor is reduced, and the duty ratio is limited by input voltage change, so that the working condition of a rear-stage DC-DC circuit is deteriorated, the design cannot be optimized, and the efficiency of the whole circuit is influenced to a certain extent.
Disclosure of Invention
The present invention is directed to overcoming the above-mentioned problems, and provides 1. a bridgeless integrated AC-DC rectifier circuit and method, comprising: the single-phase alternating current power supply, a first inductor, a first switching tube, a second switching tube, a third switching tube, a fourth switching tube and a rear-stage DC-DC circuit;
the latter stage DC-DC circuit adopts half-bridge LLC resonant circuit, includes: the single-phase alternating-current power supply comprises a single-phase alternating-current power supply, a first inductor, a first switching tube, a second switching tube, a third switching tube, a fourth switching tube, a dc capacitor, a first capacitor, a second capacitor, a first transformer, a first diode, a second diode, a third capacitor and a first load resistor;
one end of the single-phase alternating current power supply is connected with one end of the first inductor; the other end of the single-phase alternating current power supply is connected with a drain electrode of the fourth switching tube; the other end of the first inductor and the first switch tube S 1 Is connected with the source electrode of the transistor; the drain electrode of the first switching tube, the drain electrode of the third switching tube and the dc capacitor are connected with the anode of the first capacitor; the source electrode of the first switch tube is connected with the drain electrode of the second switch tube; the source electrode of the third switching tube and the drain electrode of the second switching tube are connected with one end of the primary side of the first transformer; the negative electrode of the first capacitor and the positive electrode of the second capacitor are connected with the other end of the primary side of the first transformer; the source electrode of the second switch tube, the source electrode of the fourth switch tube and the dc capacitor are connected with the negative electrode of the second capacitor; one end of the secondary side of the first transformer is connected with the anode of the first diode; the middle end of the first transformer is connected with the cathode of the third capacitor and one end of the first resistor; the third end of the secondary side of the first transformer is connected with the anode of a second diode; and the anodes of the first diode, the second diode and the third capacitor are connected with the other end of the first resistor.
2. The bridgeless integrated AC-DC rectifying circuit according to claim 1, wherein the post-stage DC-DC circuit is a half-bridge LLC resonant circuit, a full-bridge LLC resonant circuit, a dual-active full-bridge conversion circuit, or a dual-active half-bridge conversion circuit.
The single-phase alternating current power supply is used for providing input alternating current power supply; the first inductor is used for storing and releasing energy; the first switching tube is used for controlling the output of direct-current voltage; the second switching tube is used for controlling the output of direct-current voltage; the third switching tube is used for controlling the output of direct-current voltage; the fourth switching tube is used for controlling the output of direct-current voltage; the dc capacitor is used for filtering output direct-current voltage ripples; the first capacitor and the second capacitor are used for generating resonance with the inductor; the first load resistor is used for providing direct current voltage output; the first transformer is used for transmitting energy to an output side; the first diode is used for providing a current circulation path; the second diode is used for providing a current circulation path; and the third capacitor is used for filtering the direct-current voltage ripple at the output side.
A method of rectification using a bridgeless integrated AC-DC rectification circuit, comprising:
preceding stage control: the control object is a first bridge arm, the first bridge arm is low-frequency, and the duty ratio of a switch tube of the bridge arm is adjusted in real time by a sine pulse width modulation method so as to realize the function of power factor correction and the adjustment of voltage on a PFC output direct current side;
and (3) post-stage control: the control object is a second bridge arm, the second bridge arm is high-frequency, and the output voltage of the rear-stage DC-DC circuit is adjusted in real time through frequency conversion control or phase-shifting control;
the control method enables the front stage and the rear stage to be controlled respectively; meanwhile, the bridge arm can be shared with a rear-stage DC-DC conversion circuit, so that the aim of improving the efficiency is fulfilled;
the expression of the voltage on the DC side of the PFC output is as follows:
Figure GDA0003250529130000031
wherein, V dc Represents the output voltage of the PFC direct current side; v i Represents an ac side input voltage; d 1 Bridge arm switch tube S with low frequency representation 2 Duty cycle of (d); d 2 Indicating high frequency bridge arm switch tube S 3 The duty cycle of (c).
Compared with the prior art, the invention has the following advantages:
the invention has two degrees of freedom of control. The first bridge arm is a low-frequency bridge arm, and the duty ratio of a switch tube of the bridge arm is adjusted in real time through a sine pulse width modulation method so as to realize the function of power factor correction and the adjustment of the voltage on the output direct current side of the PFC; the second bridge arm is a high-frequency bridge arm, the duty ratio is fixed, the second bridge arm can share the bridge arm with a half-bridge LLC resonant DC-DC circuit, integration is realized, a soft switching effect is brought, and the purpose of improving efficiency is achieved.
Drawings
Fig. 1 is a circuit diagram of the system of the present invention.
Fig. 2 is a typical circuit diagram of a half-bridge LLC resonant DC-DC circuit followed by the system of the present invention.
FIG. 3: the driving signals of the switching tubes S3 and S4, input inductive current, resonant current, excitation inductive current and current waveforms flowing through the first diode and the second diode are in a stable working state of a typical circuit of the system.
Fig. 4 is an equivalent circuit diagram of a typical circuit of the system of the present invention during mode 1 of operation.
Fig. 5 is an equivalent circuit diagram of an exemplary circuit of the system of the present invention during mode 2 of operation.
Fig. 6 is an equivalent circuit diagram of an exemplary circuit of the system of the present invention during mode 3 of operation.
Fig. 7 is an equivalent circuit diagram of an exemplary circuit of the system of the present invention during mode 4 of operation.
Fig. 8 is an equivalent circuit diagram of an exemplary circuit of the system of the present invention during mode 5 of operation.
Fig. 9 is an equivalent circuit diagram of an exemplary circuit of the system of the present invention during operating mode 6.
Detailed Description
In order to facilitate the understanding and implementation of the present invention for those of ordinary skill in the art, the present invention is further described in detail with reference to the accompanying drawings and examples, it is to be understood that the embodiments described herein are merely illustrative and explanatory of the present invention and are not restrictive thereof.
The circuit of this embodiment is shown in figure 1,
single phase ac power supply v in First inductance L 1 A first switch tube S 1 A second switch tube S 2 Third switch tube S 3 Fourth switch tube S 4 Dc capacitor C dc First capacitor C 1 First capacitor C 2 First transformer T 1 First diode D 1 A second diode D 2 Third capacitor C 3 And a first load resistor R 1 ；
The single-phase AC power supply v in One end of (1) and the first inductor L 1 Is connected with one end of the connecting rod; the single-phase AC power supply v in And the other end of the second switch tube S 4 Is connected with the drain electrode of the transistor; the first inductor L 1 And the other end of the first switch tube S 1 Is connected with the source electrode of the transistor; the first switch tube S 1 Drain electrode of (2) and the third switching tube S 3 Drain of (b), positive electrode of said dc capacitor and said first capacitor C 1 The positive electrode of (1) is connected; the first switch tube S 1 Source electrode of and the second switch tube S 2 Is connected with the drain electrode of the transistor; the third switch tube S 3 Source electrode of, the fourth switching tube S 4 And the first transformer T 1 One end of the primary side is connected; the first capacitor C 1 Negative pole of (1), the second capacitor C 2 And the first transformer T 1 The other end of the primary side is connected; the second switch tube S 2 Source electrode of, the fourth switching tube S 4 Source of the dc capacitor, a cathode of the dc capacitor and the second capacitor C 2 Is connected with the negative pole of the anode; the first transformer T 1 One end of the secondary side and the first diode D 1 The anode of (2) is connected; the first transformer T 1 An intermediate terminal and the third capacitor C 3 Negative electrode of (1), first resistor R 1 Is connected with one end of the connecting rod; the first transformer T 1 The third end of the secondary side and a second diode D 2 The anode of (2) is connected; the first diode D 1 The second diode D 2 The third capacitor C 3 And the first resistor R 1 The other end of the connecting rod is connected.
The following describes the embodiments of the present invention with reference to fig. 2 to 9:
setting a first inductance L 1 Has a current of i in At a voltage v L First capacitor C 1 Has a voltage of v C1 A second capacitor C 2 Has a voltage of v C2 Third capacitor C 3 Has a voltage of v C3 The output voltage of the PFC DC side is v dc ，v dc ＝v C1 +v C2 Flows through the first diode D 1 Has a current of i D1 Flows through the first diode D 2 Has a current of i D2 Flows through the first transformer T 1 Current of leakage inductance is i Lr Flows through the first transformer T 1 Current of exciting inductor is i m First load resistance R 1 At a voltage v o 。
Due to S 1 ，S 2 The working processes of the upper pipe and the lower pipe are symmetrical, so that only S is analyzed 1 6 working stages when the upper pipe is opened. FIGS. 4-9 are circuit diagrams S of the circuit shown in FIG. 1 1 A schematic diagram of an equivalent circuit of a working mode when the upper tube is turned on, wherein FIG. 4 is a first switch tube S 1 A third switch tube S 3 Conducting the second switch tube S 2 And a fourth switching tube S 4 Off, the first diode D 1 Forward conducting, second diode D 2 An equivalent circuit schematic diagram when reverse turn-off is performed; FIG. 5 shows the first switch tube S 1 A third switch tube S 3 Conducting the second switch tube S 2 The fourth switch tube S 4 Off, the first diode D 1 A second diode D 2 An equivalent circuit schematic diagram when reverse turn-off is performed; FIG. 6 shows the first switch tube S 1 Conducting the second switch tube S 2 A third switch tube S 3 Turn-off and fourth switch tube S 4 Off, the first diode D 1 A second diode D 2 An equivalent circuit schematic diagram during reverse turn-off; FIG. 7 shows the first switch tube S 1 And a fourth switching tube S 4 Conducting the second switch tube S 2 A third switch tube S 3 Turn-off, first diode D 1 Reverse turn-off, second diode D 2 An equivalent circuit schematic diagram when conducting in the forward direction; FIG. 8 shows the first switch tube S 1 A second switch tube S 2 The third switch tubeS 3 Turn-off, fourth switch tube S 4 Conducting the first diode D 1 A second diode D 2 An equivalent circuit schematic diagram when reverse turn-off is performed; FIG. 9 shows the first switch tube S 1 A second switch tube S 2 A third switch tube S 3 And a fourth switching tube S 4 Off, the first diode D 1 A second diode D 2 And an equivalent circuit schematic diagram in reverse turn-off.
The working conditions of the modes are specifically analyzed, and the adjustment of the output voltage at the direct current side can be realized by adjusting the duty ratio of the low-frequency bridge arm, so that the first inductor L 1 Average current i in a single switching cycle of Lave Is constant and i will not be listed again in each modal analysis in And v C1 、v C2 、v C3 Using i in the steady state analysis Lave Instead of the first inductor L 1 Current value i of in . It is to be noted that the following processes or parameters, if any, which are not specifically described in detail are understood or implemented by those skilled in the art with reference to the prior art.
As shown in FIG. 4, modality 1 corresponds to t of FIG. 3 0 ～t 1 Time period:
at t ═ t 0 While, the first switch tube S 1 A third switch tube S 3 Zero voltage on, second switch tube S 2 The fourth switch tube S 4 Off, input current i in Passes through the first inductor L once 1 A first switch tube S 1 A third switch tube S 3 Then back to the single-phase AC supply v in . At this time, the inductor energy is increased and the inductor current i in Linearly rising, inductor current i in Can be expressed as:
Figure GDA0003250529130000061
at the same time, the first capacitor C 1 A third switch tube S 3 First transformer leakage inductance L r A first transformer resonant inductor L m A second capacitor C 2 Forming a resonant circuit, resonant current i Lr Specific exciting inductance current i m Large so that the current flowing through the primary side of the transformer is positive and negative, and the amplitude is the difference between the resonant current and the exciting current, therefore, the first diode D on the secondary side of the transformer 1 And conducting. The primary voltage of the transformer is clamped at nV o Exciting inductance current is linearly increased, exciting inductance current i Lr Can be expressed as:
Figure GDA0003250529130000062
at this stage, the resonant frequency is:
Figure GDA0003250529130000063
at t 1 And at the moment, when the resonant inductor current is equal to the excitation inductor current, ending the mode 1 working mode.
As shown in FIG. 5, modality 2 corresponds to t of FIG. 3 1 ～t 2 Time period:
at t ═ t 1 While, the first switch tube S 1 A third switch tube S 3 Continuously turning on, inputting an inductive current i in The linear increase continues until t is reached 2 The time of day reaches a maximum value. In addition, at t 1 Time of day, resonant inductor current i Lr Equal to exciting inductance current i m When the first diode is turned off, the first diode is turned off with zero current. In this mode, the first capacitor C 1 A third switch tube S 3 First transformer leakage inductance L r A first transformer resonant inductor L m A second capacitor C 2 Forming a resonant circuit, wherein the resonant frequency is as follows:
Figure GDA0003250529130000064
in this mode, since the resonant current is equal to the exciting inductor current, the first and second diodes on the secondary side of the transformer are turned off in reverse directions, so that no energy is transferred to the secondary side. At t ═ t 2 While, the third switch tube S 3 Off, modality 2 ends.
As shown in FIG. 6, isModality 3, corresponding to t of FIG. 3 2 ～t 3 Time period:
in this mode, corresponding to the dead time of the switching signal, the third switch tube S is now in the process 3 The fourth switch tube S 4 Turn off, input of inductor current i in Linear decrease, the relevant electrical parameter relationship in this mode is:
Figure GDA0003250529130000065
due to the third switch tube S 3 And a fourth switching tube S 4 The presence of parasitic capacitances, to which the excitation current is discharged, thus realizes the fourth switching tube S 4 The zero voltage of (2) turns on. When t is equal to t 3 While, the fourth switch tube S 4 And (4) opening.
As shown in FIG. 7, modality 4, corresponding to t of FIG. 3 3 ～t 4 Time period:
at t ═ t 3 While, the fourth switch tube S 4 The drain-source voltage will be zero, therefore, the fourth switch tube S 4 Zero voltage is on. In this mode, the inductor current i is input in Continuing to linearly decrease; because the fourth switch tube S 4 Conducting so that the input voltage of the resonant tank is zero. In this mode, the resonance current is larger than the exciting inductance current, and the secondary side of the transformer is provided with a second diode D according to the polarity relation of the transformer 2 The forward direction is conducted, the primary voltage of the transformer is clamped at-nV o Excitation inductance current i m The linear decrease, the magnetizing inductor current can be expressed as:
Figure GDA0003250529130000071
at t ═ t 4 When the resonance current is equal to the excitation inductance current, the mode 4 ends.
As shown in FIG. 8, modality 5 corresponds to t of FIG. 3 4 ～t 5 Time period:
fourth switch tube S 4 Keeping on state, when t is t 4 While the resonant current is equal to the exciting inductor current, and a second diode D 2 Zero current off, at t-t 5 While, the fourth switch tube S 4 The switch-off, the mode 5 end,
as shown in FIG. 9, modality 6, corresponding to t of FIG. 3 5 ～t 6 Time period:
in this mode, all the primary side switching tubes and the secondary side diodes are in the off state, which is the same as mode 3, because the third switching tube S 3 The fourth switch tube S 4 Due to the existence of the parasitic capacitors, the excitation current discharges to the parasitic capacitors, and zero-voltage switching-on of the third switching tube is realized. When t is equal to t 6 While, the third switch tube S 3 And (4) opening.
The 6 working stages when the second switch tube is switched on are similar to the above, and are not described again.
According to the analysis of the above modes, the first inductance L is 1 And analyzing the relation among all parameters of the system under the steady state condition by using a volt-second balance principle, wherein variables represented by capital characters below are steady state values of corresponding variables. A first switch tube S is arranged 1 The time of opening is (1-D) 1 )T s1 A second switch tube S 2 Time of opening is D 1 T s1 A third switching tube S 3 Time of opening is D 2 T s2 Fourth switch tube S 4 The time of opening is (1-D) 2 )T s2 Wherein D is 1 Is a first switch tube S 1 Duty cycle of (1-D) 1 ) Is a second switch tube S 2 Duty ratio of D 2 For a third switching tube S 3 Duty cycle of (1-D) 4 ) Is a fourth switching tube S 4 The duty cycle of (c). T is s1 ，T s2 Is a switching period, and T s1 Greater than T s2 . For the first inductance L 1 The method is obtained by applying a volt-second balance principle:
Figure GDA0003250529130000072
finishing the formula (5) to obtain:
Figure GDA0003250529130000081
the above-mentioned embodiment is a typical circuit of the system of the present invention, but the implementation manner of the present invention is not limited by the above-mentioned embodiment, and a typical circuit formed by changing the DC-DC circuit (including but not limited to a half-bridge LLC resonant circuit, a full-bridge LLC resonant circuit, a dual-active full-bridge inverter circuit, a dual-active half-bridge inverter circuit, etc.) of the latter stage of the system of the present invention is included in the protection scope of the present invention.
The above-mentioned embodiments are preferred embodiments of the present invention, but the present invention is not limited to the above-mentioned embodiments, and any other changes, modifications, substitutions, combinations, and simplifications which do not depart from the spirit and principle of the present invention should be construed as equivalents thereof, and all such changes, modifications, substitutions, combinations, and simplifications are intended to be included in the scope of the present invention.
It should be understood that the above description of the preferred embodiments is given for clarity and not for any purpose of limitation, and that various changes, substitutions and alterations can be made herein without departing from the spirit and scope of the invention as defined by the appended claims.

Claims (1)

1. A bridgeless integrated AC-DC rectification circuit, comprising: the single-phase alternating current power supply, the first inductor, the first switching tube, the second switching tube and the rear-stage DC-DC circuit;
the latter stage DC-DC circuit adopts half-bridge LLC resonant circuit, includes: the direct current power supply comprises a dc capacitor, a first capacitor, a second capacitor, a first transformer, a first diode, a second diode, a third capacitor, a first load resistor, a third switch tube and a fourth switch tube;
one end of the single-phase alternating current power supply is connected with one end of the first inductor; the other end of the single-phase alternating current power supply is connected with a drain electrode of the fourth switching tube; the other end of the first inductor is connected with a source electrode of the first switching tube; the drain electrode of the first switch tube and one end of the dc capacitor are connected with the anode of the first capacitor; the source electrode of the first switch tube is connected with the drain electrode of the second switch tube; the source electrode of the third switching tube is connected with the drain electrode of a fourth switching tube, and the drain electrode of the fourth switching tube is connected with one end of the primary side of the first transformer; the negative electrode of the first capacitor and the positive electrode of the second capacitor are connected with the other end of the primary side of the first transformer; the source electrode of the second switch tube, the source electrode of the fourth switch tube and the other end of the dc capacitor are connected with the negative electrode of the second capacitor; one end of the secondary side of the first transformer is connected with the anode of the first diode; the middle end of the secondary side of the first transformer is connected with the negative electrode of the third capacitor and one end of the first resistor; the third end of the secondary side of the first transformer is connected with the anode of a second diode; the cathode of the first diode, the cathode of the second diode and the anode of the third capacitor are connected with the other end of the first resistor;
the rear-stage DC-DC circuit adopts a half-bridge LLC resonant circuit, or a full-bridge LLC resonant circuit, or a double-active full-bridge conversion circuit, or a double-active half-bridge conversion circuit;
the method for rectifying by adopting the bridgeless integrated AC-DC rectifying circuit comprises the following steps:
preceding stage control: the control object is a first bridge arm, the first bridge arm is low-frequency, and the duty ratio of a switch tube of the bridge arm is adjusted in real time by a sine pulse width modulation method so as to realize the function of power factor correction and the adjustment of voltage on a PFC output direct current side;
and (3) post-stage control: the control object is a second bridge arm, the second bridge arm is high-frequency, and the output voltage of the rear-stage DC-DC circuit is adjusted in real time through frequency conversion control or phase-shift control;
the control method enables the front stage and the rear stage to be controlled respectively; meanwhile, the bridge arm can be shared with a rear-stage DC-DC conversion circuit, so that the aim of improving the efficiency is fulfilled;
the expression of the voltage on the DC side of the PFC output is as follows:
Figure FDA0003739611490000011
wherein, V dc Represents the output voltage of the PFC direct current side; v i Represents an alternating-current-side input voltage; d 1 Denotes a second switching tube S 2 Duty cycle of (d); d 2 Denotes a third switching transistor S 3 The first bridge arm comprises a first switching tube and a second switching tube; the second bridge arm comprises a third switching tube and a fourth switching tube.
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