CN110518818B - CRM (customer relationship management) buck-flyback PFC (Power factor correction) converter controlled in fixed frequency - Google Patents

CRM (customer relationship management) buck-flyback PFC (Power factor correction) converter controlled in fixed frequency Download PDF

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CN110518818B
CN110518818B CN201910561980.5A CN201910561980A CN110518818B CN 110518818 B CN110518818 B CN 110518818B CN 201910561980 A CN201910561980 A CN 201910561980A CN 110518818 B CN110518818 B CN 110518818B
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CN110518818A (en
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马春伟
姚凯
张震
邬程健
管婵波
李凌格
陈杰楠
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Nanjing University of Science and Technology
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Nanjing University of Science and Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/14—Arrangements for reducing ripples from dc input or output
    • H02M1/15—Arrangements for reducing ripples from dc input or output using active elements
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • H02M3/33592—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer having a synchronous rectifier circuit or a synchronous freewheeling circuit at the secondary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/219—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only in a bridge configuration
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Abstract

The invention provides a CRM buck-flyback PFC converter adopting fixed frequency control, which comprises a main power circuit, an input voltage digital feedforward circuit, an output voltage difference distribution circuit, a feedback circuit, a state judgment circuit, a driving signal generation circuit, a comparison circuit and a multiplication circuit. The invention introduces an input voltage digital feedforward circuit, an input voltage comparison circuit and a gating circuit to realize a double-fixed-frequency control mode, so that the switching frequencies of the switching tubes of the converter in the Buck stage and the Flyback stage are respectively kept at different constant values.

Description

CRM (customer relationship management) buck-flyback PFC (Power factor correction) converter controlled in fixed frequency
Technical Field
The invention relates to an alternating current-direct current converter technology, in particular to a CRM buck-flyback PFC converter controlled in a fixed frequency mode.
Background
Due to the problems of an input current dead zone, a switching tube floating and the like, the traditional BuckPFC converter cannot meet the design technical requirements in many application occasions. In order to solve these problems, a Buck-Flyback PFC converter has been proposed, which combines a Buck topology and a Flyback topology, the Buck topology operating when an input voltage is greater than an output voltage, and the Flyback topology operating when the input voltage is less than the output voltage. However, the traditional CRM buck-flyback PFC converter adopts fixed on-time control, and when two topologies of the converter work, the on-time is equal, so that the switching tube switching frequency variation range in a half power frequency period in the control mode is large, the power factor is low in a low-voltage range, and the performance requirement is difficult to meet.
The conduction time of the switch tube of each switching period of the traditional CRM buck-flyback PFC converter is the same. The control is simple, the power factor in a high-voltage range is high, and the diode has no reverse recovery problem; the disadvantages are that the switching tube has a large switching frequency conversion range in a half power cycle, a low voltage range has a very low power factor, an EMI design is complex, and the efficiency is low.
Disclosure of Invention
The invention aims to provide a CRM buck-flyback PFC converter controlled in a fixed frequency mode.
The technical scheme for realizing the purpose of the invention is as follows: a CRM buck-flyback PFC converter controlled in a fixed frequency mode comprises a main power circuit and a control circuit, wherein the main power circuit comprises an input power supply vinEMI filter, rectifier diode D1-D6Primary side inductance L of main circuitpSecondary inductance L of main circuitsZero-crossing detection inductor LZCDAnd a switching tube QbAnd a switching tube QfFreewheel diode DoAn output capacitor CoLoad RL(ii) a Input voltage source vinConnected with the input port of the EMI filter, the output port of the EMI filter and the rectifier diode D1-D4Composed rectifier bridge and D3-D6The input ports of the formed rectifier bridges are connected, and a rectifier diode D3、D4The positive pole is a reference potential zero point and a rectifier diode D5、D6Primary side inductance L of negative and main circuitpConnected with a primary side inductor L of a main circuitpAnd the other end of the switch tube QfIs connected with a rectifier diode D1、D2Secondary side inductance L of cathode and main circuitsConnected to and connected to a freewheeling diode DoIs connected with the negative pole of the main circuit and the secondary side inductance L of the main circuitsAnd output capacitor CoIs connected with a load RL and outputs a capacitor CoTogether with the load RL, with a freewheeling diode DoQ connected to switching tubebOne end of (1), zero-cross detection inductance LZCDAnd main circuit inductances Lp and LsAll-coupled zero-crossing detection inductor LZCDOne end of the current limiting resistor is connected with a reference potential zero point and the other end is connected with a current limiting resistor RZCDConnecting, current sampling resistor RsOne end of the switch tube is grounded and the other end is connected with the switch tube QfAnd a switching tube QbAnd (4) connecting.
Furthermore, the control circuit comprises a main control chip, a sampling circuit, a digital-to-analog conversion circuit, an output voltage difference distribution circuit, a feedback circuit, a state judgment circuit, a driving signal generation circuit, a comparison circuit and a multiplication circuit; the positive input end of the output voltage differential circuit and the output voltage positive pole V of the main circuito+ connected, negative input end of output voltage differential circuit and positive electrode V of output voltage of main circuito-connected, the output of the output voltage divider circuit being connected to a first input of the sampling circuit and to an input K of the feedback circuit, a second input of the sampling circuit being connected to the rectified voltage v of the main circuitgThe first output end of the sampling circuit is connected with an ADC2 port of the main control chip, the second output end of the sampling circuit is connected with an ADC1 port of the main control chip, the first output port of the main control chip is connected with an input port of the digital-to-analog conversion circuit, and the output port of the digital-to-analog conversion circuit is connected with an input port v of the multiplierxConnected, output port of feedback circuit and input v of multiplieryThe output end of the multiplier is connected with the positive input end of the comparison circuit, and the negative input end of the comparison circuit is connected with the main circuit vRsThe output end of the comparison circuit is connected with the input end of the drive signal generation circuit, and the input end of the drive signal generation circuit passes through the current-limiting resistor RZCDZero-crossing detection winding L with main circuitZCDThe output end of the drive signal generating circuit is connected with the input end of the state judging circuit, and the end of the drive signal generating circuit is connected with the switching tube Q of the main power circuitbA switch tube Q connected with the end of the drive signal generating circuit and the main power circuitfAnd (4) connecting.
Further, the sampling circuit converts the input voltage v of the convertergAnd outputVoltage VoConverted into voltage v capable of being collected by control chipg_dspAnd Vo_dsp(ii) a The main control chip comprises a phase locking program, an input voltage peak value sampling program, an output voltage sampling program and a control program, vg_dspConnected with ADC1 interface of control chip, and providing the sampled value to phase-locked procedure and input voltage peak value sampling procedure, Vo_dspThe sampling device is connected with an ADC2 interface of the control chip, the sampled numerical value is sent to an output voltage sampling program, the output of a phase locking program, an input voltage peak value sampling program and an output voltage sampling program is sent to a main control program, and the main control program calculates the numerical value of the profile of the inductive current; the input end of the digital-to-analog conversion circuit is connected with the control chip through an SPI protocol, and the digital-to-analog conversion circuit converts the numerical value of the received inductive current contour into voltage to be output.
Further, the state determination circuit includes a first comparator Comp1 and a resistor R5、R6Reference voltage source VboundaryTwo AND gates, a NOT gate and a drive circuit; the equidirectional input terminal of the first comparator Comp1 passes through the voltage-dividing resistor R5、R6Voltage v rectified by a diode rectifier circuit of the main power circuit (1)gConnected to the inverting input of the first comparator Comp1boundaryThe output end of the first comparator Comp1 is connected with one input end of the first AND Gate AND 1 AND the input end of the NOT Gate, the output end of the NOT Gate is connected with one input end of the second AND Gate AND 2, the output ends of the two AND gates are connected with the input end of the driving circuit, AND the output ends of the driving circuit are respectively connected with two switching tubes Q of the main power circuitbAnd Qf。
Furthermore, the driving signal generating circuit may adopt an integrated IC circuit of a model such as L6561 or L6562, the amplifier used in the output voltage differential sampling circuit and the output voltage feedback circuit may adopt an operational amplifier of a model such as TL074, TL072, LM358, LM324, etc., the multiplier may be composed of an integrated IC circuit or a discrete device, the and gate used in the state judging circuit may adopt a logic chip of a model such as SN74HC08N, CD4011BE, or 74HC32N, the driving circuit may adopt a driving chip of a model such as IR2110, TLP2590, etc., or a totem-pole driving circuit, and the control chip may adopt a chip such as TMS320F28335 or TMS320F28377, etc.
Compared with the prior art, the invention has the following advantages: (1) when the constant switching frequency is adopted for control, the switching frequencies of the two topological switching tubes in a half power frequency period are kept to be constant values; (2) compared with the traditional constant on-time control, the constant switching frequency control is controlled in a low-voltage range, and the PF is obviously improved; (3) the effective value of the inductive current is greatly reduced in a constant switching frequency type control mode, so that the efficiency of the whole converter is greatly improved; (4) the output voltage ripple under the control of the constant switching frequency is greatly reduced compared with the traditional constant on-time control.
The invention is further described below with reference to the accompanying drawings.
Drawings
Fig. 1 is a schematic diagram of a CRM buck-flyback PFC converter main circuit.
Fig. 2 is a waveform diagram of the inductor current and the switching tube current of the CRM buck-flyback PFC converter in one switching period.
Fig. 3 is a waveform diagram of an input current under a conventional control.
Fig. 4 is a graph of switching frequency variation under conventional control.
Fig. 5 is a graph showing a PF change curve under conventional control.
FIG. 6 is a diagram showing the PF variation under the control of constant switching frequency at different turn ratios.
Fig. 7 is a diagram of the variation curve of the switching frequency under two controls.
FIG. 8 is a graph showing the PF change under two controls.
Fig. 9 is a schematic diagram of the input current harmonic curves under two controls.
Fig. 10 is a graph illustrating the effective values of the primary and secondary currents of the inductor under two types of control.
Fig. 11 is a graph showing the variation curve of the output voltage ripple under two kinds of control.
Fig. 12 is a combined schematic diagram of a main power circuit and a control circuit of a CRM buck-flyback PFC converter with fixed frequency control.
Main symbol names in the above figures: v. ofinSupply voltage iinInput current, RB-rectifier bridge, vg-rectified output voltage, iLInductor current, L-inductor, Qb—QfSwitching tube, Dfw—Dsk-diode, CoOutput filter capacitor, RL-load, VoOutput voltage, Rs_b/b—Rs_b/bSampling resistance, VrefReference voltage, v, of feedback control of output voltageEAOutput voltage feedback controlled error voltage signal output, t-time, ω -input voltage angular frequency, VmInput voltage peak value, vgs_b-switching tube QbV drive voltage ofgs_b/b-switching tube QfDriving voltage of Ts-converter switching period, fs-converter switching frequency, PF-power factor, IL_pkPeak value of inductor current, Iin_rmsEffective value of input current, tonConverter on-time, toffConverter off time, iinInput current, PinInput power, Δ Vo-output voltage ripple.
Detailed Description
Fig. 1 is a Buck-flyback pfc converter main circuit.
The following assumptions are made: (1) all devices are ideal elements; (2) the output voltage ripple is very small compared to its dc amount; (3) the switching frequency is much higher than the input voltage frequency.
Fig. 2 shows waveforms of the switching tube current and the inductor current in one switching period when the inductor current is critical and continuous, wherein fig. 2(a) is a waveform diagram when the Buck topology works, and fig. 2(b) is a waveform diagram when the Flyback topology works. When the input voltage vgLess than the output voltage VoTime, Flyback topology works, QbOff, QfWhen conducting, DoCut-off, inductance LpVoltage across vgCurrent of i thereofLpStarting from zero with vg/LpIs increased linearly, and outputs a filter capacitor CoSupplying power to a load; when Q isfAt turn-off, according to ampere-turn conservation iLsBy DoFollow current, at this time LsThe voltage across is-Vo，iLsWith Vo/LsIs decreased, and iLsMay drop to zero at the start of a new cycle. When the input voltage vgGreater than the output voltage VoTime, Buck topology works, QfOff, QbWhen conducting, DfwCut-off, inductance LsVoltage across vg-VoCurrent of i thereofLsStarting from zero with (v)g-Vo)/LsIs linearly increased, vgTo an output filter capacitor CoAnd load power supply; when Q isbWhen turned off, iLsBy DoFollow current, at this time LsThe voltage across is-Vo，iLsWith VoThe slope of/L decreases, and iLsMay drop to zero at the start of a new cycle.
Without loss of generality, v defining the input AC voltageinIs expressed as
vin＝Vmsinωt (1)
Wherein VmAnd ω is the amplitude and angular frequency of the input ac voltage, respectively.
Then the voltage of the input voltage rectified by the rectifier bridge becomes
vg＝Vm|sinωt| (2)
In a half power frequency period, the converter is divided into two working stages of Buck topology work and Flyback topology work. When the rectified input voltage vgGreater than the output voltage VoThe Buck topology works. If the conduction time of the Buck topological switch tube is assumed to be ton_buckWhen the switch tube is on, the input voltage vgThe peak value i of the secondary side inductance current in a switching period in a Buck mode can be obtained by supplying power to the inductor, the output capacitor and the loadLs_pk_buckThe expression of (a) is as follows:
Figure GDA0002984375120000051
wherein VoTo output a voltage, LsThe inductance value of the secondary side inductor.
When the switch tube turns off the inductor to follow current, the turn-off time t can be obtained according to the volt-second balance principleoff_buckExpression (c):
Figure GDA0002984375120000052
an expression of switching frequency in Buck stage can be obtained according to the formula (4):
Figure GDA0002984375120000053
according to the formula (3) and the formula (5), the average value of the current flowing through the inductor in one switching period when the Buck topology works is obtained as
Figure GDA0002984375120000054
When the input voltage vgLess than the output voltage VoThe Flyback topology works. If the conduction time of the Flyback topological switch tube is assumed to be ton_flybackWhen the switch tube is on, the input voltage vgThe peak value i of the current of the primary side inductor in one switching period can be obtained by supplying power to the primary side inductorLp_pk_flybackIs expressed as follows
Figure GDA0002984375120000055
In the formula LpThe inductance value of the primary side inductance is shown.
When the switch tube is turned off, the secondary side inductor LsFollow current and inductance supply power to the capacitor and the load, and the secondary side inductance peak value i can be obtained according to the ampere-turn conservation and volt-second balance principleLs_pkAnd off time toff_flybackExpression (c):
iLs_pk(ωt)＝niLp_pk(ωt) (8)
Figure GDA0002984375120000061
wherein
Figure GDA0002984375120000062
An expression of the switching frequency in the Flyback stage can be obtained from equation (9):
Figure GDA0002984375120000063
according to the formula (7) and the formula (10), the average value of the input current flowing through one switching period of the Flyback topology operation is obtained
Figure GDA0002984375120000064
Because the Flyback topology works in the dead zone where the Buck circuit cannot work, the converter has no dead zone in the whole power frequency period. The input current i is derived from the aboveinComprises the following steps:
Figure GDA0002984375120000065
wherein
Figure GDA0002984375120000066
According to the formula (6) and the formula (11), the expression of the switching frequency in a half power frequency period can be obtained:
Figure GDA0002984375120000067
when the conduction times in the two operating phases are the same and constant, i.e.
ton＝ton_buck＝ton_flyback (14)
The waveform of the input current within a half power frequency period at different input voltages can be plotted according to equation (13), as shown in fig. 3; the waveform of the switching frequency variation of the switching tube within a half power frequency period under different input voltages can be plotted according to equation (14), as shown in fig. 4. As can be seen from fig. 3, although the Flyback converter compensates the dead zone part of the input current of the Buck converter in the low voltage range, the waveform of the input current is far from the sine wave, and the harmonic content is much. As can be seen from fig. 4, the switching frequency of the switching tubes of the Buck topology and the Flyback topology based on the constant on-time control method has a large variation range within a half power frequency period, which is not favorable for the design of EMI.
Assuming 100% efficiency of the converter, i.e. input power equal to output power, it is possible to obtain from the power balance
Figure GDA0002984375120000071
The conventional control may be formulated for on-time according to equation (16):
Figure GDA0002984375120000072
calculating the expression of PF according to the above formula
Figure GDA0002984375120000073
The PF curve under the conventional control can be made according to equation (18), as shown in fig. 5. As can be seen from the graph, the PF value increases as the input voltage increases. In the low voltage range, the PF value is low, and at an input voltage of 90V, the PF value is only 0.888.
The above reasoning shows that under the conventional control, the switching frequency variation range of the switching tube in both Buck topology and Flyback topology is relatively large, which is very unfavorable for the design of EMI. The idea of the constant switching frequency control strategy is to make the switching frequencies of the switching tubes of the Buck topology and the Flyback topology constant and equal in a half power frequency period, namely to make
Figure GDA0002984375120000074
The expressions of the conduction time of the Buck topology switching tube and the Flyback topology switching tube in a half power frequency period can be obtained according to the expression (18):
Figure GDA0002984375120000075
Figure GDA0002984375120000081
according to the expressions (7), (12), (21) and (22), an expression of the input current in a half power frequency period can be obtained:
Figure GDA0002984375120000082
2.2 optimal turns ratio selection
Suppose the output power of the converter is PoThe efficiency of the converter is 100%, which can be obtained according to the power balance principle:
Figure GDA0002984375120000083
an expression of K can be obtained from equation (24).
Figure GDA0002984375120000084
An expression of PF at double fixed frequencies can be written according to the formula (22)
Figure GDA0002984375120000085
The power factor curve of the CRM Buck-Flyback PFC converter with constant switching frequency control can be plotted according to equation (24) at different turns ratios, as shown in fig. 6. As can be seen from the figure, when the turn ratio n is 2, the PF curve of the converter is the highest, and the converter can obtain the optimal power factor.
From the above analysis, it can be seen that to keep the switching frequency of the converter constant in the switching period, it is only necessary to make the converter operate in the Buck mode and the Flyback mode for the on-time ton_buckAnd ton_flybackBoth of them vary according to formula (19) and formula (20), but formula (19) and formula (20) relate to Vo、VmL and PoThe function (2) has more independent variables, and if an analog circuit is used for building the control circuit, the feedforward control circuit is very complicated. The design adopts digital feedforward circuit control to input the voltage peak value VmAn output voltage VoThe equal parameters are input into the control chip through the sampling circuit and sampled, the peak value envelope curve of the inductive current is calculated through the calculation function of the control chip, and the peak value envelope curve is output to the drive generation circuit through the digital-to-analog conversion circuit. The control circuit is shown in fig. 12.
The drive signal generating circuit can adopt an integrated IC circuit of a model L6561 or a model L6562, an amplifier used in the output voltage differential sampling circuit (4) and the output voltage feedback circuit (6) is an operational amplifier of a model TL074, a model TL072, a model LM358, a model LM324 and the like, the multiplier (10) is formed by adopting an integrated IC circuit or a discrete device, an AND gate used in the state judging circuit (7) and the drive signal generating circuit (8) is a logic chip of a model SN74HC08N, a model CD4011BE or a model 74HC32N and the like, the drive circuit can adopt a drive chip of a model IR2110, a model TLP2590 and the like or adopt a totem pole drive circuit, and a control chip can adopt a chip of a model TMS320F28335 or TMS320F28377 and the like.
It can be seen from equation (18) that the switching frequency of the two topology switching tubes in a half power frequency period is kept at a constant value when the constant switching frequency is adopted for control under different input voltage levels. Compared with the traditional control, the change range of the switching frequency of the two switching tubes is greatly reduced, and the design of EMI is simplified. FIG. 7 shows the switching frequency variation waveforms for different control modes in the power frequency cycle at 110VAC and 220 VAC.
PF value change curves in the conventional control and the constant switching frequency control can be made from equations (24) and (17), respectively, as shown in fig. 8. As can be seen from the figure, compared with the traditional constant on-time control in a low voltage range, the constant switching frequency control has the advantages that the PF is obviously improved; the PF controlled by the constant switching frequency is reduced to a certain extent in a high-voltage range, but compared with the lowest PF of constant conduction time, namely '0.88', the PF controlled by the constant switching frequency can be kept above 0.91, so that the PF controlled by the constant switching frequency is obviously improved compared with the conventional PF controlled by the constant conduction frequency.
In order to analyze the harmonics of the input current, fourier analysis may be performed on equation (21). The fourier decomposed form of the input current is:
Figure GDA0002984375120000091
wherein
Figure GDA0002984375120000101
Figure GDA0002984375120000102
In the formula TlineIs the power frequency cycle.
The harmonic waves contained in the input current under the control of the constant switching frequency can be obtained by calculating the formula (21) instead of the formula (26). Wherein, the cosine component and the even-order sine component are both 0.
The 3, 5 and 7 harmonic current amplitudes I of constant switching frequency control and traditional constant on-time control can be obtained by the formulas (21) and (26)3、I5、I7For fundamental current amplitude I1Per unit value of
Figure GDA0002984375120000103
As shown in fig. 9.
According to IEC61000-3-2, Class D standard requirements, the ratio of the input current of 3, 5, 7, 9 subharmonics to the input power should satisfy the formula (27)
Figure GDA0002984375120000104
Figure GDA0002984375120000105
Figure GDA0002984375120000106
It can be seen from fig. 9 that the 3, 5, 7 th harmonics are below the limits of IEC61000-3-2, Class D standard at any input voltage.
The square of the effective values of the primary side inductance current and the secondary side inductance current in the Buck mode and the Flyback mode in one switching period is obtained from the formula (3), the formula (4), the formula (7) and the formula (9)
Figure GDA0002984375120000107
Figure GDA0002984375120000108
Figure GDA0002984375120000109
The root mean square is calculated from the above formula in half power frequency period to obtain the effective current values of the primary side and the secondary inductance
Figure GDA0002984375120000111
Figure GDA0002984375120000112
By substituting the above equations (16), (19) and (20), the effective value variation curve of the inductor current under the conventional control method and the constant switching frequency control can be obtained, as shown in fig. 10. As can be seen from the figure, the effective value of the inductor current is greatly reduced in the constant switching frequency control mode, and the efficiency of the entire converter is greatly improved.
Calculating formula according to instantaneous input power per unit value
Figure GDA0002984375120000113
Substituting the equation (12) into the above equation to obtain the per unit value p of the instantaneous input power under the conventional control* in_cot(ii) a Substituting the equation (21) into the above equation can obtain the per unit value p of instantaneous input power under the control of constant switching frequency* in_dcf。
When in use
Figure GDA0002984375120000114
Time, energy storage capacitor CoCharging; when in use
Figure GDA0002984375120000115
When, CoAnd (4) discharging. Assume that ω t is 0, and constant on-time control and variable on-time control are performed
Figure GDA0002984375120000116
The time axis coordinate corresponding to the intersection of the waveform of (1) and (1) is t1And t2Then energy storage capacitor CoThe per unit values of the maximum energy (the reference value is the output energy in half power frequency period) stored in half power frequency period are respectively
Figure GDA0002984375120000117
Figure GDA0002984375120000118
According to the calculation formula of the capacitance energy storage,
Figure GDA0002984375120000119
and
Figure GDA00029843751200001110
and can be expressed as
Figure GDA00029843751200001111
Figure GDA00029843751200001112
Wherein Δ Vo_1And Δ Vo_2The ripple value of the output voltage under the control of fixed on-time and constant switching frequency respectively.
Fig. 8 is made from equations (36) and (37), and it can be seen that the output voltage ripple under the control with the constant switching frequency is much reduced compared to the conventional constant on-time control, and when the input voltage is 264VAC, the output voltage ripple under the control with the constant switching frequency is only 46.5% of that under the conventional control.

Claims (4)

1. A CRM buck-flyback PFC converter controlled in a fixed frequency mode is characterized by comprising a main power circuit (1) and a control circuit, wherein the main power circuit (1) comprises an input power supplyv in EMI filter, rectifier diodeD 1 -D 6 Primary side inductor of main circuitL p Secondary inductance of main circuitL s Zero-crossing detection inductorL ZCD Switch tubeQ b And a switching tubeQ f Freewheel diodeD o Output capacitorC o Load, and method of operating the sameR L ；
Input voltage sourcev in Is connected to the input port of the EMI filter,
output port and rectifier diode of EMI filterD 1 -D 4 Composed of a rectifier bridge andD 3 -D 6 the input ports of the formed rectifier bridges are connected,
rectifier diodeD 3 、D 4 The positive electrode is a reference potential zero point,
rectifier diodeD 5 、D 6 Primary side inductance of negative and main circuitL p The connection is carried out in a connecting way,
primary side inductor of main circuitL p The other end of the switch tubeQ f One end of the first and second connecting rods is connected,
rectifier diodeD 1 、D 2 Secondary side inductance of cathode and main circuitL s Is connected with the freewheeling diodeD o The negative electrode of the anode is connected with the anode,
secondary inductor of main circuitL s Different name terminal and output capacitorC o Is connected to one end of a load RL,
output capacitorC o With the other end of the load RL together with the freewheeling diodeD o The anode being connected to the switching tubeQ b At one end of the first and second arms,
zero-crossing detection inductorL ZCD With main circuit inductanceLpAndL s are all coupled to each other and are,
zero-crossing detection inductorL ZCD One end of the current limiting resistor is connected with a reference potential zero point and the other end is connected with a current limiting resistorR ZCD The connection is carried out by connecting the two parts,
current sampling deviceResistance deviceR s One end of the switch tube is grounded and the other end of the switch tube is connected with the switch tubeQ f And a switching tubeQ b Connecting;
the control circuit comprises a main control chip (2), a sampling circuit (3), a digital-to-analog conversion circuit (5), an output voltage difference circuit (4), a feedback circuit (6), a state judgment circuit (7), a driving signal generation circuit (8), a comparison circuit (9) and a multiplier (10);
the positive input end of the output voltage differential circuit (4) and the output voltage positive electrode of the main power circuit (1)V o +The connection is carried out in a connecting way,
the negative input end of the output voltage differential circuit (4) and the output voltage positive electrode of the main power circuit (1)V o -The connection is carried out in a connecting way,
the output end of the output voltage differential circuit (4) is connected with the first input end of the sampling circuit (3) and the input end of the feedback circuit (6),
the second input end of the sampling circuit (3) and the rectified voltage of the main circuitv g The connection is carried out in a connecting way,
the first output end of the sampling circuit (3) is connected with an ADC2 port of the main control chip (2),
the second output end of the sampling circuit (3) is connected with an ADC1 port of the main control chip (2),
a first output port of the main control chip (2) is connected with an input port of the digital-to-analog conversion circuit (5),
an output port of the digital-to-analog conversion circuit (5) and an input end of the multiplierv x The connection is carried out by connecting the two parts,
the output port of the feedback circuit (6) and the input end of the multiplierv y The connection is carried out by connecting the two parts,
the output end of the multiplier is connected with the positive input end of the comparison circuit (9),
negative input end of comparison circuit (9) and main circuitv Rs The connection is carried out by connecting the two parts,
the output end of the comparison circuit (9) is connected with the input end of the driving signal generation circuit (8),
the input end of the driving signal generating circuit (8) passes through a current limiting resistorR ZCD Zero-crossing detection winding of main circuitL ZCD The connection is carried out by connecting the two parts,
the output end of the driving signal generating circuit (8) is connected with the input end of the state judging circuit (7),
end of the driving signal generating circuit (8) and a switch tube of the main power circuit (1)Q b The connection is carried out by connecting the two parts,
end of the driving signal generating circuit (8) and a switch tube of the main power circuit (1)Q f And (4) connecting.
2. The converter according to claim 1,
the sampling circuit (3) converts the input voltage of the converterv g And an output voltageV o Converting into voltage capable of being collected by control chipv g_dsp AndV o_dsp ；
the main control chip (2) comprises a phase locking program, an input voltage peak value sampling program, an output voltage sampling program and a control program,v g_dsp and is connected with ADC1 interface of control chip, and can utilize the sampled value to give it to phase-locking procedure and input voltage peak value sampling procedure,V o_dsp the sampling device is connected with an ADC2 interface of the control chip, the sampled numerical value is sent to an output voltage sampling program, the output of a phase locking program, an input voltage peak value sampling program and an output voltage sampling program is sent to a main control program, and the main control program calculates the numerical value of the profile of the inductive current;
the input end of the digital-to-analog conversion circuit (5) is connected with the control chip through an SPI protocol, and the digital-to-analog conversion circuit converts the numerical value of the received inductive current profile into voltage to be output.
3. Converter according to claim 1, characterized in that said state decision circuit (7) comprises a first comparator Comp1, a resistorR 5 、R 6 Reference voltage sourceV boundary Two AND gates, a NOT gate and a drive circuit;
the equidirectional input terminal of the first comparator Comp1 passes through a voltage-dividing resistorR 5 、R 6 Voltage rectified by diode rectifying circuit of main power circuit (1)v g The connection is carried out in a connecting way,
first comparatorComp1 inverting input terminal and reference voltage sourceV boundary The connection is carried out by connecting the two parts,
first comparatorCompThe output terminal of 1 is connected to one input terminal of the first AND Gate1 AND the input terminal of the not Gate,
the output of the not-Gate is connected to one input of a second AND-Gate AND 2,
the output ends of the two AND gates are connected with the input end of a driving circuit, and the output ends of the driving circuit are respectively connected with two switching tubes of the main power circuit (1)Q b AndQ f 。
4. the converter according to claim 1, wherein the driving signal generating circuit is an integrated IC circuit of model L6561 or L6562, the amplifiers used in the output voltage difference circuit (4) and the feedback circuit (6) are operational amplifiers of model TL074, TL072, LM358 or LM324, the multiplier (10) is composed of an integrated IC circuit or discrete devices, the and gate used in the state judging circuit (7) is a logic chip of model SN74HC08N, CD4011BE or 74HC32N, the driving circuit is a driving chip of model IR2110 or TLP2590 or a totem pole driving circuit, and the control chip is a chip of model TMS320F28335 or TMS320F 28377.
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Citations (8)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5552695A (en) * 1994-03-22 1996-09-03 Linear Technology Corporation Synchronously rectified buck-flyback DC to DC power converter
US5636114A (en) * 1995-11-30 1997-06-03 Electronic Measurements, Inc. Lossless snubber circuit for use in power converters
CN103647448A (en) * 2013-12-09 2014-03-19 杭州士兰微电子股份有限公司 Integrated step-down-flyback type high power factor constant current circuit and device
CN103856059A (en) * 2012-11-30 2014-06-11 英飞凌科技奥地利有限公司 Buck-flyback converter
CN102647100B (en) * 2012-05-11 2014-09-17 杭州电子科技大学 Integrated Buck-flyback high power factor converter
CN107800312A (en) * 2017-11-13 2018-03-13 四川大学 A kind of output ripple and low pfc converter
CN109149963A (en) * 2017-06-28 2019-01-04 南京理工大学 The DCM of switch periods optimal utilization rate control is depressured pfc converter
CN109309447A (en) * 2017-07-28 2019-02-05 南京理工大学 The CRM of constant switching frequency control is depressured pfc converter

Patent Citations (8)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5552695A (en) * 1994-03-22 1996-09-03 Linear Technology Corporation Synchronously rectified buck-flyback DC to DC power converter
US5636114A (en) * 1995-11-30 1997-06-03 Electronic Measurements, Inc. Lossless snubber circuit for use in power converters
CN102647100B (en) * 2012-05-11 2014-09-17 杭州电子科技大学 Integrated Buck-flyback high power factor converter
CN103856059A (en) * 2012-11-30 2014-06-11 英飞凌科技奥地利有限公司 Buck-flyback converter
CN103647448A (en) * 2013-12-09 2014-03-19 杭州士兰微电子股份有限公司 Integrated step-down-flyback type high power factor constant current circuit and device
CN109149963A (en) * 2017-06-28 2019-01-04 南京理工大学 The DCM of switch periods optimal utilization rate control is depressured pfc converter
CN109309447A (en) * 2017-07-28 2019-02-05 南京理工大学 The CRM of constant switching frequency control is depressured pfc converter
CN107800312A (en) * 2017-11-13 2018-03-13 四川大学 A kind of output ripple and low pfc converter

Non-Patent Citations (2)

* Cited by examiner, † Cited by third party
Title
Variable-On-Time Control to Achieve High Input Power Factor for a CRM-Integrated Buck-Flyback PFC Converter;Qingwei Chen等;《IEEE TRANSACTIONS ON POWER ELECTRONICS》;20170731;第32卷(第7期);第5312-5322页 *
一种同步整流Buck-flyback拓扑的研究;周岩等;《电力电子技术》;20100131;第44卷(第1期);第39-40,54页 *

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