CN113514079B

CN113514079B - Frequency modulation gyro Lissajous modulation and self-correction test system

Info

Publication number: CN113514079B
Application number: CN202110318786.1A
Authority: CN
Inventors: 李崇; 刘志鹏; 孟相睿; 王雨晨; 侯佳坤; 王鑫宁
Original assignee: Ocean University of China
Current assignee: Ocean University of China
Priority date: 2021-03-25
Filing date: 2021-03-25
Publication date: 2023-12-22
Anticipated expiration: 2041-03-25
Also published as: CN113514079A

Abstract

The invention belongs to the technical field of MEMS gyroscopes, and discloses a Lissajous modulation and self-correction test system of a frequency modulation gyro, which aims at the defect that the existing Lissajous frequency modulation gyro lacks a self-correction system, and designs a novel ADC auxiliary topology structure. The invention utilizes the topological structure to carry out frequency tracking control and amplitude stabilizing control, is in a closed-loop operation mode, realizes high-stability control of vibration frequency and amplitude by separating and calculating a driving signal and an angular rate signal, and inhibits zero drift error caused by resonance frequency change and damping mismatch.

Description

Frequency modulation gyro Lissajous modulation and self-correction test system

Technical Field

The invention belongs to the technical field of MEMS gyroscopes, and relates to a frequency modulation gyro Lissajous modulation and self-correction test system.

Background

In recent years, the MEMS gyro technology has been rapidly developed, and its application in the fields of industry, military and the like is becoming wider due to its small size, low power consumption, low cost and the like, but its application in the tip fields of high performance tactical weapons, robots and the like is limited due to its poor zero point stability. The novel MEMS gyroscope modulation and self-correction technology is an effective way for improving the zero point stability of the MEMS gyroscope. The modulation mode of the MEMS gyroscope is divided into two types of amplitude modulation and frequency modulation.

The traditional scheme is mainly an amplitude modulation system for an MEMS gyroscope modulation and self-correction system, but because of the characteristic that amplitude signals are easy to interfere, the amplitude modulation gyroscope still has quite obvious zero drift problem even under a constant temperature control environment and various correction methods. With the continuous progress of technology, the frequency modulation gyro technology has the advantages that compared with amplitude signals, frequency signals have ultrahigh stability and are not easy to be interfered by external environments, and various defects of the amplitude modulation gyro can be well overcome.

The frequency modulation gyro is divided into quadrature frequency modulation and Lissajous frequency modulation. Most of the conventional frequency modulation gyro systems are orthogonal frequency modulation systems, and have the advantages of mode matching, unlimited bandwidth, reliable scale factors and the like, but the stability of the resonant frequency still has strong dependence, and the requirement on a frequency measurement circuit is extremely high, so that the application of the frequency modulation gyro system is very limited. Quadrature frequency modulation systems have extremely stringent frequency matching requirements and zero rate offset directly related to modal natural frequency drift, which theoretically cannot take advantage of fm gyroscopes. The Lissajous frequency modulation well solves the defect of quadrature frequency modulation, has the characteristics of low requirement on the symmetry of the gyroscope, is insensitive to temperature change, has high stability, has low requirement on hardware and the like in theory, and can achieve better performance.

However, existing lissajous frequency modulation gyro systems lack a correction system formulated for mechanical errors and are mostly open-loop operation methods, so that zero drift errors in MEMS gyro mechanical systems cannot be counteracted.

Disclosure of Invention

Aiming at the technical problems of the Lissajous frequency modulation gyro system in the prior art, the invention provides a frequency modulation gyro Lissajous modulation and self-correction test system, which adopts the following technical scheme:

a kind of modulation and self-correction test system of the Lissajous of frequency modulation gyro, including FDC circuit, ADC circuit, IQ demodulation circuit, coherent demodulation circuit, phase modulator, phase-locked loop circuit, automatic gain control circuit, numerical control oscillator, multiplier, adder, error control circuit, DAC circuit, low-pass filter and electrical interface;

the FDC circuits are two, namely a first FDC circuit and a second FDC circuit;

the two ADC circuits are respectively a first ADC circuit and a second ADC circuit;

the IQ demodulation circuits are two, namely a first IQ demodulation circuit and a second IQ demodulation circuit;

the coherent demodulation circuit and the phase modulator are respectively provided with one phase modulator;

the phase-locked loop circuits are two, namely a first phase-locked loop circuit and a second phase-locked loop circuit;

The automatic gain control circuits are two, namely a first automatic gain control circuit and a second automatic gain control circuit;

the number of the numerical control oscillators is two, namely a first numerical control oscillator and a second numerical control oscillator;

the number of the multipliers is two, namely a first multiplier and a second multiplier; the adder is provided with one adder;

the two error control circuits are respectively a first error control circuit and a second error control circuit;

the DAC circuits are two, namely a first DAC circuit and a second DAC circuit;

three low-pass filters are respectively a first low-pass filter, a second low-pass filter and a third low-pass filter;

six electrical interfaces are respectively a first electrical interface, a second electrical interface, a third electrical interface, a fourth electrical interface, a fifth electrical interface and a sixth electrical interface;

the input end of the first FDC circuit and the input end of the first ADC circuit are connected with a sixth electrical interface, and the input end of the second FDC circuit and the input end of the second ADC circuit are connected with a fifth electrical interface;

the output end of the first ADC circuit is connected with the input end of the first IQ demodulation circuit; the output end of the first IQ demodulation circuit is respectively connected with the input ends of the first phase-locked loop circuit and the first automatic gain control circuit;

The output end of the second ADC circuit is connected with the input end of the second IQ demodulation circuit; the output end of the second IQ demodulation circuit is respectively connected with the input ends of the second phase-locked loop circuit and the second automatic gain control circuit;

the output ends of the first phase-locked loop circuit and the first automatic gain control circuit are connected with the input end of the first numerical control oscillator;

the output ends of the second phase-locked loop circuit and the second automatic gain control circuit are connected with the input end of the second digital control oscillator;

the output end of the first numerical control oscillator is respectively connected to the input end of the first multiplier, the input end of the second multiplier, the input end of the first error control circuit and the input end of the first DAC circuit; the output ends of the first error control circuit and the first DAC circuit are respectively connected with the fourth electrical interface and the first electrical interface;

the output end of the second digital control oscillator is connected with the input end of the first multiplier, the input end of the second error control circuit and the input end of the second DAC circuit; the output ends of the second error control circuit and the second DAC circuit are respectively connected with the third electric interface and the second electric interface;

output end of the first multiplier the output end of the first multiplier is connected with the input ends of the first low-pass filter and the phase modulator in turn; the output end of the second multiplier is sequentially connected with the input ends of the second low-pass filter and the phase modulator;

The output end of the phase modulator is connected with the input end of the coherent demodulation circuit; the output ends of the first FDC circuit and the second FDC circuit are respectively connected to the input end of the adder, and the output end of the adder is connected with the input end of the coherent demodulation circuit;

the output end of the coherent demodulation circuit is connected with a third low-pass filter, and the third low-pass filter is connected with a rate signal output interface.

Preferably, the frequency modulation gyro comprises a first driving input electrode, a second driving input electrode, a first tuning input electrode, a second tuning input electrode, a first sensing output electrode and a second sensing output electrode;

wherein the first drive input electrode is connected with the first electrical interface, and the second drive input electrode is connected with the second electrical interface;

the first tuning input electrode is connected with the fourth electric interface, and the second tuning input electrode is connected with the third electric interface;

the first inductive output electrode is connected with the sixth electrical interface, and the second inductive output electrode is connected with the fifth electrical interface.

Preferably, the frequency modulation gyro adopts an MEMS gyro equivalent circuit, which comprises a first vibration mode circuit, a second vibration mode circuit and a coupling circuit positioned between the first vibration mode circuit and the second vibration mode circuit;

The first vibration mode circuit comprises a first resistor, a first capacitor and a first inductor which are sequentially connected in series;

the second vibration mode circuit comprises a second resistor, a second capacitor and a second inductor which are sequentially connected in series;

the coupling circuit comprises an operational amplifier, a mutual inductor, a VGA and a potentiometer;

the number of the operational amplifiers is six, namely a first operational amplifier, a second operational amplifier, a third operational amplifier, a fourth operational amplifier, a fifth operational amplifier and a sixth operational amplifier;

sixteen transformers are respectively a first transformer, a second transformer, a third transformer, a fourth transformer, a fifth transformer, a sixth transformer, a seventh transformer, an eighth transformer, a ninth transformer, a tenth transformer, an eleventh transformer, a twelfth transformer, a thirteenth transformer, a fourteenth transformer, a fifteenth transformer and a sixteenth transformer;

the two VGAs are respectively a first VGA and a second VGA;

eight potentiometers are respectively a first potentiometer, a second potentiometer, a third potentiometer, a fourth potentiometer, a fifth potentiometer, a sixth potentiometer, a seventh potentiometer and an eighth potentiometer;

the input end of the first transformer is connected with a first driving input electrode, and the output end of the first transformer is connected in series with a first vibration mode circuit;

the input end of the second transformer is connected with a second driving input electrode, and the output end of the second transformer is connected in series with a second vibration mode circuit;

the positive phase input end and the negative phase input end of the first operational amplifier are respectively connected to one end part of the first capacitor, and the output end of the first operational amplifier is connected to the input end of the twelfth transformer; the output end of the twelfth transformer is connected in series with a second vibration mode circuit;

The positive phase input end and the negative phase input end of the sixth operational amplifier are respectively connected to one end part of the second capacitor, and the output end of the sixth operational amplifier is connected to the input end of the sixth transformer; the output end of the sixth transformer is connected in series with the first vibration mode circuit;

the input end of the fifth transformer is connected in series with the first vibration mode circuit, and the output end of the fifth transformer is respectively connected with the non-inverting input end and the inverting input end of the second operational amplifier; the output end of the second operational amplifier is connected to the input end of the eleventh transformer;

the output end of the eleventh transformer is connected in series with a second vibration mode circuit;

the input end of the thirteenth transformer is connected in series with the second vibration mode circuit, and the output end of the thirteenth transformer is respectively connected with the non-inverting input end and the inverting input end of the fifth operational amplifier; the output end of the fifth operational amplifier is connected to the input end of the fourth transformer;

the output end of the fourth transformer is connected in series with the first vibration mode circuit;

the input end of the seventh transformer is connected in series with the first vibration mode circuit, and the output end of the seventh transformer is respectively connected with the non-inverting input end and the inverting input end of the third operational amplifier; the output end of the third operational amplifier is sequentially connected with the input ends of the first VGA and the tenth transformer;

the output end of the tenth transformer is connected in series with a second vibration mode circuit;

The input end of the fifteenth transformer is connected in series with a second vibration mode circuit, and the output end of the fifteenth transformer is respectively connected with the non-inverting input end and the inverting input end of the fourth operational amplifier; the output end of the fourth operational amplifier is sequentially connected with the input ends of the second VGA and the third transformer;

the output end of the third transformer is connected in series with a first vibration mode circuit;

one end of the first potentiometer is connected with the non-inverting input end of the first operational amplifier, and the other end of the first potentiometer is grounded;

the second potentiometer is connected between the negative phase input end of the first operational amplifier and the output end of the first operational amplifier;

the third potentiometer is connected between the negative phase input end of the second operational amplifier and the output end of the second operational amplifier;

the fourth potentiometer is connected between the negative phase input end of the third operational amplifier and the output end of the third operational amplifier;

one end of the fifth potentiometer is connected with the non-inverting input end of the sixth operational amplifier, and the other end of the fifth potentiometer is grounded;

the sixth potentiometer is connected between the negative phase input end of the sixth operational amplifier and the output end of the sixth operational amplifier;

the seventh potentiometer is connected between the negative phase input end of the fifth operational amplifier and the output end of the fifth operational amplifier;

The eighth potentiometer is connected between the negative phase input end of the fourth operational amplifier and the output end of the fourth operational amplifier;

the input end of the eighth transformer is connected with the first tuning input electrode, and the output end of the eighth transformer is connected in series with the first vibration mode circuit; the input end of the fourteenth transformer is connected with a second tuning input electrode, and the output end of the fourteenth transformer is connected in series with a second vibration mode circuit;

the input end of the ninth transformer is connected in series with the first vibration mode circuit, and the output end of the ninth transformer is connected with the first induction output electrode; the input end of the sixteenth transformer is connected in series with the second vibration mode circuit, and the output end of the sixteenth transformer is connected with the second induction output electrode.

The invention has the following advantages:

as described above, the invention relates to a Lissajous modulation and self-correction test system of a frequency modulation gyro, which aims at the defect that the existing Lissajous frequency modulation gyro lacks a self-correction system, and designs a novel topological structure with an ADC auxiliary type to carry out frequency tracking control and amplitude stabilizing control.

Drawings

FIG. 1 is a schematic diagram of a system for testing the modulation and self-calibration of a frequency-modulated gyro Lissajous modulation system in an embodiment of the present invention;

FIG. 2 is a block diagram of an equivalent circuit of a MEMS gyroscope in an embodiment of the invention;

fig. 3 is a schematic structural diagram of an equivalent circuit of the MEMS gyroscope according to an embodiment of the present invention.

Wherein, 101 a-a first resistor, 101 b-a second resistor, 101 c-a third resistor, 101 d-a fourth resistor, 101 e-a fifth resistor, and 101 f-a sixth resistor; 102 a-first capacitance, 102 b-second capacitance, 103 a-first inductance, 103 b-second inductance; 104 a-first operational amplifier, 104 b-second operational amplifier, 104 c-third operational amplifier, 104 d-fourth operational amplifier, 104 e-fifth operational amplifier, 104 f-sixth operational amplifier; 105 a-first, 105 b-second, 105 c-third, 105 d-fourth, 105 e-fifth, 105 f-sixth, 105 g-seventh, 105 h-eighth, 105 i-ninth, 105 j-tenth, 105 k-eleventh, 105 l-twelfth, 105 m-thirteenth, 105 n-fourteenth, 105 o-fifteenth, 105 p-sixteenth; 106 a-first VGA,106 b-second VGA;107 a-first potentiometer, 107 b-second potentiometer, 107 c-third potentiometer, 107 d-fourth potentiometer, 107 e-fifth potentiometer, 107 f-sixth potentiometer, 107 g-seventh potentiometer, 107 h-eighth potentiometer; 108 a-first drive input electrode, 108 b-second drive input electrode, 109 a-first tuning input electrode, 109 b-second tuning input electrode, 110 a-first sense output electrode, 110 b-second sense output electrode; 111 a-first electrical interface, 111 b-second electrical interface, 111 c-third electrical interface, 111 d-fourth electrical interface, 111 e-fifth electrical interface, 111 f-sixth electrical interface; 201 a-first FDC circuit, 201 b-second FDC circuit, 202 a-first ADC circuit, 202 b-second ADC circuit, 203 a-first IQ demodulation circuit, 203 b-second IQ demodulation circuit, 204-coherent demodulation circuit, 205-phase modulator, 206 a-first phase-locked loop circuit, 206 b-second phase-locked loop circuit, 207 a-first automatic gain control circuit, 207 b-second automatic gain control circuit, 208 a-first digitally controlled oscillator, 208 b-second digitally controlled oscillator, 209 a-first multiplier, 209 b-second multiplier, 210-adder, 211 a-first error control circuit, 211 b-second error control circuit, 212 a-first DAC circuit, 212 b-second DAC circuit, 213 a-first low pass filter, 213 b-second low pass filter, 213 c-third low pass filter, 214-rate signal output interface.

Detailed Description

The invention is described in further detail below with reference to the attached drawings and detailed description:

examples

As shown in fig. 1, the invention relates to a frequency modulation gyro lissajous modulation and self-correction test system, which comprises an FDC circuit, an ADC circuit, an IQ demodulation circuit, a coherent demodulation circuit, a phase modulator, a phase-locked loop circuit, an automatic gain control circuit, a digital controlled oscillator, a multiplier, an adder, an error control circuit, a DAC circuit, a low-pass filter and an electrical interface.

The FDC circuit, the ADC circuit, the IQ demodulation circuit, the coherent demodulation circuit, the phase modulator, the phase-locked loop circuit, the automatic gain control circuit, the IQ demodulation circuit, the error control circuit, the DAC circuit and the like can adopt a mature scheme.

There are two FDC circuits, a first FDC circuit 201a and a second FDC circuit 201b, respectively.

There are two ADC circuits, a first ADC circuit 202a and a second ADC circuit 202b, respectively.

The IQ demodulation circuits include two IQ demodulation circuits, namely a first IQ demodulation circuit 203a and a second IQ demodulation circuit 203b.

The coherent demodulation circuit 204 and the phase modulator 205 are each one.

There are two phase-locked loop circuits, a first phase-locked loop circuit 206a and a second phase-locked loop circuit 206b.

The automatic gain control circuit includes a first automatic gain control circuit 207a and a second automatic gain control circuit 207b.

There are two digitally controlled oscillators, a first digitally controlled oscillator 208a and a second digitally controlled oscillator 208b, respectively.

Two multipliers are respectively a first multiplier 209a and a second multiplier 209b; adder 210 has one.

There are two error control circuits, a first error control circuit 211a and a second error control circuit 211b, respectively.

There are two DAC circuits, a first DAC circuit 212a and a second DAC circuit 212b, respectively.

The low-pass filters include a first low-pass filter 213a, a second low-pass filter 213b, and a third low-pass filter 213c.

There are six electrical interfaces, namely, a first electrical interface 111a, a second electrical interface 111b, a third electrical interface 111c, a fourth electrical interface 111d, a fifth electrical interface 111e and a sixth electrical interface 111f.

An input of the first FDC circuit 201a is connected to the sixth electrical interface 111f, and an output of the first vibration mode of the fm gyro is connected to the input of the first FDC circuit 201a via the sixth electrical interface 111f.

An input of the second FDC circuit 201b is connected to the fifth electrical interface 111e, and an output of the second vibration mode of the fm gyro is connected to the input of the second FDC circuit 201b via the fifth electrical interface 111 e.

In addition, the output of the first vibration mode of the fm gyro is further connected to the input terminal of the first ADC circuit 202a through the sixth electrical interface 111f, and the first ADC circuit 202a is configured to convert an analog signal into a digital signal.

The output of the second vibration mode of the fm gyroscope is further coupled to the input of the second ADC circuit 202b via the fifth electrical interface 111e, and the second ADC circuit 202b is configured to convert an analog signal to a digital signal.

An output terminal of the first ADC circuit 202a is connected to an input terminal of the first IQ demodulation circuit 203 a. The output signal of the first ADC circuit 202a is IQ-demodulated in the first IQ demodulation circuit 203a, resulting in an in-phase signal and a quadrature signal.

The output terminal of the first IQ demodulation circuit 203a is connected to the input terminals of the first phase-locked loop circuit 206a and the first automatic gain control circuit 207a, respectively, so as to lock the resonance frequency point and control the excitation gain.

The output end of the second ADC circuit 202b is connected to the input end of the second IQ demodulation circuit 203 b; the output signal of the second ADC circuit 202b is IQ-demodulated in the second IQ demodulation circuit 203b, resulting in an in-phase signal and a quadrature signal.

The output terminal of the second IQ demodulation circuit 203b is connected to the input terminals of the second phase-locked loop circuit 206b and the second automatic gain control circuit 207b, respectively, so as to lock the resonance frequency point and control the excitation gain.

The output terminals of the first pll circuit 206a and the first agc circuit 207a are connected to the input terminal of the first digitally controlled oscillator 208a, and are capable of generating an excitation signal. The outputs of the second phase-locked loop circuit 206b and the second automatic gain control circuit 207b are connected to the input of the second numerically controlled oscillator 208b, which is capable of generating an excitation signal.

The output terminals of the first digitally controlled oscillator 208a and the second digitally controlled oscillator 208b are connected to the input terminals of the first multiplier 209a and the second multiplier 209b, respectively.

An output of the first multiplier 209a is connected to an input of a first low-pass filter 213a, and an output of the first low-pass filter 213a is connected to one input of the phase modulator 205.

An output of the second multiplier 209b is connected to an input of the second low-pass filter 213 b; the output of the second low pass filter 213b is connected to the other input of the phase modulator 205.

An output of the phase modulator 205 is connected to an input of the coherent demodulation circuit 204.

The sine and cosine signals generated by the first digitally controlled oscillator 208a and the second digitally controlled oscillator 208b are multiplied and filtered before being input to the phase modulator 205, and the output of the phase modulator 205 provides a reference signal for the coherent demodulation circuit 204.

In addition, the output terminal of the first numerically controlled oscillator 208a is further connected to the input terminal of the first error control circuit 211a, and the output terminal of the second numerically controlled oscillator 208b is further connected to the input terminal of the second error control circuit 211 b.

The output end of the first error control circuit 211a is connected to the fourth electrical interface 111d, and the tuning signal generated by the first error control circuit 211a is input to the first tuning electrode of the fm gyro through the fourth electrical interface 111 d.

The output end of the second error control circuit 211b is connected to the third electrical interface 111c, and the tuning signal generated by the second error control circuit 211b is input to the second tuning electrode of the fm gyro through the third electrical interface 111 c.

In addition, the output terminal of the first digitally controlled oscillator 208a is also connected to the input terminal of the first D AC circuit 212 a; an output of the first DAC circuit 212a is connected to the first electrical interface 111 a. The first D AC circuit 212a is configured to convert a digital signal into an analog signal, and then input the analog signal to the first driving input electrode of the fm gyro as a driving excitation signal of the first vibration mode.

The output of the second numerically controlled oscillator 208b is also connected to an input of a second DAC circuit 212b, the output of the second DAC circuit 212b being connected to the second electrical interface 111 b. The second D AC circuit 212b is configured to convert a digital signal into an analog signal, and then input the analog signal to the second driving input electrode of the fm gyro as a driving excitation signal of the second vibration mode.

The outputs of the first and second FDC circuits 201a, 201b are connected to inputs of an adder 210, respectively, the outputs of which are added in the adder, the output of the adder 210 being connected to an input of the coherent demodulation circuit 204.

An output terminal of the coherent demodulation circuit 204 is connected to an input terminal of the third low-pass filter 213c, and an output terminal of the third low-pass filter 213c is connected to the rate signal output interface 214, and filters and outputs the demodulated signal.

The trend of the signal flow in the frequency modulation gyro Lissajous modulation and self-correction test system is as follows:

1. the first FDC circuit 201a is connected to the sixth electrical interface 111f, and is configured to extract a frequency signal of the sensing output signal of the first sensing output electrode 110 a; the second FDC circuit 201b is connected to the fifth electrical interface 111e for extracting the frequency signal of the second sensing output signal 110b sensing output signal.

2. The first ADC circuit 202a is connected to the sixth electrical interface 111f, and is configured to perform analog-to-digital conversion on the sensed output signal of the first sensing output electrode 110a, i.e. convert the analog signal into a digital signal. The digital signal undergoes IQ demodulation by the first IQ demodulation circuit 203a, and the demodulated signal enters the first phase-locked loop circuit 206a and the first automatic gain control circuit 207a.

The second ADC circuit 202b is connected to the fifth electrical interface 111e, and is configured to perform analog-to-digital conversion on the sensed output signal of the second sensing output electrode 110b, i.e. convert the analog signal into a digital signal. The digital signal undergoes IQ demodulation by the second IQ demodulation circuit 203b, and the demodulated signal enters the second phase-locked loop circuit 206b and the second automatic gain control circuit 207b.

According to the embodiment of the invention, a phase-locked loop circuit is used for frequency tracking, the actual resonance frequency of the gyroscope is affected by the environment and the effect of residual stress release is achieved, if the gyroscope is driven by a fixed frequency, the driving frequency is inconsistent with the resonance frequency of the gyroscope, so that zero drift is generated by the gyroscope, the frequency of the numerical control oscillator is maintained at the respective tracking points by using the phase-locked loop to track the two-mode resonance frequency points, and the gyroscope always works in a resonance state, so that the gyroscope is ensured to have higher sensitivity and errors caused by the part are effectively reduced.

Meanwhile, the embodiment of the invention stably controls the amplitudes of two modes of the gyroscope through the automatic gain control circuit, and ensures the measurement stability and precision of the gyroscope. Due to the anisotropic property of the silicon material and the tolerance in the manufacturing process, the damping coefficient/quality factor of the two modes are different, and the damping mismatch error can cause serious nonlinear drift of the gyroscope; in the long-term running process, the damping coefficient can also generate slow change due to heating and stress release, so that the angular rate of the gyroscope can randomly walk, the gyroscope can be equivalently driven into two damping-matched modes by using an automatic gain control circuit, the amplitude instability in the long-term running process is counteracted, and the medium-long-term zero point stability of the gyroscope is finally effectively improved.

3. The output signals of the first phase-locked loop circuit 206a and the first automatic gain control circuit 207a enter the first digital controlled oscillator 208a to generate a two-axis digital signal sin (ω) _x t) and cos (omega) _x t). The output signals of the second phase-locked loop circuit 206b and the second automatic gain control circuit 207b enter the second digital oscillator 208b to generate a two-axis digital signal sin (omega) _y t) and cos (omega) _y t)。

4. The signals output by the first FDC circuit 201a and the second FDC circuit 201b are added, and enter the coherent demodulation circuit 204, and after the MEMS gyro signal is processed by the FDC circuit, the omega is output _z sin (Δωt).

In the formula, delta omega is continuously changed and the frequency is lower, harmonic distortion is very easy to occur in the traditional signal coherent demodulation process, so that the signal to noise ratio of the signal is reduced, and great challenges are brought to the improvement of zero point instability of the output angular rate of the gyroscope.

The embodiment of the invention can recover the accurate demodulation reference signal by utilizing the combination of the numerical control oscillator, the filter and the phase modulator.

5. The signal generated by the first digitally controlled oscillator 208a and the signal generated by the second digitally controlled oscillator 208b are multiplied to obtain:

after passing through the first low-pass filter and the second low-pass filter, the high-frequency component cos [ (omega) can be filtered out _x +ω _y )t]And sin [ (omega) _x +ω _y )t]The reserved component cos [ (omega) _x -ω _y )t]And sin [ (omega) _x -ω _y )t]Is a critical demodulation reference signal.

Since the low-pass filter introduces phase delay, the invention recovers demodulation reference signals cos (delta ωt) and sin (delta ωt) with accurate phase through the 32bit dual-channel phase modulator 205 designed by the FPGA, and considers the phase change caused by operation and transmission in an actual system, and the scheme also designs the phase modulator to ensure the quadrature/in-phase relation between the demodulation reference signals and the demodulated signals.

6. The demodulation signals cos (Δωt) and sin (Δωt) output by the phase modulator 205 enter the coherent demodulation circuit 204, and are coherently demodulated and filtered by using the accurate demodulation signals, so that an accurate rate output signal can be obtained;

the rate output signal obtained by coherent demodulation and filtering is output through the rate signal output port 214.

7. The output signal of the first numerically controlled oscillator 208a enters the first error control circuit 211a, and the tuning signal generated by the first error control circuit 211a is input to the first tuning electrode of the fm gyro through the fourth electrical interface 111 d.

The output signal of the second numerically controlled oscillator 208b enters the second error control circuit 211b, and the tuning signal generated by the second error control circuit 211b is input to the second tuning electrode of the fm gyro through the third electrical interface 111 c.

Frequency splitting is a critical parameter for precisely solving the angular rate of the frequency-modulated gyroscope, and the zero position of the gyroscope is directly determined. In reality, because of the nonideal factors such as temperature change, residual stress release and the like of the gyroscope tube core, the resonance frequencies of the two modes are constantly changed in the running state, and the change directions of the resonance frequencies are inconsistent due to the anisotropism of the monocrystalline silicon material. In other words, the key parameter of the gyro frequency difference Δω is drift with time, so that the zero point stability of the gyro, especially the middle-long term stability, is seriously affected.

In order to minimize the influence of frequency splitting on the system, the present embodiment proposes a dc tuning voltage control method. Under the Lissajous frequency modulation operation mode, the two modes of the gyroscope are driven by one digital control oscillator respectively, and the frequency splitting values of the two modes can be extracted in real time by matching with the respective phase-locked loop circuits. After the frequency splitting value is extracted, a tuning voltage may be input to the third electrical interface 111c and the fourth electrical interface 111d to dynamically tune the frequency difference between the XY two modes.

8. The output signal of the first digitally controlled oscillator 208a enters the first DAC circuit 212a, and the first DAC circuit 212a converts the digital signal into an analog signal and then enters the first electrical interface 111a, thereby completing the phase-locked loop and automatic gain control.

The output signal of the second numerically controlled oscillator 208b enters the second DAC circuit 212b, and the second DAC circuit 212b converts the digital signal into an analog signal and then enters the second electrical interface 111b, thereby completing the phase-locked loop and automatic gain control.

Compared with the prior art, the Lissajous modulation and self-correction test system in the embodiment has the following advantages:

(1) the invention adopts a frequency modulation method, utilizes the FDC circuit to extract the frequency signal, and solves the problem that the amplitude signal is easy to be interfered in the amplitude modulation scheme.

(2) The invention adopts the Lissajous frequency modulation method, can demodulate the rate signal without mode matching, and solves the problems of strong stability dependence of the normal quadrature frequency modulation on the resonance frequency point and extremely high requirement on circuit symmetry in the current frequency modulation system.

(3) The current Lissajous frequency modulation system is an open loop system and lacks a self-correction system.

According to the invention, the ADC auxiliary topological structure is adopted to enable the system to form a closed loop, drift in the mechanical system is counteracted by the phase-locked loop, the automatic gain control and the error correction loop, and the problem that the current open-loop operation of the frequency modulation system cannot automatically correct external interference is solved.

The closed loop is that an analog signal of a gyro system is converted into a digital signal by an ADC circuit, the digital signal is demodulated into an in-phase signal and a quadrature signal by an IQ demodulation circuit, a digital control oscillator is controlled by a phase-locked loop circuit and an automatic gain control circuit to generate an excitation signal, the excitation signal is converted into an analog signal by a DAC circuit and is fed back to an excitation input end of the gyro system, and a system excitation closed loop is formed; the excitation signal generates an error control signal through an error control circuit and is fed back to the tuning input end of the gyro system to form a system error control closed loop.

As shown in fig. 2, the fm gyro further includes a first electrical interface 111a, a second electrical interface 111b, a third electrical interface 111c, a fourth electrical interface 111d, a fifth electrical interface 111e, and a sixth electrical interface 111f.

The first electrical interface 111a is connected to the first driving input electrode 108a, and the first electrical interface 111a is configured to receive an input driving signal of the first vibration mode and input the input driving signal to the first driving input electrode 108a.

The second electrical interface 111b is connected to the second driving input electrode 108b, and the second electrical interface 111b is configured to receive an input driving signal of the second vibration mode and input the input driving signal to the second driving input electrode 108b.

The third electrical interface 111c is connected to the second tuning input electrode 109b, and the third electrical interface 111c is configured to receive a tuning voltage signal of the second vibration mode and input the tuning voltage signal to the second tuning input electrode 109b.

The fourth electrical interface 111d is connected to the first tuning input electrode 109a, and the fourth electrical interface 111d is configured to receive a tuning voltage signal of the first vibration mode and input the tuning voltage signal to the first tuning input electrode 109a.

The fifth electrical interface 111e is connected to the second sensing output electrode 110b, and the second vibration mode of the fm gyroscope is output to the fifth electrical interface 111e through the second sensing output electrode 110 b.

The sixth electrical interface 111f is connected to the first sensing output electrode 110a, and the first vibration mode output of the fm gyro is output to the sixth electrical interface 111f through the first sensing output electrode 110 a.

Because most of the existing frequency modulation gyroscopes are mechanical gyroscopes, the gyroscopes have the following problems in verification of a modulation system:

1. because the MEMS gyroscopes are different in types, production processes and even production batches, the problems and the errors introduced are different, the zero point of the MEMS gyroscopes is very sensitive to temperature change due to the mechanical characteristics of the MEMS gyroscopes, and the effect of the same circuit modulation system is very different for different gyroscopes, so that the conventional MEMS gyroscopes are used, and the effect of a verification modulation system is unreasonable.

2. Because the mechanical structure error of the MEMS cannot be expressed exactly, a plurality of electronic errors and mechanical errors which are introduced by a circuit in the existing Lissajous frequency modulation gyro are mixed together, the modal coupling effect is not clear, the error source which causes zero drift in an electronic system cannot be determined, and corresponding optimization is performed.

3. The current software simulation speed with larger usage amount is too slow, and the time required for one simulation is too long after the required precision is achieved, so that the quick test is inconvenient.

4. In the testing process, the mechanical structure of the gyroscope is fixed, so that the internal parameters of the gyroscope cannot be changed, the testing result is single, and the gyroscope can be replaced only to change the parameters of the tested object, so that the performance of the system is not conveniently tested by using a large number of experimental results.

Based on the principle, the embodiment of the invention also provides an MEMS gyroscope equivalent circuit which is a circuit which is built by using devices such as capacitance, inductance and resistance and completely simulates the internal working principle of the MEMS gyroscope, the mechanical error of the gyroscope is changed by controlling the value of the devices, and the two-mode coupling condition is simulated by an independent voltage source.

As shown in fig. 2 and 3, the MEMS gyroscope equivalent circuit includes a first vibration mode circuit, a second vibration mode circuit, and a coupling circuit between the first vibration mode circuit and the second vibration mode circuit.

The first vibration mode circuit comprises a first resistor 101a, a first capacitor 102a and a first inductor 103a, and the first resistor 101a, the first capacitor 102a and the first inductor 103a are sequentially connected in series to form an RLC resonant circuit.

The second vibration mode circuit includes a second resistor 101b, a second capacitor 102b, and a second inductor 103b, where the second resistor 101b, the second capacitor 102b, and the second inductor 103b are sequentially connected in series to form an RLC resonant circuit.

The coupling circuit is realized by a mutual inductor and an operational amplifier, and the tuning input electrode adopts an error control circuit input scheme for restoring the gyro function.

As shown in fig. 3, the coupling circuit includes an operational amplifier, a transformer, a VGA, a potentiometer, and the like.

The number of operational amplifiers is six, and the first operational amplifier 104a, the second operational amplifier 104b, the third operational amplifier 104c, the fourth operational amplifier 104d, the fifth operational amplifier 104e and the sixth operational amplifier 104f are respectively.

Sixteen transformers are provided, namely, a first transformer 105a, a second transformer 105b, a third transformer 105c, a fourth transformer 105d, a fifth transformer 105e, a sixth transformer 105f, a seventh transformer 105g, an eighth transformer 105h, a ninth transformer 105i, a tenth transformer 105j, an eleventh transformer 105k, a twelfth transformer 105l, a thirteenth transformer 105m, a fourteenth transformer 105n, a fifteenth transformer 105o and a sixteenth transformer 105p.

There are two VGAs, a first VGA106a and a second VGA106b.

Eight potentiometers are respectively a first potentiometer 107a, a second potentiometer 107b, a third potentiometer 107c, a fourth potentiometer 107d, a fifth potentiometer 107e, a sixth potentiometer 107f, a seventh potentiometer 107g and an eighth potentiometer 107h.

The input end of the first transformer 105a is connected with a first driving input electrode 108a for inputting an excitation signal, and the output end of the first transformer 105a is connected in series to a first vibration mode circuit.

The input end of the second transformer 105b is connected with a second driving input electrode 108b for inputting an excitation signal, and the output end of the second transformer 105b is connected in series to a second vibration mode circuit.

The non-inverting and inverting input terminals of the first operational amplifier 104a are respectively connected to one end of the first capacitor 102a, and the output terminal of the first operational amplifier 104a is connected to the input terminal of the twelfth transformer 105 l.

A third resistor 101c is connected between the first capacitor 102a and the non-inverting input terminal of the first operational amplifier 104a, and a fourth resistor 101d is connected between the first capacitor 102a and the non-inverting input terminal of the first operational amplifier 104 a.

The output end of the twelfth transformer 105l is connected in series to the second vibration mode circuit.

Since the two input terminals of the first operational amplifier 104a are connected to one end of the first capacitor 102a, respectively, the voltage across the first capacitor 102a can be amplified, which is equivalent to stiffness coupling.

The non-inverting and inverting input terminals of the sixth operational amplifier 104f are connected to one end of the second capacitor 102b, respectively, and the output terminal of the sixth operational amplifier 104f is connected to the input terminal of the sixth transformer 105 f.

A fifth resistor 101e is connected between the second capacitor 102b and the non-inverting input terminal of the second operational amplifier 104b, and a sixth resistor 101f is connected between the second capacitor 102b and the non-inverting input terminal of the second operational amplifier 104 b.

The output end of the sixth transformer 105f is connected in series to the first vibration mode circuit.

Since the two input terminals of the sixth operational amplifier 104f are connected to one end of the second capacitor 102b, respectively, the voltage across the second capacitor 102b can be amplified, which is equivalent to stiffness coupling.

The input end of the fifth transformer 105e is connected in series to the first vibration mode circuit, and the output end of the fifth transformer 105e is connected with the non-inverting input end and the inverting input end of the second operational amplifier 104b respectively. The output end of the second operational amplifier 104b is connected to the input end of the eleventh transformer 105k, and the output end of the eleventh transformer 105k is connected in series to the second vibration mode circuit.

Since the input end of the fifth transformer 105e is connected in series to the first vibration mode circuit and the output end is connected to the input end of the second operational amplifier 104, the current in the first vibration mode circuit can be amplified to be a voltage, which is equivalent to damping coupling.

The input end of the thirteenth transformer 105m is connected in series to the second vibration mode circuit, and the output end of the thirteenth transformer 105m is connected to the non-inverting input end and the inverting input end of the fifth operational amplifier 104e respectively. The output end of the fifth operational amplifier 104e is connected to the input end of the fourth transformer 105d, and the output end of the fourth transformer 105d is connected in series to the first vibration mode circuit.

Since the input end of the thirteenth transformer 105m is connected in series to the second vibration mode circuit and the output end is connected to the input end of the fifth operational amplifier 104e, the current in the second vibration mode circuit can be amplified to be a voltage, which is equivalent to damping coupling.

The input end of the seventh transformer 105g is connected in series to the first vibration mode circuit, and the output end of the seventh transformer 105g is connected with the non-inverting input end and the inverting input end of the third operational amplifier 104c respectively.

The output end of the third operational amplifier 104c is sequentially connected with the input ends of the first VGA106a and the tenth transformer 105j, and the output end of the tenth transformer 105j is connected in series to the second vibration mode circuit.

Since the input end of the seventh transformer 105g is connected in series to the first vibration mode circuit and the output ends thereof are respectively connected to the input ends of the third operational amplifier 104c, the current in the circuit can be amplified to a voltage, which is equivalent to an angular rate signal.

The input end of the fifteenth transformer 105o is connected in series with the second vibration mode circuit, and the output end of the fifteenth transformer 105o is connected with the non-inverting input end and the inverting input end of the fourth operational amplifier 104d respectively.

The output end of the fourth operational amplifier 104d is sequentially connected with the input ends of the second VGA106b and the third transformer 105c, and the output end of the third transformer 105c is connected in series to the first vibration mode circuit.

Since the input end of the fifteenth transformer 105o is connected in series to the second vibration mode circuit and the output ends thereof are respectively connected to the input ends of the fourth operational amplifier 104d, the current in the circuit can be amplified to a voltage, which is equivalent to an angular rate signal.

One end of the first potentiometer 107a is connected to the non-inverting input terminal of the first operational amplifier 104a, and the other end is grounded.

The second potentiometer 107b is connected between the negative input terminal and the output terminal of the first operational amplifier 104 a.

The third potentiometer 107c is connected between the negative input and the output of the second operational amplifier 104 b.

The fourth potentiometer 107d is connected between the negative input and the output of the third operational amplifier 104 c.

One end of the fifth potentiometer 107e is connected to the non-inverting input terminal of the sixth operational amplifier 104f, and the other end is grounded.

The sixth potentiometer 107f is connected between the negative input and the output of the sixth operational amplifier 104 f.

The seventh potentiometer 107g is connected between the negative input and the output of the fifth operational amplifier 104 e.

The eighth potentiometer 107h is connected between the negative input terminal and the output terminal of the fourth operational amplifier 104 d.

The first potentiometer 107a, the second potentiometer 107b, the third potentiometer 107c, the fourth potentiometer 107d, the fifth potentiometer 107e, the sixth potentiometer 107f, the seventh potentiometer 107g and the eighth potentiometer 107h are all high-precision digital potentiometers.

The output end of the eighth transformer 105h is connected in series to the first vibration mode circuit, and the input end of the eighth transformer 105h is connected with the first tuning input electrode 109a, so as to realize the tuning function of the first vibration mode circuit.

The output end of the fourteenth transformer 105n is connected in series to the second vibration mode circuit, and the input end of the fourteenth transformer 105n is connected with the second tuning input electrode 109b, so as to realize the tuning function of the second vibration mode circuit.

The input end of the ninth transformer 105i is connected in series to the first vibration mode circuit, and the output end of the ninth transformer 105i is connected with the first induction output electrode 110a, so that the circuit signal is output by the first vibration mode circuit.

The input end of the sixteenth transformer 105p is connected in series to the second vibration mode circuit, and the output end of the sixteenth transformer 105p is connected with the second induction output electrode 110b, so that the output circuit signal of the second vibration mode circuit is realized.

The trend of the signal flow in the MEMS gyroscope equivalent circuit is as follows:

1. the coriolis force of the first vibration mode circuit is input into a differential amplifying circuit built by a fourth operational amplifier 104d, the current obtained by a fifteenth transformer 105o in the second vibration mode circuit is amplified by 2L lambda times, the amplification factor is regulated by an eighth potentiometer 107h, and then omega is amplified by a second VGA106b _Z The amplification is equivalent to the angular velocity and then reverse-serially connected into the first vibration mode circuit through the third transformer 105c to achieve the amplification of-2L lambda.

2. The coriolis force of the second vibration mode circuit is input into a differential amplifying circuit built by a third operational amplifier 104c, the current obtained by the seventh transformer 105g in the first vibration mode circuit is amplified by 2L lambda times, the amplification factor is regulated by a fourth potentiometer 107d, and then omega is amplified by a first VGA106a _Z The amplification is equivalent to the angular velocity and then reverse-serially connected into the second vibration mode circuit through the tenth transformer 105j to achieve the amplification of-2L lambda.

3. The transimpedance amplifier with the damping coupling input of the first vibration mode circuit built by the fifth operational amplifier 104e will pass through the thirteenth mutualThe current in the second vibration mode circuit obtained by the sensor 105m is amplified to R _xy The voltage of the multiple (i.e., the damping coupling multiple of the second vibration mode to the first vibration mode), the amplification factor is controlled by the seventh potentiometer 107g, and then is serially connected into the first vibration mode circuit through the fourth transformer 105 d.

4. The transimpedance amplifier built by the second operational amplifier 104b amplifies the current in the first vibration mode circuit, which is derived by the fifth transformer 105e, to R _yx The voltage of the multiple (i.e., the damping coupling multiple of the first vibration mode to the second vibration mode), the amplification factor is controlled by the third potentiometer 107c, and then is serially connected into the second vibration mode circuit through the eleventh transformer 105 k.

5. The stiffness coupling input in the first vibration mode circuit is formed by amplifying the voltage at two ends of the second capacitor 102b in the second vibration mode circuit through a transimpedance amplifier built by a sixth operational amplifier 104fThe amplification factor (i.e., the stiffness coupling factor of the second vibration mode to the first vibration mode) is controlled by the fifth potentiometer 107e and the sixth potentiometer 107f (specifically, the amplification factor is changed by controlling the introduction resistance of the fifth potentiometer 107e and the sixth potentiometer 107 f), and then is connected in series into the first vibration mode circuit through the sixth transformer 105 f.

Wherein c _y Indicating the size, c, of the second capacitor 102b _yx Representing the magnitude of the coupling capacitance obtained during the derivation of the formula.

6. The stiffness coupling in the second vibration mode circuit amplifies the voltage across the first capacitor 102a in the first vibration mode circuit by a transimpedance amplifier built up by the first operational amplifier 104aThe magnification (i.e., the stiffness coupling of the first vibrational mode to the second vibrational mode) is controlled by the first potentiometer 107a and the second potentiometer 107b (specifically, by controlling the first potentiometer 107a and the second potentiometer The magnitude of the induced resistance of the transformer 107b changes the amplification factor) and then is serially connected into the second vibration mode circuit through the eleventh transformer 105 k.

Wherein c _x Representing the size, c, of the first capacitor 102a _xy Representing the magnitude of the coupling capacitance obtained during the derivation of the formula.

7. The tuning input in the first vibration mode circuit is input from the dc power supply of the first tuning input electrode 109a, and is connected in series to the first vibration mode circuit through the eighth transformer 105 h. The tuning input in the second vibration mode circuit is input from the dc power supply of the second tuning input electrode 109b, and is connected in series to the second vibration mode circuit through the fourteenth transformer 105 n.

The MEMS gyroscope equivalent circuit is used for carrying out quick test verification on the gyroscope measurement and control circuit. The invention can self-define the mechanical error of the equivalent gyroscope, verify the influence of each error on the system and facilitate the subsequent circuit debugging; the invention overcomes the defect of unstable parameters caused by the adoption of the entity gyroscope in the traditional debugging method; according to the invention, the diversified gyro model is obtained by changing the equivalent gyro circuit parameters, so that a large number of samples are obtained through experiments, and the research of the gyro error model is facilitated.

Compared with the existing mechanical MEMS gyroscope, the MEMS gyroscope equivalent circuit has the following advantages:

(1) The MEMS gyroscope equivalent circuit adopts a circuit to simulate the function of the MEMS gyroscope, is more stable and is not easily interfered by the outside compared with the mechanical structure of the MEMS gyroscope, and solves the problem that the MEMS gyroscope of the testing system has low stability and is easily affected by factors such as temperature and the like to cause the inaccuracy of the testing system.

(2) The MEMS gyroscope equivalent circuit changes parameters such as damping coupling, rigidity coupling, angular rate signals and the like according to requirements through amplifier multiple setting, and the damping coupling and the rigidity coupling are set to zero, so that the electronic error introduced by a subsequent circuit can be clearly seen, and the defect that the electronic error cannot be separated from the mechanical error during performance test of a gyroscope modulation system by using the MEMS gyroscope, and therefore correction cannot be well carried out is overcome.

(3) The running speed of the MEMS gyroscope equivalent circuit can obtain an output result within a few seconds after parameters are changed, and the software simulation needs a few hours or more if the software simulation needs high precision, so that the experiment time can be saved and the problem of low simulation speed can be solved by the method for testing the MEMS gyroscope equivalent circuit.

(4) According to the invention, the sizes of the resistor, the capacitor and the inductor in the first vibration mode circuit and the second vibration mode circuit are changed, so that the size of the resonant frequency can be changed, the diversity of the measured gyroscopes is increased, the defect that gyroscopes with different resonant frequencies need to be replaced when multiple resonant frequency data are required to be acquired in the test process is overcome, a large amount of experimental data can be obtained, and the diversity adaptability of the measurement and control circuit is better checked.

The foregoing description is, of course, merely illustrative of preferred embodiments of the present invention, and it should be understood that the present invention is not limited to the above-described embodiments, but is intended to cover all modifications, equivalents and alternatives falling within the spirit and scope of the present invention as defined by the appended claims.

Claims

1. The system is characterized by comprising an FDC circuit, an ADC circuit, an IQ demodulation circuit, a coherent demodulation circuit, a phase modulator, a phase-locked loop circuit, an automatic gain control circuit, a numerical control oscillator, a multiplier, an adder, an error control circuit, a DAC circuit, a low-pass filter and an electrical interface;

the FDC circuits are two, namely a first FDC circuit and a second FDC circuit;

the DAC circuits are two, namely a first DAC circuit and a second DAC circuit;

the output end of the first numerical control oscillator is connected with the input end of the first multiplier, the input end of the second multiplier, the input end of the first error control circuit and the input end of the first DAC circuit;

the output ends of the first error control circuit and the first DAC circuit are respectively connected with the fourth electrical interface and the first electrical interface;

the output end of the second digital control oscillator is connected with the input end of the first multiplier, the input end of the second error control circuit and the input end of the second DAC circuit;

the output ends of the second error control circuit and the second DAC circuit are respectively connected with the third electric interface and the second electric interface;

the output end of the first multiplier is sequentially connected with one input end of the first low-pass filter and one input end of the phase modulator; the output end of the second multiplier is sequentially connected with the other input ends of the second low-pass filter and the phase modulator;

The output end of the coherent demodulation circuit is connected with a third low-pass filter, and the third low-pass filter is connected with a rate signal output interface;

the frequency modulation gyroscope comprises a first driving input electrode, a second driving input electrode, a first tuning input electrode, a second tuning input electrode, a first induction output electrode and a second induction output electrode;

the first driving input electrode is connected with the first electric interface, and the second driving input electrode is connected with the second electric interface;

2. The fm gyro lissajous modulation and self-calibration test system of claim 1, wherein,

the frequency modulation gyroscope adopts an MEMS gyroscope equivalent circuit and comprises a first vibration mode circuit, a second vibration mode circuit and a coupling circuit positioned between the first vibration mode circuit and the second vibration mode circuit;

the two VGAs are respectively a first VGA and a second VGA;

the input end of the thirteenth transformer is connected in series with the second vibration mode circuit, and the output end of the thirteenth transformer is respectively connected with the non-inverting input end and the inverting input end of the fifth operational amplifier; the output end of the fifth operational amplifier is connected to the input end of the fourth transformer,