CN113514079A

CN113514079A - Frequency modulation gyro Lissajous modulation and self-correction test system

Info

Publication number: CN113514079A
Application number: CN202110318786.1A
Authority: CN
Inventors: 李崇; 刘志鹏; 孟相睿; 王雨晨; 侯佳坤; 王鑫宁
Original assignee: Ocean University of China
Current assignee: Ocean University of China
Priority date: 2021-03-25
Filing date: 2021-03-25
Publication date: 2021-10-19
Anticipated expiration: 2041-03-25
Also published as: CN113514079B

Abstract

The invention belongs to the technical field of MEMS (micro-electromechanical systems) gyros and discloses a Lissajous modulation and self-correction test system of a frequency modulation gyroscope, which designs an ADC (analog-to-digital converter) -assisted novel topological structure aiming at the defect that the existing Lissajous frequency modulation gyroscope is lack of a self-correction system. The invention utilizes the topological structure to carry out frequency tracking control and amplitude stabilization control, is a closed-loop operation mode, realizes high-stability control of vibration frequency and amplitude by separately resolving the driving signal and the angular rate signal, and inhibits zero drift error caused by resonance frequency change and damping mismatching.

Description

Frequency modulation gyro Lissajous modulation and self-correction test system

Technical Field

The invention belongs to the technical field of MEMS (micro-electromechanical systems) gyroscopes, and relates to a frequency modulation gyroscope Lissajous modulation and self-correction test system.

Background

In recent years, the MEMS gyroscope technology has been rapidly developed, and its application in the fields of industry, military and the like is more and more extensive due to its small size, low power consumption, low cost and the like, but its application in the advanced fields of high-performance tactical weapons, robots and the like is limited due to its poor zero stability. And the novel MEMS gyro modulation and self-correction technology is an effective way for improving the zero point stability of the MEMS gyro. The modulation mode of the MEMS gyroscope is divided into amplitude modulation and frequency modulation.

Most of MEMS gyro modulation and self-correction systems are amplitude modulation systems in the traditional scheme, but because of the characteristic that amplitude signals are easily interfered, even under a constant-temperature and temperature-control environment and various correction methods, the amplitude modulation gyro still has a very obvious zero drift problem. With the continuous progress of the technology, the frequency modulation gyro technology is developed, and compared with an amplitude signal, a frequency signal has ultrahigh stability and is not easily interfered by an external environment, so that various defects of the amplitude modulation gyro can be well overcome.

The frequency-modulated gyroscope is divided into orthogonal frequency modulation and Lissajous frequency modulation. Most of the prior frequency modulation gyro systems are orthogonal frequency modulation systems, and have the advantages of mode matching, unlimited bandwidth, reliable scale factor and the like, but still have stronger dependence on the stability of resonant frequency, and have extremely high requirements on a frequency measurement circuit, so that the application of the gyro system is very limited. The quadrature frequency modulation system has extremely strict frequency matching requirements and zero rate offset directly related to modal natural frequency drift, and theoretically cannot exert the advantages of a frequency modulation gyroscope. Lissajous frequency modulation well overcomes the defect of orthogonal frequency modulation, has no strict requirement on the symmetry of a gyroscope, and theoretically has the characteristics of insensitivity to temperature change, high stability, low hardware requirement and the like, so that better performance can be achieved.

However, the existing lissajous fm gyro system lacks a correction system made for mechanical errors, and mostly adopts an open-loop operation method, so that zero drift errors in the MEMS gyro mechanical system cannot be offset.

Disclosure of Invention

Aiming at the technical problems of the Lissajous FM gyro system in the prior art, the invention provides a Lissajous modulation and self-correction test system of the FM gyro, which adopts the following technical scheme:

a frequency modulation gyro Lissajous modulation and self-correction test system comprises an FDC circuit, an ADC circuit, an IQ demodulation circuit, a coherent demodulation circuit, a phase modulator, a phase-locked loop circuit, an automatic gain control circuit, a numerical control oscillator, a multiplier, an adder, an error control circuit, a DAC circuit, a low-pass filter and an electrical interface;

the two FDC circuits are respectively a first FDC circuit and a second FDC circuit;

the two ADC circuits are respectively a first ADC circuit and a second ADC circuit;

two IQ demodulation circuits are provided, namely a first IQ demodulation circuit and a second IQ demodulation circuit;

the coherent demodulation circuit and the phase modulator are respectively provided with one phase modulator;

the two phase-locked loop circuits are respectively a first phase-locked loop circuit and a second phase-locked loop circuit;

the two automatic gain control circuits are respectively a first automatic gain control circuit and a second automatic gain control circuit;

the two numerically-controlled oscillators are respectively a first numerically-controlled oscillator and a second numerically-controlled oscillator;

the number of the multipliers is two, namely a first multiplier and a second multiplier; one adder is provided;

the two error control circuits are respectively a first error control circuit and a second error control circuit;

the two DAC circuits are respectively a first DAC circuit and a second DAC circuit;

the number of the low-pass filters is three, and the three low-pass filters are respectively a first low-pass filter, a second low-pass filter and a third low-pass filter;

the number of the electrical interfaces is six, namely a first electrical interface, a second electrical interface, a third electrical interface, a fourth electrical interface, a fifth electrical interface and a sixth electrical interface;

the input end of the first FDC circuit and the input end of the first ADC circuit are connected with the sixth electrical interface, and the input end of the second FDC circuit and the input end of the second ADC circuit are connected with the fifth electrical interface;

the output end of the first ADC circuit is connected with the input end of the first IQ demodulation circuit; the output end of the first IQ demodulation circuit is respectively connected with the input ends of the first phase-locked loop circuit and the first automatic gain control circuit;

the output end of the second ADC circuit is connected with the input end of the second IQ demodulation circuit; the output end of the second IQ demodulation circuit is respectively connected with the input ends of the second phase-locked loop circuit and the second automatic gain control circuit;

the output ends of the first phase-locked loop circuit and the first automatic gain control circuit are connected with the input end of the first numerical control oscillator;

the output ends of the second phase-locked loop circuit and the second automatic gain control circuit are connected with the input end of the second numerical control oscillator;

the output end of the first digital controlled oscillator is respectively connected to the input end of the first multiplier, the input end of the second multiplier, the input end of the first error control circuit and the input end of the first DAC circuit; the output ends of the first error control circuit and the first DAC circuit are respectively connected with the fourth electrical interface and the first electrical interface;

the output end of the second digital controlled oscillator is connected with the input end of the first multiplier, the input end of the second error control circuit and the input end of the second DAC circuit; the output ends of the second error control circuit and the second DAC circuit are respectively connected with the third electrical interface and the second electrical interface;

the output end of the first multiplier is connected with the input ends of the first low-pass filter and the phase modulator in sequence; the output end of the second multiplier is sequentially connected with the input ends of the second low-pass filter and the phase modulator;

the output end of the phase modulator is connected with the input end of the coherent demodulation circuit; the output ends of the first FDC circuit and the second FDC circuit are respectively connected to the input end of the adder, and the output end of the adder is connected with the input end of the coherent demodulation circuit;

the output end of the coherent demodulation circuit is connected with a third low-pass filter, and the third low-pass filter is connected with a rate signal output interface.

Preferably, the frequency modulation gyroscope comprises a first driving input electrode, a second driving input electrode, a first tuning input electrode, a second tuning input electrode, a first sensing output electrode and a second sensing output electrode;

the first driving input electrode is connected with the first electrical interface, and the second driving input electrode is connected with the second electrical interface;

the first tuning input electrode is connected with the fourth electrical interface, and the second tuning input electrode is connected with the third electrical interface;

the first sensing output electrode is connected with the sixth electrical interface, and the second sensing output electrode is connected with the fifth electrical interface.

Preferably, the frequency modulation gyroscope adopts an MEMS gyroscope equivalent circuit, which includes a first vibration mode circuit, a second vibration mode circuit, and a coupling circuit located between the first vibration mode circuit and the second vibration mode circuit;

the first vibration mode circuit comprises a first resistor, a first capacitor and a first inductor which are sequentially connected in series;

the second vibration mode circuit comprises a second resistor, a second capacitor and a second inductor which are sequentially connected in series;

the coupling circuit comprises an operational amplifier, a mutual inductor, a VGA and a potentiometer;

six operational amplifiers are respectively a first operational amplifier, a second operational amplifier, a third operational amplifier, a fourth operational amplifier, a fifth operational amplifier and a sixth operational amplifier;

sixteen transformers are respectively a first transformer, a second transformer, a third transformer, a fourth transformer, a fifth transformer, a sixth transformer, a seventh transformer, an eighth transformer, a ninth transformer, a tenth transformer, an eleventh transformer, a twelfth transformer, a thirteenth transformer, a fourteenth transformer, a fifteenth transformer and a sixteenth transformer;

the two VGAs are respectively a first VGA and a second VGA;

eight potentiometers are provided, namely a first potentiometer, a second potentiometer, a third potentiometer, a fourth potentiometer, a fifth potentiometer, a sixth potentiometer, a seventh potentiometer and an eighth potentiometer;

the input end of the first mutual inductor is connected with a first driving input electrode, and the output end of the first mutual inductor is connected in series with a first vibration mode circuit;

the input end of the second mutual inductor is connected with a second driving input electrode, and the output end of the second mutual inductor is connected in series with a second vibration mode circuit;

the positive phase input end and the negative phase input end of the first operational amplifier are respectively connected to one end part of the first capacitor, and the output end of the first operational amplifier is connected to the input end of the twelfth transformer; the output end of the twelfth mutual inductor is connected in series with a second vibration mode circuit;

the positive phase input end and the negative phase input end of the sixth operational amplifier are respectively connected to one end part of the second capacitor, and the output end of the sixth operational amplifier is connected to the input end of the sixth mutual inductor; the output end of the sixth mutual inductor is connected in series with the first vibration mode circuit;

the input end of the fifth mutual inductor is connected in series with the first vibration mode circuit, and the output end of the fifth mutual inductor is respectively connected with the positive phase input end and the negative phase input end of the second operational amplifier; the output end of the second operational amplifier is connected to the input end of the eleventh mutual inductor;

the output end of the eleventh mutual inductor is connected in series with a second vibration mode circuit;

the input end of the thirteenth mutual inductor is connected with the second vibration mode circuit in series, and the output end of the thirteenth mutual inductor is connected with the positive phase input end and the negative phase input end of the fifth operational amplifier respectively; the output end of the fifth operational amplifier is connected to the input end of the fourth mutual inductor;

the output end of the fourth transformer is connected in series with the first vibration mode circuit;

the input end of the seventh mutual inductor is connected in series with the first vibration mode circuit, and the output end of the seventh mutual inductor is respectively connected with the positive phase input end and the negative phase input end of the third operational amplifier; the output end of the third operational amplifier is sequentially connected with the input ends of the first VGA and the tenth mutual inductor;

the output end of the tenth mutual inductor is connected in series with a second vibration mode circuit;

the input end of the fifteenth transformer is connected in series with the second vibration mode circuit, and the output end of the fifteenth transformer is connected with the positive phase input end and the negative phase input end of the fourth operational amplifier respectively; the output end of the fourth operational amplifier is sequentially connected with the second VGA and the input end of the third mutual inductor;

the output end of the third mutual inductor is connected in series with the first vibration mode circuit;

one end of the first potentiometer is connected with the positive phase input end of the first operational amplifier, and the other end of the first potentiometer is grounded;

the second potentiometer is connected between the negative phase input end of the first operational amplifier and the output end of the first operational amplifier;

the third potentiometer is connected between the negative phase input end of the second operational amplifier and the output end of the second operational amplifier;

the fourth potentiometer is connected between the negative phase input end of the third operational amplifier and the output end of the third operational amplifier;

one end of the fifth potentiometer is connected with the positive phase input end of the sixth operational amplifier, and the other end of the fifth potentiometer is grounded;

the sixth potentiometer is connected between the negative phase input end of the sixth operational amplifier and the output end of the sixth operational amplifier;

the seventh potentiometer is connected between the negative phase input end of the fifth operational amplifier and the output end of the fifth operational amplifier;

the eighth potentiometer is connected between the negative phase input end of the fourth operational amplifier and the output end of the fourth operational amplifier;

the input end of the eighth mutual inductor is connected with the first tuning input electrode, and the output end of the eighth mutual inductor is connected in series with the first vibration mode circuit; the input end of the fourteenth mutual inductor is connected with the second tuning input electrode, and the output end of the fourteenth mutual inductor is connected in series with the second vibration mode circuit;

the input end of the ninth mutual inductor is connected with the first vibration mode circuit in series, and the output end of the ninth mutual inductor is connected with the first induction output electrode; and the input end of the sixteenth mutual inductor is connected in series with the second vibration mode circuit, and the output end of the sixteenth mutual inductor is connected with the second induction output electrode.

The invention has the following advantages:

as described above, the invention provides a frequency modulation gyro Lissajous modulation and self-correction test system, aiming at the defect that the existing Lissajous frequency modulation gyro lacks a self-correction system, an ADC (analog to digital converter) auxiliary novel topological structure is designed for frequency tracking control and amplitude stabilization control, and the topological structure is a closed-loop operation mode and is used for resolving a driving signal and an angular rate signal separately, so that high-stability control of vibration frequency and amplitude is realized, and zero drift errors caused by mismatching of resonance frequency change and damping are suppressed.

Drawings

FIG. 1 is a schematic structural diagram of a FM gyrosaccha modulation and self-calibration test system according to an embodiment of the present invention;

FIG. 2 is a block diagram of an equivalent circuit of an MEMS gyroscope according to an embodiment of the present invention;

FIG. 3 is a schematic structural diagram of an MEMS gyroscope equivalent circuit in the embodiment of the present invention.

101 a-a first resistor, 101 b-a second resistor, 101 c-a third resistor, 101 d-a fourth resistor, 101 e-a fifth resistor, and 101 f-a sixth resistor; 102 a-a first capacitor, 102 b-a second capacitor, 103 a-a first inductor, 103 b-a second inductor; 104 a-a first operational amplifier, 104 b-a second operational amplifier, 104 c-a third operational amplifier, 104 d-a fourth operational amplifier, 104 e-a fifth operational amplifier, 104 f-a sixth operational amplifier; 105 a-a first transformer, 105 b-a second transformer, 105 c-a third transformer, 105 d-a fourth transformer, 105 e-a fifth transformer, 105 f-a sixth transformer, 105 g-a seventh transformer, 105 h-an eighth transformer, 105 i-a ninth transformer, 105 j-a tenth transformer, 105 k-an eleventh transformer, 105 l-a twelfth transformer, 105 m-a thirteenth transformer, 105 n-a fourteenth transformer, 105 o-a fifteenth transformer, 105 p-a sixteenth transformer; 106 a-a first VGA, 106 b-a second VGA; 107 a-a first potentiometer, 107 b-a second potentiometer, 107 c-a third potentiometer, 107 d-a fourth potentiometer, 107 e-a fifth potentiometer, 107 f-a sixth potentiometer, 107 g-a seventh potentiometer, 107 h-an eighth potentiometer; 108 a-a first driving input electrode, 108 b-a second driving input electrode, 109 a-a first tuning input electrode, 109 b-a second tuning input electrode, 110 a-a first sensing output electrode, 110 b-a second sensing output electrode; 111 a-a first electrical interface, 111 b-a second electrical interface, 111 c-a third electrical interface, 111 d-a fourth electrical interface, 111 e-a fifth electrical interface, 111 f-a sixth electrical interface; 201 a-a first FDC circuit, 201 b-a second FDC circuit, 202 a-a first ADC circuit, 202 b-a second ADC circuit, 203 a-a first IQ demodulation circuit, 203 b-a second IQ demodulation circuit, 204-a coherent demodulation circuit, 205-a phase modulator, 206 a-a first phase locked loop circuit, 206 b-a second phase locked loop circuit, 207 a-a first automatic gain control circuit, 207 b-a second automatic gain control circuit, 208 a-a first digitally controlled oscillator, 208 b-a second digitally controlled oscillator, 209 a-a first multiplier, 209 b-a second multiplier, 210-an adder, 211 a-a first error control circuit, 211 b-a second error control circuit, 212 a-a first DAC circuit, 212 b-a second DAC circuit, 213 a-a first low pass filter, 213 b-second low pass filter, 213 c-third low pass filter, 214-rate signal output interface.

Detailed Description

The invention is described in further detail below with reference to the following figures and detailed description:

examples

As shown in fig. 1, the present invention relates to a frequency modulation gyro lissajous modulation and self-calibration test system, which comprises an FDC circuit, an ADC circuit, an IQ demodulation circuit, a coherent demodulation circuit, a phase modulator, a phase locked loop circuit, an automatic gain control circuit, a digitally controlled oscillator, a multiplier, an adder, an error control circuit, a DAC circuit, a low-pass filter and an electrical interface.

The FDC circuit, the ADC circuit, the IQ demodulation circuit, the coherent demodulation circuit, the phase modulator, the phase-locked loop circuit, the automatic gain control circuit, the IQ demodulation circuit, the error control circuit, the DAC circuit and the like can adopt mature schemes.

There are two FDC circuits, a first FDC circuit 201a and a second FDC circuit 201 b.

There are two ADC circuits, a first ADC circuit 202a and a second ADC circuit 202 b.

There are two IQ demodulation circuits, namely a first IQ demodulation circuit 203a and a second IQ demodulation circuit 203 b.

The coherent demodulation circuit 204 and the phase modulator 205 are each provided with one.

There are two pll circuits, a first pll circuit 206a and a second pll circuit 206 b.

The automatic gain control circuit includes a first automatic gain control circuit 207a and a second automatic gain control circuit 207 b.

The two dcgs are a first dcg 208a and a second dcg 208 b.

There are two multipliers, a first multiplier 209a and a second multiplier 209 b; there is one adder 210.

There are two error control circuits, a first error control circuit 211a and a second error control circuit 211 b.

There are two DAC circuits, a first DAC circuit 212a and a second DAC circuit 212 b.

The low pass filters include a first low pass filter 213a, a second low pass filter 213b, and a third low pass filter 213 c.

There are six electrical interfaces, which are the first electrical interface 111a, the second electrical interface 111b, the third electrical interface 111c, the fourth electrical interface 111d, the fifth electrical interface 111e, and the sixth electrical interface 111 f.

The input end of the first FDC circuit 201a is connected to the sixth electrical interface 111f, and the output of the first vibration mode of the fm gyroscope is connected to the input end of the first FDC circuit 201a through the sixth electrical interface 111 f.

The input end of the second FDC circuit 201b is connected to the fifth electrical interface 111e, and the output of the second vibration mode of the fm gyro is connected to the input end of the second FDC circuit 201b through the fifth electrical interface 111 e.

In addition, the output of the first vibration mode of the fm gyroscope is further connected to the input terminal of the first ADC circuit 202a through the sixth electrical interface 111f, and the first ADC circuit 202a is configured to convert the analog signal into a digital signal.

The output of the second vibration mode of the fm gyroscope is further connected to the input of a second ADC circuit 202b through a fifth electrical interface 111e, and the second ADC circuit 202b is configured to convert the analog signal into a digital signal.

An output terminal of the first ADC circuit 202a is connected to an input terminal of the first IQ demodulation circuit 203 a. The output signal of the first ADC circuit 202a is IQ-demodulated in the first IQ demodulation circuit 203a to obtain an in-phase signal and a quadrature signal.

The output terminal of the first IQ demodulation circuit 203a is connected to the input terminals of the first phase-locked loop circuit 206a and the first automatic gain control circuit 207a, respectively, and is capable of locking the resonance frequency point and controlling the excitation gain.

The output end of the second ADC circuit 202b is connected to the input end of the second IQ demodulation circuit 203 b; the output signal of the second ADC circuit 202b is IQ-demodulated in the second IQ demodulation circuit 203b to obtain an in-phase signal and a quadrature signal.

The output terminal of the second IQ demodulation circuit 203b is connected to the input terminals of the second phase-locked loop circuit 206b and the second automatic gain control circuit 207b, respectively, and is capable of locking the resonance frequency point and controlling the excitation gain.

The outputs of the first phase locked loop circuit 206a and the first automatic gain control circuit 207a are coupled to the input of a first digitally controlled oscillator 208a, which is capable of generating an excitation signal. The outputs of the second phase locked loop circuit 206b and the second automatic gain control circuit 207b are connected to the input of a second numerically controlled oscillator 208b capable of generating an excitation signal.

The outputs of the first and second numerically controlled

oscillators

208a, 208b are connected to the inputs of first and

second multipliers

209a, 209b, respectively.

The output of the first multiplier 209a is coupled to an input of a first low-pass filter 213a, the output of the first low-pass filter 213a being coupled to an input of the phase modulator 205.

The output of the second multiplier 209b is connected to the input of a second low-pass filter 213 b; the output of second low pass filter 213b is connected to the other input of phase modulator 205.

The output of phase modulator 205 is connected to the input of coherent demodulation circuit 204.

The sine and cosine signals generated by the first and second digitally controlled

oscillators

208a and 208b are multiplied and filtered before being input to the phase modulator 205, and the output of the phase modulator 205 provides a reference signal for the coherent demodulation circuit 204.

Furthermore, the output of the first digitally controlled oscillator 208a is coupled to an input of a first error control circuit 211a, and the output of the second digitally controlled oscillator 208b is coupled to an input of a second error control circuit 211 b.

The output end of the first error control circuit 211a is connected to the fourth electrical interface 111d, and the tuning signal generated by the first error control circuit 211a is input to the first tuning electrode of the fm gyroscope through the fourth electrical interface 111 d.

The output end of the second error control circuit 211b is connected to the third electrical interface 111c, and the tuning signal generated by the second error control circuit 211b is input to the second tuning electrode of the fm gyroscope through the third electrical interface 111 c.

Furthermore, the output of the first digitally controlled oscillator 208a is also connected to the input of the first D AC circuit 212 a; the output of the first DAC circuit 212a is connected to the first electrical interface 111 a. The first D-AC circuit 212a is configured to convert the digital signal into an analog signal, and then input the analog signal to the first driving input electrode of the fm gyro as a driving excitation signal of the first vibration mode.

The output of the second numerically controlled oscillator 208b is also connected to an input of a second DAC circuit 212b, the output of the second DAC circuit 212b being connected to the second electrical interface 111 b. The second D-AC circuit 212b is configured to convert the digital signal into an analog signal, and then input the analog signal to the second driving input electrode of the fm gyro as a driving excitation signal of the second vibration mode.

The outputs of the first FDC circuit 201a and the second FDC circuit 201b are connected to the inputs of an adder 210, respectively, and their outputs are added in the adder 210, and the output of the adder 210 is connected to the input of the coherent demodulation circuit 204.

The output end of the coherent demodulation circuit 204 is connected to the input end of the third low-pass filter 213c, and the output end of the third low-pass filter 213c is connected to the rate signal output interface 214, so as to filter and output the demodulated signal.

The trend of the signal flow in the frequency modulation gyro Lissajous modulation and self-correction test system is as follows:

1. the first FDC circuit 201a is connected to the sixth electrical interface 111f, and is configured to extract a frequency signal of an induction output signal of the first induction output electrode 110 a; the second FDC circuit 201b is connected to the fifth electrical interface 111e, and is configured to extract a frequency signal of the sensing output signal of the second sensing output electrode 110 b.

2. The first ADC circuit 202a is connected to the sixth electrical interface 111f, and is configured to perform analog-to-digital conversion on the sensing output signal of the first sensing output electrode 110a, i.e. convert the analog signal into a digital signal. The digital signal is IQ-demodulated by the first IQ demodulation circuit 203a, and the demodulated signal enters the first phase-locked loop circuit 206a and the first automatic gain control circuit 207 a.

The second ADC circuit 202b is connected to the fifth electrical interface 111e, and is configured to perform analog-to-digital conversion on the sensing output signal of the second sensing output electrode 110b, i.e. convert the analog signal into a digital signal. The digital signal is IQ-demodulated by the second IQ demodulation circuit 203b, and the demodulated signal enters the second phase-locked loop circuit 206b and the second automatic gain control circuit 207 b.

In the embodiment of the invention, the frequency tracking uses the phase-locked loop circuit, the actual resonance frequency of the gyroscope is influenced by the environment and the residual stress is released, the actual resonance frequency of the gyroscope is constantly changed, if the gyroscope is driven by using a fixed frequency, the driving frequency is inconsistent with the resonance frequency of the gyroscope, so that the gyroscope generates zero drift, the frequency of the numerical control oscillator is maintained at each tracking point by using the phase-locked loop to track the two-mode resonance frequency points, and the gyroscope always works in a resonance state, so that the gyroscope is ensured to have higher sensitivity and the error caused by the parts is effectively reduced.

Meanwhile, the amplitude of two modes of the gyroscope is stably controlled through the automatic gain control circuit, so that the measurement stability and precision of the gyroscope are ensured. Due to the anisotropic characteristic of silicon materials and the tolerance in the manufacturing process, the damping coefficients/quality factors of two modes are different, and the mismatch error of the damping can cause the serious nonlinear drift of the gyroscope; in the long-term operation process, the damping coefficient can also generate slow change due to heating and stress release, so that the angular rate of the gyroscope can randomly walk, the gyroscope can be equivalently driven into two damping matching modes by using the automatic gain control circuit, the amplitude instability in the long-term operation can be counteracted, and the medium-term and long-term zero point stability of the gyroscope is finally effectively improved.

3. The first PLL circuit 206a and the first AGC circuit 207a output signals to a first VCO 208a to generate a two-axis digital sin (ω) signal_xt) and cos (. omega.) of_xt). The second PLL circuit 206b and the second AGC circuit 207b output signals to a second digital oscillator 208b to generate a two-axis digital signal sin (ω)_yt) and cos (. omega.) of_yt)。

4. The signals output by the first FDC circuit 201a and the second FDC circuit 201b are added and enter the coherent demodulation circuit 204, and the MEMS gyro signal is processed by the FDC circuit and then outputs omega_zsin (Δ ω t).

In the formula, Δ ω changes continuously and has a low frequency, and harmonic distortion is easily generated in the process of coherent demodulation of a traditional signal, so that the signal-to-noise ratio of the signal is reduced, and great challenge is brought to improvement of zero instability of the output angular rate of the gyroscope.

The embodiment of the invention can recover the accurate demodulation reference signal by using the combination of the numerical control oscillator, the filter and the phase modulator.

5. The signal generated by the first digitally controlled oscillator 208a and the signal generated by the second digitally controlled oscillator 208b are multiplied to obtain:

the high frequency component cos [ (omega) can be filtered after passing through the first low pass filter and the second low pass filter_x+ω_y)t]And sin [ (omega)_x+ω_y)t]The remaining component cos [ (omega [ #]_x-ω_y)t]And sin [ (omega)_x-ω_y)t]The reference signal is demodulated for the key.

Because a low-pass filter can introduce phase delay, the invention recovers demodulation reference signals cos (delta omega t) and sin (delta omega t) with accurate phases through a 32-bit dual-channel phase modulator 205 designed by FPGA, and considers phase change caused by operation and transmission in an actual system, and the scheme also designs a phase modulator to ensure the orthogonal/in-phase relation of the demodulation reference signals and the demodulated signals.

6. Demodulated signals cos (Δ ω t) and sin (Δ ω t) output by the phase modulator 205 enter the coherent demodulation circuit 204, and coherent demodulation and filtering are performed by using the accurate demodulated signals, so that accurate rate output signals can be obtained;

the rate output signal obtained by coherent demodulation and filtering is output via the rate signal output port 214.

7. The output signal of the first digitally controlled oscillator 208a enters the first error control circuit 211a, and the tuning signal generated by the first error control circuit 211a is input to the first tuning electrode of the fm gyro through the fourth electrical interface 111 d.

The output signal of the second digital controlled oscillator 208b enters the second error control circuit 211b, and the tuning signal generated by the second error control circuit 211b is input to the second tuning electrode of the fm gyro through the third electrical interface 111 c.

The frequency cracking is a key parameter for accurately resolving the angular rate of the frequency modulation gyroscope, and directly determines the zero position of the gyroscope. In reality, because the gyroscope tube core has undesirable factors such as temperature change, residual stress release and the like, the resonant frequencies of the two modes are constantly changed in the running state, and the change directions of the resonant frequencies are inconsistent due to the anisotropy of the monocrystalline silicon material. In other words, the gyro frequency difference Δ ω is a key parameter, and is shifted with time, thereby seriously affecting the zero-point stability, especially the stability of the middle and long periods of the gyro.

In order to minimize the influence of frequency cracking on the system, the present embodiment proposes a dc tuning voltage control method. Under the Lissajous frequency modulation operation mode, two modes of the gyroscope are respectively driven by a numerical control oscillator and matched with respective phase-locked loop circuits, so that frequency cracking numerical values of the two modes can be extracted in real time. After extracting the frequency-resolved values, the tuning voltage may be input to the third electrical interface 111c and the fourth electrical interface 111d to dynamically tune the frequency difference between the two XY modes.

8. The output signal of the first digitally controlled oscillator 208a enters the first DAC circuit 212a, and the first DAC circuit 212a converts the digital signal into an analog signal and then enters the first electrical interface 111a, thereby completing the phase-locked loop and the automatic gain control.

The output signal of the second digital controlled oscillator 208b enters the second DAC circuit 212b, and the second DAC circuit 212b converts the digital signal into an analog signal and then enters the second electrical interface 111b, thereby completing the phase-locked loop and the automatic gain control.

Compared with the prior art, the Lissajous modulation and self-correction test system in the embodiment has the following advantages:

the invention adopts a frequency modulation method and utilizes the FDC circuit to extract frequency signals, thereby solving the problem that amplitude signals are easy to interfere in an amplitude modulation scheme.

Secondly, the invention adopts a Lissajous frequency modulation method, can demodulate rate signals without mode matching, and solves the problems that the commonly used orthogonal frequency modulation in the current frequency modulation system has strong stability dependence on resonance frequency points and extremely high requirement on circuit symmetry.

The current Lissajous frequency modulation system is an open loop system and lacks a self-correcting system.

According to the invention, the system forms a closed loop through an ADC (analog to digital converter) auxiliary topological structure, and the drift in the mechanical system is counteracted through a phase-locked loop, automatic gain control and an error correction loop, so that the problem that the external interference cannot be corrected by self during open-loop operation of the current frequency modulation system is solved.

The closed loop is that an analog signal of a gyro system is converted into a digital signal by an ADC (analog to digital converter) circuit, the signal is demodulated into an in-phase signal and a quadrature signal by an IQ (in-phase and quadrature) demodulation circuit, a numerically-controlled oscillator is controlled by a phase-locked loop circuit and an automatic gain control circuit to generate an excitation signal, the excitation signal is converted into an analog signal by a DAC (digital to analog converter) circuit and is fed back to an excitation input end of the gyro system to form a system excitation closed loop; the excitation signal generates an error control signal through an error control circuit and feeds the error control signal back to the tuning input end of the gyro system to form a system error control closed loop.

As shown in fig. 2, the fm gyro further includes a first electrical interface 111a, a second electrical interface 111b, a third electrical interface 111c, a fourth electrical interface 111d, a fifth electrical interface 111e, and a sixth electrical interface 111 f.

The first electrical interface 111a is connected to the first driving input electrode 108a, and the first electrical interface 111a is configured to receive an input driving signal in the first vibration mode and input the input driving signal to the first driving input electrode 108 a.

The second electrical interface 111b is connected to the second driving input electrode 108b, and the second electrical interface 111b is configured to receive an input driving signal of the second vibration mode and input the input driving signal to the second driving input electrode 108 b.

The third electrical interface 111c is connected to the second tuning input electrode 109b, and the third electrical interface 111c is configured to receive the tuning voltage signal of the second vibration mode and input the tuning voltage signal to the second tuning input electrode 109 b.

The fourth electrical interface 111d is connected to the first tuning input electrode 109a, and the fourth electrical interface 111d is configured to receive the tuning voltage signal of the first vibration mode and input the tuning voltage signal to the first tuning input electrode 109 a.

The fifth electrical interface 111e is connected to the second sensing output electrode 110b, and the second vibration mode of the fm gyroscope is output to the fifth electrical interface 111e through the second sensing output electrode 110 b.

The sixth electrical interface 111f is connected to the first sensing output electrode 110a, and the first vibration mode output of the fm gyroscope is output to the sixth electrical interface 111f through the first sensing output electrode 110 a.

Because most of the existing frequency modulation gyros are mechanical gyros, the gyros have the following problems when a modulation system is verified:

1. due to the fact that the MEMS gyroscopes are different in types and production processes, even different in production batches, the problems and introduced errors are different, the zero point of the MEMS gyroscope is very sensitive to temperature change due to the mechanical characteristics of the MEMS gyroscope, the effect of the same circuit modulation system on different gyroscopes is very different, and therefore the effect of the modulation system is not reasonable when the conventional MEMS gyroscope is used.

2. Because the MEMS mechanical structure error cannot be accurately represented, many electronic errors and mechanical errors introduced by a circuit in the existing Lissajous FM gyroscope are mixed together, the modal coupling effect is not clear, and the error source causing zero drift in an electronic system cannot be determined and optimized correspondingly.

3. At present, the software with larger use amount has too low simulation speed, and the simulation time is too long for one time after the required precision is reached, so that the rapid test is inconvenient.

4. In the testing process, because the mechanical structure of the gyroscope is fixed, the internal parameters of the gyroscope cannot be changed, the testing result is simplified, the gyroscope can only be replaced to change the parameters of the tested object, and the performance of the system is not convenient to be checked by using a large number of experimental results.

Based on the above, the embodiment of the invention further provides an MEMS gyroscope equivalent circuit, the MEMS gyroscope equivalent circuit is a circuit which is built by using devices such as capacitors, inductors and resistors and completely simulates the internal working principle of the MEMS gyroscope, the mechanical error of the gyroscope is changed by controlling the value of the devices, and the two-mode coupling condition is simulated by an independent voltage source.

As shown in fig. 2 and 3, the MEMS gyro equivalent circuit includes a first vibration mode circuit, a second vibration mode circuit, and a coupling circuit located between the first vibration mode circuit and the second vibration mode circuit.

The first vibration mode circuit comprises a first resistor 101a, a first capacitor 102a and a first inductor 103a, wherein the first resistor 101a, the first capacitor 102a and the first inductor 103a are sequentially connected in series to form an RLC resonance circuit.

The second vibration mode circuit comprises a second resistor 101b, a second capacitor 102b and a second inductor 103b, and the second resistor 101b, the second capacitor 102b and the second inductor 103b are sequentially connected in series to form an RLC resonance circuit.

The coupling circuit is realized by a mutual inductor and an operational amplifier, and an error control circuit is adopted for inputting a tuning input electrode for restoring the function of the gyroscope.

As shown in fig. 3, the coupling circuit includes an operational amplifier, a transformer, a VGA, and a potentiometer.

Six operational amplifiers are provided, namely a first operational amplifier 104a, a second operational amplifier 104b, a third operational amplifier 104c, a fourth operational amplifier 104d, a fifth operational amplifier 104e and a sixth operational amplifier 104 f.

There are sixteen transformers, which are respectively a first transformer 105a, a second transformer 105b, a third transformer 105c, a fourth transformer 105d, a fifth transformer 105e, a sixth transformer 105f, a seventh transformer 105g, an eighth transformer 105h, a ninth transformer 105i, a tenth transformer 105j, an eleventh transformer 105k, a twelfth transformer 105l, a thirteenth transformer 105m, a fourteenth transformer 105n, a fifteenth transformer 105o, and a sixteenth transformer 105 p.

There are two VGAs, a first VGA106a and a second VGA106 b.

The number of the potentiometers is eight, and the potentiometers are respectively a first potentiometer 107a, a second potentiometer 107b, a third potentiometer 107c, a fourth potentiometer 107d, a fifth potentiometer 107e, a sixth potentiometer 107f, a seventh potentiometer 107g and an eighth potentiometer 107 h.

The input end of the first transformer 105a is connected with a first driving input electrode 108a for inputting an excitation signal, and the output end of the first transformer 105a is connected in series to the first vibration mode circuit.

The input end of the second transformer 105b is connected with a second driving input electrode 108b for inputting an excitation signal, and the output end of the second transformer 105b is connected in series to the second vibration mode circuit.

The non-inverting and inverting input terminals of the first operational amplifier 104a are connected to one end of the first capacitor 102a, respectively, and the output terminal of the first operational amplifier 104a is connected to the input terminal of the twelfth transformer 105 l.

A third resistor 101c is connected between the first capacitor 102a and the positive-phase input terminal of the first operational amplifier 104a, and a fourth resistor 101d is connected between the first capacitor 102a and the negative-phase input terminal of the first operational amplifier 104 a.

The output end of the twelfth transformer 105l is connected in series to the second vibration mode circuit.

Since the two input terminals of the first operational amplifier 104a are respectively connected to one end of the first capacitor 102a, the voltage across the first capacitor 102a can be amplified, which is equivalent to rigid coupling.

The non-inverting and inverting input terminals of the sixth operational amplifier 104f are connected to one end of the second capacitor 102b, respectively, and the output terminal of the sixth operational amplifier 104f is connected to the input terminal of the sixth transformer 105 f.

A fifth resistor 101e is connected between the second capacitor 102b and the positive phase input terminal of the second operational amplifier 104b, and a sixth resistor 101f is connected between the second capacitor 102b and the negative phase input terminal of the second operational amplifier 104 b.

The output end of the sixth transformer 105f is connected in series to the first vibration mode circuit.

Since the two input terminals of the sixth operational amplifier 104f are respectively connected to one end of the second capacitor 102b, the voltage across the second capacitor 102b can be amplified, which is equivalent to rigid coupling.

The input end of the fifth transformer 105e is connected in series to the first vibration mode circuit, and the output end of the fifth transformer 105e is connected to the positive phase input end and the negative phase input end of the second operational amplifier 104b, respectively. The output end of the second operational amplifier 104b is connected to the input end of the eleventh transformer 105k, and the output end of the eleventh transformer 105k is connected in series to the second vibration mode circuit.

Since the input end of the fifth transformer 105e is connected in series to the first vibration mode circuit, and the output end is connected to the input end of the second operational amplifier 104, the current in the first vibration mode circuit can be amplified to be a voltage, which is equivalent to damping coupling.

The input end of the thirteenth transformer 105m is connected in series to the second vibration mode circuit, and the output end of the thirteenth transformer 105m is connected to the positive phase input end and the negative phase input end of the fifth operational amplifier 104e, respectively. The output end of the fifth operational amplifier 104e is connected to the input end of the fourth transformer 105d, and the output end of the fourth transformer 105d is connected in series to the first vibration mode circuit.

Since the input end of the thirteenth transformer 105m is connected in series to the second vibration mode circuit, and the output end is connected to the input end of the fifth operational amplifier 104e, the current in the second vibration mode circuit can be amplified to be a voltage, which is equivalent to damping coupling.

The input end of the seventh transformer 105g is connected in series to the first vibration mode circuit, and the output end of the seventh transformer 105g is connected to the positive phase input end and the negative phase input end of the third operational amplifier 104c, respectively.

The output end of the third operational amplifier 104c is connected to the first VGA106a and the input end of the tenth transformer 105j in sequence, and the output end of the tenth transformer 105j is connected to the second vibration mode circuit in series.

Since the input end of the seventh transformer 105g is connected in series to the first vibration mode circuit, and the output end is connected to the input end of the third operational amplifier 104c, the current in the circuit can be amplified to be a voltage, which is equivalent to an angular rate signal.

The input end of the fifteenth transformer 105o is connected in series to the second vibration mode circuit, and the output end of the fifteenth transformer 105o is connected to the positive phase input end and the negative phase input end of the fourth operational amplifier 104d, respectively.

The output end of the fourth operational amplifier 104d is connected to the input ends of the second VGA106b and the third transformer 105c in sequence, and the output end of the third transformer 105c is connected to the first vibration mode circuit in series.

Since the input end of the fifteenth transformer 105o is connected in series to the second vibration mode circuit, and the output end is connected to the input end of the fourth operational amplifier 104d, the current in the circuit can be amplified to be a voltage, which is equivalent to an angular rate signal.

The first potentiometer 107a has one terminal connected to the non-inverting input terminal of the first operational amplifier 104a and the other terminal connected to ground.

The second potentiometer 107b is connected between the negative phase input terminal and the output terminal of the first operational amplifier 104 a.

The third potentiometer 107c is connected between the negative phase input terminal and the output terminal of the second operational amplifier 104 b.

The fourth potentiometer 107d is connected between the negative phase input terminal and the output terminal of the third operational amplifier 104 c.

One end of the fifth potentiometer 107e is connected to the non-inverting input terminal of the sixth operational amplifier 104f, and the other end is grounded.

The sixth potentiometer 107f is connected between the negative-phase input terminal and the output terminal of the sixth operational amplifier 104 f.

The seventh potentiometer 107g is connected between the negative phase input terminal and the output terminal of the fifth operational amplifier 104 e.

The eighth potentiometer 107h is connected between the negative-phase input terminal and the output terminal of the fourth operational amplifier 104 d.

The first potentiometer 107a, the second potentiometer 107b, the third potentiometer 107c, the fourth potentiometer 107d, the fifth potentiometer 107e, the sixth potentiometer 107f, the seventh potentiometer 107g, and the eighth potentiometer 107h are all high-precision digital potentiometers.

The output end of the eighth transformer 105h is connected in series to the first vibration mode circuit, and the input end of the eighth transformer 105h is connected to the first tuning input electrode 109a, so that the tuning function of the first vibration mode circuit is realized.

The output end of the fourteenth transformer 105n is connected in series to the second vibration mode circuit, and the input end of the fourteenth transformer 105n is connected to the second tuning input electrode 109b, so that the tuning function of the second vibration mode circuit is realized.

The input end of the ninth transformer 105i is connected in series to the first vibration mode circuit, and the output end of the ninth transformer 105i is connected with the first sensing output electrode 110a, so that the first vibration mode circuit outputs a circuit signal.

The input end of the sixteenth transformer 105p is connected in series to the second vibration mode circuit, and the output end of the sixteenth transformer 105p is connected with the second sensing output electrode 110b, so that the second vibration mode circuit outputs a circuit signal.

The signal flow in the MEMS gyroscope equivalent circuit runs as follows:

1. the Coriolis force of the first vibration mode circuit is input to a differential amplifier circuit constructed by a fourth operational amplifier 104dThe current in the second vibration mode circuit obtained through the fifteenth transformer 105o is amplified by 2L lambda times, the amplification factor is adjusted through the eighth potentiometer 107h, and then the current is amplified by omega through the second VGA106b_ZAnd the amplification is equivalent to the angular rate, and then the amplification is carried out by-2L lambda times through the third mutual inductor 105c reversely connected in series into the first vibration mode circuit.

2. The coriolis force of the second vibration mode circuit is input to a differential amplifier circuit built by a third operational amplifier 104c, the current obtained by a seventh transformer 105g in the first vibration mode circuit is amplified by 2L λ, the amplification factor is adjusted by a fourth potentiometer 107d, and then the current is amplified by omega through a first VGA106a_ZAnd the amplification is equivalent to the angular rate, and then the amplification is carried out by-2L lambda times through the tenth transformer 105j reversely connected in series into the second vibration mode circuit.

3. The damping coupling of the first vibration mode circuit is input into a transimpedance amplifier built up from a fifth operational amplifier 104e which amplifies the current in the second vibration mode circuit obtained through a thirteenth transformer 105m to R_xyThe voltage multiplied by the damping coupling factor (i.e., the damping coupling factor of the second vibration mode to the first vibration mode), the amplification factor being controlled by the seventh potentiometer 107g, is then serially connected into the first vibration mode circuit through the fourth transformer 105 d.

4. Damping coupling in second vibration mode circuit transimpedance amplifier built up from second operational amplifier 104b amplifies the current in the first vibration mode circuit derived through fifth transformer 105e to R_yxThe voltage multiplied by the damping coupling factor (i.e., the damping coupling factor of the first vibration mode to the second vibration mode), the amplification factor being controlled by the third potentiometer 107c, is then connected in series into the second vibration mode circuit through the eleventh transformer 105 k.

5. Rigid coupling input in the first vibration mode circuit the transimpedance amplifier built up from the sixth operational amplifier 104f amplifies the voltage across the second capacitor 102b in the second vibration mode circuit

Multiple (i.e. stiffness coupling multiple of the second vibration mode to the first vibration mode), amplification multiple from fifth vibration modeThe potentiometer 107e and the sixth potentiometer 107f are controlled (specifically, the amplification factor is changed by controlling the introduced resistance of the fifth potentiometer 107e and the sixth potentiometer 107 f), and then are connected in series into the first vibration mode circuit through the sixth transformer 105 f.

Wherein, c_yRepresents the size, c, of the second capacitor 102b_yxRepresenting the coupling capacitance magnitude obtained in the derivation of the formula.

6. Rigid coupling in second vibration mode circuit transimpedance amplifier built up from first operational amplifier 104a amplifies the voltage across first capacitor 102a in the first vibration mode circuit

The amplification factor (i.e., the stiffness coupling factor of the first vibration mode to the second vibration mode) is controlled by the first potentiometer 107a and the second potentiometer 107b (specifically, the amplification factor is changed by controlling the magnitude of the introduced resistance of the first potentiometer 107a and the second potentiometer 107 b), and then the first potentiometer 105k is connected in series into the second vibration mode circuit.

Wherein, c_xRepresents the size, c, of the first capacitor 102a_xyRepresenting the coupling capacitance magnitude obtained in the derivation of the formula.

7. The tuning input in the first vibration mode circuit is input by the dc power supply of the first tuning input electrode 109a, and is connected in series to the first vibration mode circuit through the eighth transformer 105 h. The tuning input in the second vibration mode circuit is input by the dc power supply of the second tuning input electrode 109b, and is connected in series to the second vibration mode circuit through the fourteenth transformer 105 n.

The invention is used for carrying out rapid test verification on the gyro measurement and control circuit through the MEMS gyro equivalent circuit. The invention can self-define the mechanical error of the equivalent gyroscope, verify the influence of each error on the system and facilitate the subsequent circuit debugging; the invention makes up the defect of unstable parameters caused by adopting the entity gyroscope in the traditional debugging method; the invention obtains diversified gyro models by changing the equivalent gyro circuit parameters, thereby obtaining a large number of samples through tests and facilitating the research of gyro error models.

Compared with the existing mechanical MEMS gyroscope, the MEMS gyroscope equivalent circuit has the following advantages:

the MEMS gyroscope equivalent circuit adopts a circuit to simulate the function of the MEMS gyroscope, is more stable and is not easily interfered by the outside compared with the mechanical structure of the MEMS gyroscope, and solves the problem that the test system is inaccurate due to low stability of the MEMS gyroscope, high possibility of being influenced by factors such as temperature and the like.

And secondly, the MEMS gyroscope equivalent circuit is set by the multiple of the amplifier, parameters such as damping coupling, stiffness coupling, angular rate signals and the like are changed according to requirements, and the damping coupling and the stiffness coupling are set to be zero, so that the electronic error introduced by a subsequent circuit can be clearly seen, and the defect that the mechanical error and the electronic error cannot be separated when the MEMS gyroscope is used for carrying out a performance test on a gyroscope modulation system, and thus the correction cannot be well carried out is overcome.

The operating speed of the MEMS gyroscope equivalent circuit can obtain an output result within a few seconds after parameters are changed, and software simulation needs several hours or more time if high precision is required, so that the method for testing the MEMS gyroscope equivalent circuit can save the experimental time and solve the problem of low simulation speed.

The invention can change the resonance frequency by changing the resistance, capacitance and inductance of the first and second vibration mode circuits, increase the diversity of the tested gyroscope, change the defect that the gyroscope with different resonance frequency needs to be replaced when the multi-resonance frequency data is needed to be obtained in the test process, obtain a large amount of experimental data and better test the diversified adaptability of the test control circuit.

It should be understood, however, that the description herein of specific embodiments is not intended to limit the invention to the particular forms disclosed, but on the contrary, the intention is to cover all modifications, equivalents, and alternatives falling within the spirit and scope of the invention as defined by the appended claims.

Claims

1. A frequency modulation gyro Lissajous modulation and self-correction test system is characterized by comprising an FDC circuit, an ADC circuit, an IQ demodulation circuit, a coherent demodulation circuit, a phase modulator, a phase-locked loop circuit, an automatic gain control circuit, a numerical control oscillator, a multiplier, an adder, an error control circuit, a DAC circuit, a low-pass filter and an electrical interface;

the output end of the first digital controlled oscillator is connected with the input end of the first multiplier, the input end of the second multiplier, the input end of the first error control circuit and the input end of the first DAC circuit;

the output ends of the first error control circuit and the first DAC circuit are respectively connected with the fourth electrical interface and the first electrical interface;

the output end of the second digital controlled oscillator is connected with the input end of the first multiplier, the input end of the second error control circuit and the input end of the second DAC circuit;

the output ends of the second error control circuit and the second DAC circuit are respectively connected with the third electrical interface and the second electrical interface;

the output end of the first multiplier is sequentially connected with one input end of the first low-pass filter and one input end of the phase modulator; the output end of the second multiplier is sequentially connected with the other input ends of the second low-pass filter and the phase modulator;

2. A FM gyrolax modulation and self-calibration test system as claimed in claim 1,

the frequency modulation gyroscope comprises a first driving input electrode, a second driving input electrode, a first tuning input electrode, a second tuning input electrode, a first induction output electrode and a second induction output electrode;

3. A FM gyrolax modulation and self-calibration test system as claimed in claim 2,

the frequency modulation gyroscope adopts an MEMS gyroscope equivalent circuit, and comprises a first vibration mode circuit, a second vibration mode circuit and a coupling circuit positioned between the first vibration mode circuit and the second vibration mode circuit;

the two VGAs are respectively a first VGA and a second VGA;

the input end of the thirteenth mutual inductor is connected with the second vibration mode circuit in series, and the output end of the thirteenth mutual inductor is connected with the positive phase input end and the negative phase input end of the fifth operational amplifier respectively; the output end of the fifth operational amplifier is connected with the input end of the fourth mutual inductor,