CN1129497A - Pulse width modulated DC-To-DC boost converter - Google Patents
Pulse width modulated DC-To-DC boost converter Download PDFInfo
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- CN1129497A CN1129497A CN95190525A CN95190525A CN1129497A CN 1129497 A CN1129497 A CN 1129497A CN 95190525 A CN95190525 A CN 95190525A CN 95190525 A CN95190525 A CN 95190525A CN 1129497 A CN1129497 A CN 1129497A
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03K—PULSE TECHNIQUE
- H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
- H03K17/08—Modifications for protecting switching circuit against overcurrent or overvoltage
- H03K17/081—Modifications for protecting switching circuit against overcurrent or overvoltage without feedback from the output circuit to the control circuit
- H03K17/0814—Modifications for protecting switching circuit against overcurrent or overvoltage without feedback from the output circuit to the control circuit by measures taken in the output circuit
- H03K17/08142—Modifications for protecting switching circuit against overcurrent or overvoltage without feedback from the output circuit to the control circuit by measures taken in the output circuit in field-effect transistor switches
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
- H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
- H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
- H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
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- Dc-Dc Converters (AREA)
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Abstract
A pulse width modulated DC-to-DC boost converter circuit (300) in which switching losses are minimized. The circuit operates at a constant frequency and in continuous mode (at a constant current). Turn-on losses are minimized by causing the parasitic intrinsic capacitance of the main switching device (302) to be discharged before the switching device is switched to the conducting state, and reducing reverse recovery current from the output rectifier (320) of the circuit when the output rectifier turns off. Turn-off losses in the switching device are minimized by causing a condition of zero volts across the switching device at turn-off.
Description
Invention field
The present invention relates to the electric pressure converter of DC-to-DC, relate in particular to those and switching loss in transducer semiconductor device is reduced to minimum DC-To-DC boost converter (boost converter) with the zero-pressure handoff technique.
Background technology
DC-To-DC boost converter often is elected to be the front end power stage in the ac/dc AC/DC transducer.DC-to-DC converter has two types: pulse-width modulation (PWM) transducer and resonance converter.PWM transducer blocking-up power flow is also controlled occupation efficiency, so that handle power supply.Resonance converter is then handled power supply with sinusoidal form.The PWM transducer is worked under constant frequency and variable pulse duration, the then frequency work to change under constant pulse width of resonance converter.Current main use the PWM transducer, because its circuit is simple and be easy to control.
In the PWM step-up converter circuit, quick-make switch produces high voltage with the two ends at an inductor.When cut-off switch, inductive current is given output capacitor charging by a diode, and to produce than the voltage that originally provided at output be high voltage.Fig. 1 shows basic (PWM) step-up converter circuit 100, and it is made up of mos field effect transistor (MOSFET) power transistor (MOSFET) 102, inductor 104, diode 106 and capacitor 108.The gate terminal of MOSFET 102 " g " is connected to the voltage source (Vswitch) 116 that an external pulse switches.The drain electrode end of MOSFET 102 " d " links to each other with diode 106 with inductor 104.The source terminal of MOSFET 102 " s " ground connection.Fig. 1 shows the voltage source (Vin) 112 of the input that is connected in circuit 100 and in the output and capacitor 108 load in parallel 114 of circuit 100.Fig. 1 also shows the electric current I L104 that flows through inductor 104, flow through the electric current I D106 of diode 106 and flow through the electric current marks such as electric current I DS102 of MOSFET102; And the voltage VL104 at inductor 104 two ends, the voltage VDS102 at MOSFET102 two ends, the voltage VC108 at capacitor 108 two ends and the voltage marks such as voltage VL at load 114 two ends.
MOSFET 102 plays a part an electronic switch, crosses the electric current I L104 of inductor 104 in order to control flows.In cycle, Vswitch 116 adds to pulse voltage the grid of MOSFET 102 at converter switch.This pulse voltage circulates MOSFET 102 between " connection " (conducting) state and " disconnection " (non-conduction) state.When MOSFET 102 connects and during conducting, be zero across the drain source voltage VDS102 of MOSFET 102, and order about electric current I L104 from Vin by inductor 104 and MOSFET 102 to ground.At this moment, the electric current I L104 in the inductor 104 is identical with electric current I DS102 among the MOSFET 102.In switch periods during this period of time, by capacitor 108 the last week the interim voltage VC108 that fills be added in the load as VL.Diode 106 stops reverse current to flow into MOSFET 102 and ground from capacitor 108.
When MOSFET 102 disconnected, the electric current I DS102 that is prevented from flowing through MOSFET 102 produced a higher voltage at the two ends of inductor 104.Disconnect constantly, the voltage at inductor 104 two ends changes polarity at once, and is increased to poor between Vin and the VL.Now, diode 106 forward bias, thus the energy that is stored in the inductor 104 is defeated by capacitor 108 and load by electric current I D106 through diode 106.The electric current that flows through inductor 104 descends, and the voltage VC108 at capacitor 108 two ends raises.
Repeat the switching circulation of above-mentioned MOSFET 102 conductings and disconnection.After time through one section setting, MOSFET 102 connects once more.Transducer is control automatically, so the average current in the inductor equals load current.Order about once more electric current from Vin through inductor 104 and MOSFET 102 to ground, the rechargeable electrical energy supply load 114 stored in the previous cycle of capacitor 108 usefulness simultaneously.The average voltage at capacitor 108 two ends relies on the pulse duration of Vswitch116 output.The cycle of switching mosfet 102 repeats with very high speed between conducting and disconnection.The pulse repetition frequency that is applied by Vswitch 116 generally can be the 30-50 KHz.Because higher frequency allows to use numerical value and all less inductor of size, use high switching frequency so wish transducer.So the volume of transducer can be done lessly, and weight is lighter.
Yet a shortcoming of operation transducer is that power switched loss meeting increases with the rising of switching frequency under higher switching frequency.In fact, these switching losses are limiting factors of selector switch frequency.Transducer is moved with high switching frequency, and the switching loss in the transducer switching device to be reduced to minimum once be a target of converter design.
In the booster circuit 100 of Fig. 1, MOSFET 102 can produce loss during switching on and off.MOSFET such as MOSFET 102 has intrinsic parasitic capacitance, and it is actually across the electric capacity between the two poles of the earth, leakage-source.This leakage-source electric capacity can make MOSFET102 be disconnected by inductive ground, capacitively connects.The due to voltage spikes that off period, leakage inductance cause produces noise and voltage stress.During the connection, the energy that is stored in leakage-source electric capacity of MOSFET 102 is consumed by internal exergy dissipation.Connect loss and depend on switching frequency and be stored in Lou-energy in the electric capacity of source.
The cause of another kind of switching loss is the connection loss among the switching transistor MOS-FET 102 in the booster circuit 100, and it is owing to the reverse recovery current in the diode 106 before diode 106 disconnections produces.When connecting MOSFET 102, one section finite time of needs comes the electric charge in the compound diode 106.To produce the negative sense spike of reverse recovery current in the diode 106, these charge recombination in diode 106.Energy from this current spike is depleted in MOSFET 102.
The connection switching loss that the reverse recovery current of leakage-source electric capacity because of MOSFET 102 and diode 106 causes has been shown among Fig. 2 A-2D, has described among the figure that MOSFET 102 connects the electric current and the voltage waveform in stages in the switch periods.Fig. 2 A shows the waveform of drain source voltage VDS102 on the MOSFET1 02.Fig. 2 B shows the waveform of the electric current I D106 that flows through diode 106.Fig. 2 C shows the waveform of the leakage-source electric current I DS102 that flows through MOSFET 102.Fig. 2 D shows the waveform of the gate source voltage Vswitch that is added on the MOSFET 102.Seen in Fig. 2 A-2D, the conducting phase of MOSFET 102 switch periods can be divided into five interval I-V.
During interval I, Vswitch is zero and MOSFET 102 disconnects.VDS102 is output-voltage levels and diode 106 two ends pressure drop sums.When the period, I began, along with Vswitch is raise by pulse, it began to rise, to connect MOSFET 102.During interval I I, Vswitch is lower than and connects the required threshold voltage of MOSFET 102.During interval I II, in the time of on Vswitch is elevated to the connection threshold value, MOSFET 102 connects.When the leakage of MOSFET 102-source capacitor discharge, VDS102 descends, and diode 106 becomes reverse bias and begins and disconnects.Electric current I D106 becomes negative because of the big pulse reverse restoring current in the diode 106.Owing to do not have current-limiting resistance to connect, so electric current I D106 is quite big with MOSFET 102 and diode 106.Because voltage VDS102 is still very high, so produce bigger power loss among the MOSFET 102.In interval I V, diode 106 has disconnected.Along with VDS102 drops to zero, produce more loss among the MOSFET 102.In period V, Vswitch rises and makes MOSFET 102 saturated, connects fully.
MOSFET 102 can limit by such method in the interval I II and the loss in the IV of connecting the cycle, promptly reduces across the voltage VDS102 of MOSFET 102 as far as possible and flowed into the reverse recovery current ID106 of MOSFET 102 by diode 106 during the connection stage of switch periods.Desirable on-condition is that the VDS102 across MOSFET 102 is set to no-voltage.VDS102 is no-voltage during utilize connecting, so that the product so the power loss of voltage and electric current is zero among the MOSFET 102.Known zero voltage switching technology is applied on the basic step-up converter circuit of Fig. 1, just can realizes this target.
The zero voltage switching technology is that the leakage-source electric capacity that makes MOSFET 102 discharges with the form of quasi-sine-wave, so that switch it when the device both end voltage is zero.Traditionally, once the PWM transducer is converted to the mixed type of PWM transducer and resonance converter with the zero voltage switching technology.These mixed types are called quasi-resonant converter.Although these quasi-resonant converters reduce the power switched loss, they do not resemble moves the real PWM transducer.In quasi-resonant converter, the voltage at switching device two ends can be as high as the twice of output voltage.Therefore, quasi-resonant converter requirement switching device can bear the voltage above the output voltage twice, and contrasts with it, and the PWM transducer only requires that switching device can bear output voltage.Quasi-resonant converter is also with variable frequency work.But people wish to obtain a kind of PWM boost converter, and it can be under continuous pattern (current constant), with constant frequency work and switching loss minimum.
The invention provides and a kind ofly can when connecting, carry out zero voltage switching and the minimum PWM converter apparatus of switching loss.This PWM boost conversion apparatus resembles and moves the real PWM transducer.In addition, it can be worked under higher frequency, and size is less, and weight is lighter.Also have, because it moves under constant frequency, therefore converter circuit of the present invention uses better simply input and output filter.Summary of the invention
On the one hand, the invention provides a kind of step-up converter circuit, it comprises first inductance device that is used for receiving the forward current that offers circuit input end and first electronic switch that is coupled with first inductance device.First electronic switch with intrinsic parasitic capacitance switches in conducting state and nonconducting state intercropping cycle, under conducting state, forward current flows through first electronic switch from first inductance device, and under nonconducting state, forward current flow to the output of circuit from first inductance device.Circuit also comprises first capacitive means with the output coupling, and is connected in first rectifying device between first inductance device and first capacitive means.First capacitive means provides an output voltage, and is recharged when first electronic switch is in nonconducting state.When first electronic switch was in nonconducting state, first rectifying device made forward current flow to first capacitive means, and when switch was in conducting state, first rectifying device stoped reverse current to flow out from first capacitive means.
Circuit of the present invention also comprises the device of the parasitic capacitance discharge that makes first electronic switch, and described device comprises second inductance device that is connected between first inductance device and first rectifying device and second electronic switch that is coupled with second inductance device.When first electronic switch was in nonconducting state, second electronic switch was switched to conducting state periodically, so that make forward current deliver to second inductance device from first rectifying device.Then, by electric current from mobile the make parasitic capacitance discharge of first electronic switch to second inductance device.When the second electronic switch conducting, second inductance device restriction is from the forward current of first inductance device with from the reverse recovery current of first rectifying device.
On the other hand, the invention provides a kind of method of using in step-up converter circuit, described circuit comprises first inductance element and is used to control first electronic switch that forward current flows to first rectifier cell from first inductance element.First switch with parasitic capacitance is switched on and off periodically so that control forward current flowing to first rectifier cell.Method of the present invention can make the parasitic capacitance discharge of first switch, makes connection loss in first switch and minimum from the caused loss of the reverse recovery current of first rectifier cell simultaneously.This method comprises when first switch disconnects, causing second inductance element from the forward current of first inductance element with from the reverse recovery current of rectifier cell, so that allow the step of parasitic capacitance discharge.This method also comprises when first switch connection, the electric current from second inductance device is caused the step of second rectifying device.Connect second electronic switch, causing second inductance device, and disconnect second electronic switch, then the electric current from second inductance element is caused second rectifier cell from the forward current of first inductance element with from the reverse recovery current of rectifier cell.
Summary of drawings
In order more in depth to understand the present invention and for other purposes of the present invention and advantage, now can be in conjunction with the accompanying drawings with reference to following description, wherein:
Fig. 1 is the electrical schematic diagram of the step-up converter circuit of prior art;
Fig. 2 A-2D is a voltage and current waveform of connecting power loss in depiction 1 circuit;
Fig. 3 is the electrical schematic diagram that comprises first step-up converter circuit of the principle of the invention;
The voltage and current waveform of switch periods in Fig. 4 A-4F depiction 3 circuit;
Fig. 5 is the electrical schematic diagram that comprises second step-up converter circuit of the principle of the invention;
Fig. 6 A-6F is the voltage and current waveform of switch periods in depiction 5 circuit.
Preferred forms of the present invention
At first, there is shown first embodiment, step-up converter circuit 300 according to principle of the invention structure with reference to Fig. 3.Circuit 300 comprises MOSFET power transistor (MOSFET) 302, inductor 304, diode 306, capacitor 308, MOSFET power transistor (MOSFET) 310, inductor 312, saturated inductor 314, diode 316, diode 318, diode 320 and resistor 322.Voltage source (Vin) 324 is connected on the input of circuit 300, and load 326 is connected on the output of circuit 300.The grid of MOSFET 302 is connected to pulse switch voltage source (Vswitch) 328, and the grid of MOSFET 310 is connected to auxiliary switch voltage source (Vswitch) 330.Vswitch 328 adds to the grid of MOSFET 302 with pulse voltage, and MOSFET 302 is switched on and off.Vswitch 330 adds to the grid of MOSFET 310 with pulse voltage, and MOSFET 310 is switched on and off.Fig. 3 also shows electric current I L304, the electric current I D306 that flows through diode 306, the electric current I L312 that flows through inductor 312, the electric current I D320 that flows through diode 320 that flows through inductor 304, the electric current marks such as electric current I R322 that flow through resistor 322; And across the voltage VDS302 of MOS-FET 302 leakage-source electrodes, across the voltage VDS310 of MOSFET 310 leakage-source electrodes, across the voltage VC308 of capacitor 308, across the voltage VD316 of diode 316 with across the voltage marks such as voltage VL of load 326.
In service, it is the same with conventional boost converter that circuit 300 plays a part, and reduces to loss that is caused by the reverse recovery current in the diode 306 and the connection loss among the MOSFET 302 minimum simultaneously.Design circuit 300 also like this makes that the loss and the connection loss among the MOSFET 310 that are caused by reverse recovery current in the diode 320 are minimum.
Circuit 300 uses the resonance handoff technique, and by this technology, the leakage of MOSFET 302-source electric capacity discharged by MOSFET 310 with mode of resonance before MOSFET 302 connects.Then, when VDS302 equals zero, MOSFET 302 is connected.Resonance switches only in the use of the connection stage of MOSFET 302 switch periods.Connection loss among the MOSFET 310 is to utilize the rising in MOSFET 310 and reduce to minimum when connecting of inductor 314 restriction electric currents.
With reference to the switch periods waveform shown in Fig. 4 A-4F, can understand the working condition of circuit 300 better.Fig. 4 A shows the waveform of drain source voltage VDS302 on the MOSFET 302.Fig. 4 B shows drain source voltage on the MOSFET 310 and the waveform of booster diode 316 both end voltage sum VDS310+VD316.Fig. 4 C shows the waveform of the electric current I D306 that flows through diode 306.Fig. 4 D shows the waveform of the electric current I L312 that flows through secondary inductor 312.Fig. 4 E shows the waveform of the voltage Vswitch 328 that adds to MOSFET 302 grids.Fig. 4 F shows the waveform of the voltage Vswitch 330 that adds to MOSFET 310 grids.In Fig. 4 A-4F, the switch periods of visible circuit 300 is divided into 6 sections interval I-VI.
With reference to Fig. 3-4, during interval I, Vswitch 328 and Vswitch 330 are zero, and MOSFET 302 and MOSFET 310 are off-state.Voltage VDS302 and VDS310+VD316 be load voltage with across diode 306 two ends pressure drop sums.At this moment, the electric current I L304 in the inductor 304 equates with electric current I D306 in the diode 306, and flow to capacitor 308 and load 326.When the period, II began, Vswitch330 is added pulse, to connect MOSFET 310.VDS310+VD316 reduces to zero, and the electric current I L312 in the inductor 312 rises, and the electric current I D306 in the diode 306 descends.Because it is the inductance of inductor 304 is quite big, for example, is the 1 milihenry order of magnitude, therefore during MOSFET 310 connects that holding current IL304 is constant.So along with the rising of electric current I L312 in the inductor 312, the electric current I D306 in the diode 306 can descend with identical speed.At first saturated inductor 314 limits the rising of electric current I L312 in the inductor 312, and this electric current is through MOSFET 310 inflow places.This has limited the connection loss of MOSFET 310.At last, saturated inductor 314 is saturated, and the climbing speed of electric current I L312 will depend on the inductance value of inductor 312 in the inductor 312.When the electric current I L312 in the inductor 312 equals to flow through the electric current I L304 of inductor 304, there is reverse recovery current in the diode 306 when disconnecting owing to diode 306, so beginning to become, electric current I D306 bears.Reverse recovery current is subjected to the restriction of electric current I D306 fall off rate.The fall off rate of ID306 is by the inductance value decision of inductor 312.This undershoot of ID306 will make the electric current I L312 in the inductor 312 be increased to a numerical value that is higher than electric current in the inductor 304, disconnect fully until diode.
When diode 306 ended, interval I II began.Electric current I L312 in the inductor 312 is greater than the electric current that flows through inductor 304 now.The part that IL312 current value in the inductor 312 surpasses the inductor 304 electric current I L304 that flow through flows out through inductor 312 from MOSFET 302, remains unchanged so that flow through the electric current I L312 of inductor 312.This electric current that exceeds makes leakage-source capacitor discharge of MOSFET 302.When the capacitor discharge of leakage-source, the diode of MOS-FET 302 will conduct this electric current that exceeds.When the VDS302 vanishing, Vswitch 328 adds pulse, to connect MOSFET 302.Because leakage-source electric capacity discharges, will can not produce Lou-the source capacity loss so connect.
When interval I V began, MOSFET 310 was disconnected.Electric current I L312 reversed flow in the inductor 312 is to diode 320 and capacitor 308.Electric current I L312 in the inductor 312 will descend with a certain speed according to the inductance value of output voltage and inductor 312.Simultaneously, the electric current among the MOSFET 302 will raise, thereby total current remains unchanged, and equal the electric current I L304 in the inductor 304.When the electric current I L312 in the inductor 312 approached zero, inductor 314 was unsaturated and fall off rate IL312 is further slack-off.Owing to reverse recovery current is arranged in the diode 320, so when diode 320 reverse bias and when beginning to disconnect, IL312 presents negative current spike.This reverse recovery current will be subjected to the restriction of inductor 314.
When diode 320 disconnected, the period, V began.At this moment, negative current ID320 commutates and flows out from ground as electric current I R322, through resistor 322, diode 318 and inductor 312 and 314.Diode 316 prevents that the body diode through MOSFET 310 flows into inductor 312 to electric current from ground.Now the electric current I L312 in the inductor 312 descends with the speed of the resistance value that depends on resistor 322.The resistive damping effect of resistor 322 prevents to take place in inductor 312, inductor 314, diode 318 and the diode 306 excessive damped oscillation.This can prevent across the voltage of diode 316 and diode 320 excessive when diode 320 disconnects.
When Vswitch 328 being made as zero so that MOSFET 302 when disconnecting, the cycle finishes in period VI.Electric current among the MOSFET 302 by reversed flow to diode 306, diode 306 conductings then, and the electric current I D306 in the diode 306 raises.Then begin next switch periods, and proceed as mentioned above in interval I.
Fig. 5 shows second embodiment according to the present invention's structure, step-up converter circuit 500.Circuit 500 comprises MOSFET power transistor (MOSFET) 502, inductor 504, diode 506, capacitor 508, switch MOS FET power transistor (MOSFET) 510, inductor 512, saturated inductor 514, diode 516, diode 518, diode 520, resistor 522, diode 534 and capacitor 536.Voltage source (Vin) 524 is connected on the input of circuit 500, and load 526 is connected on the output of circuit 500.Vswitch 528 adds to the grid of MOSFET 502 with a pulse voltage, and switching on and off MOSFET 502, and Vswitch 530 adds to the grid of MOS-FET 510 with a pulse voltage, to switch on and off MOSFET 510.
Fig. 5 also shows electric current I L504, the electric current I D506 that flows through diode 506, the electric current I L512 that flows through inductor 512, the electric current I D520 that flows through diode 520 that flow through inductor 504, flow through the electric current I R522 of resistor 522 and flow through the electric current marks such as electric current I D534 of diode 534; And across the voltage VDS502 of MOSFET 502 leakage-source electrodes, across the voltage VC508 of capacitor 508, across the voltage VDS510 of MOSFET 510 leakage-source electrodes, across the voltage VC536 of capacitor 536, across the voltage VD516 of diode 51 6 with across the voltage marks such as voltage VL of load 526.
In service, it is the same with conventional boost converter that circuit 500 plays a part, and loss that according to the present invention reverse recovery current in the diode 506 is caused simultaneously and the loss that switches on and off among the MOSFET 502 are reduced to minimum.Like this design circuit 500 also makes to switch on and off the loss minimum among loss that the reverse recovery current in the diode 520 causes and the MOSFET 510.
Fig. 6 A shows the waveform of drain source voltage VDS502 on the MOSFET 502.Fig. 6 B shows the drain source voltage and waveform across the voltage sum VDS510+VD516 of booster diode 516 on the MOSFET 510.Fig. 6 C shows the waveform of the electric current I D506 that flows through diode 506.Fig. 6 D shows the waveform of the electric current I L512 that flows through secondary inductor 512.Fig. 6 E shows the waveform of the voltage Vswitch 528 that adds to MOSFET 502 grids.Fig. 6 F shows the waveform of the voltage Vswitch 530 that adds to MOSFET 510 grids.The switch periods of Fig. 5 circuit can be divided into 6 sections interval I-VI.
In interval I-III, the element 502-522 in Fig. 5 circuit 500 is identical with element 302-322 function among Fig. 3 respectively basically.During interval I V-VI, the working condition of the circuit 500 relevant work situation with foregoing circuit 300 basically is identical, and the voltage conditions that VC536 upward exists during just owing to MOSFET 502 or MOSFET 510 disconnections makes the disconnection loss minimum in MOSFET 502 and 510.The working condition of circuit 500 in during interval I V-VI below described.
With reference to Fig. 5-6, when interval I V began, Vswitch 530 switched to zero and MOSFET 510 disconnects.Electric current I L512 in the inductor 512 is by 536 chargings of 520 pairs of capacitors of diode now.Capacitor 536 begins discharge and VC536 is zero.When capacitor 536 is charging, equal voltage VC536 on the capacitor 536 across the voltage VDS510 of MOSFET 510.Since when blocking-up VDS510 and VD516 be zero, institute is so that the disconnection loss among the MOSFET 510 reduces to minimum.This makes in the interval I V among Fig. 6 the climbing speed of VDS510 and VD516 slower than the climbing speed of VDS310 in the interval I V among Fig. 4 and VD316.When capacitor 536 charged to output voltage, the electric current I L512 in the inductor 512 flowed into diode 534 as electric current I D534, and diode 534 becomes forward bias and connection.Electric current I L512 in the inductor 512 descends with a certain speed according to the voltage VL of output and the inductance of inductor 512 now.When the electric current I L512 in the inductor 512 near zero the time, the fall off rate of the no longer saturated and IL512 that further slows down of inductor 514.Owing in diode 520 and the diode 534 reverse recovery current is arranged, so when diode 520 and diode 534 became reverse bias and begin to disconnect, electric current I L512 presented negative current spike.This reverse recovery current will be subjected to the restriction of inductor 514 then.
When the period, V began, diode 520 and diode 534 disconnected, and the medium and small reverse current IL512 commutation of inductor 512 and flow out from ground as electric current I R522, through resistor 522, diode 518 and inductor 514 and 512, flowed into MOSFET502 at last.Diode 516 prevents that the body diode by MOSFET 510 flows into inductor 512 to electric current from ground.Electric current I L512 descends with the fall off rate of the resistance value that depends on resistor 522.When diode 520 disconnected, the damping effect of resistor 522 prevented that the voltage on diode 516 and the diode 520 is excessive.Electric current flows through inductor 504 and MOSFET502 now, and equals output voltage across the voltage VC536 of capacitor 536.
In period VI zero hour, Vswitch 528 goes to zero and MOSFET 502 disconnects.Now VDS502 equals VL and deducts voltage VC536 across capacitor 536, and approaches zero.Disconnection loss minimum among the MOSFET 502 thus.Electric current I L504 in the inductor 504 makes capacitor 536 be discharged to zero by diode 534.This makes the climbing speed of the interior VDS502 of period VI among Fig. 6 be slightly slower than the climbing speed of the interior VDS302 of period VI among Fig. 4.After capacitor 536 discharges fully, diode 506 will be connected, and diode 534 will disconnect.Switch periods finishes now.
Below be a tabulation, illustration can be used to set up element step-up converter circuit, typical industry standard and circuit parameter with service chart 3 and Fig. 5.
| ?MOSFET?302，502 | ????????????IRF460 |
| ?MOSFET?310，510 | ????????????IRF840 |
| Diode 306,506 | ???????????APT30D60B |
| Diode 316,516 | ????????Philips?BYM26C |
| Diode 318,518 | ????????Philips?BYM26C |
| Diode 320,520 | ????????Philips?BYM26C |
| Diode 534 | ????????Philips?BYM26C |
| Resistor 322,522 | 20 ohm |
| Inductor 304,504 | 1 milihenry |
| Inductor 312,512 | 4 microhenrys |
| Inductor 314,514 | 6 circles are on Toshiba SA14 * 8 * 4.5 |
| Capacitor 536 | 6.8 nanofarad |
| Capacitor 308,508 | C=1 millifarad (variable) |
| ?????? |
400 volts |
| ??????Vin | 230 volts |
| Switching frequency | 50 KHz |
It will be understood by those skilled in the art that these component values give an example as representative value, and available many different component values and the circuit parameter circuit of realizing Fig. 3 and Fig. 5.It is evident that equally, can not break away from the spirit and scope of the present invention structure content shown in the accompanying drawing and that discuss is in conjunction with the accompanying drawings done various changes.Therefore, should think the present invention be not limited to shown in and described particular content.
Claims (18)
1. a step-up converter circuit is characterized in that, comprising:
First inductance device is used to receive the forward current that offers described circuit input end;
First electronic switch, it links to each other with described first inductance device, described first electronic switch has intrinsic parasitic capacitance and the cycle of doing between conducting state and nonconducting state switches, wherein under conducting state, forward current flows through described switch from described first inductance device, under nonconducting state, forward current flow to the output of described circuit from described first inductance device;
First capacitive means, it links to each other with the described output that output voltage is provided, and when described first electronic switch was in nonconducting state, described capacitive means was recharged;
First rectifying device, it is connected between described first inductance device and described first capacitive means, when described first electronic switch is in nonconducting state, described first rectifying device makes forward current flow to described first capacitive means, and when described switch was in conducting state, described first rectifying device stoped reverse current to flow out from described first capacitive means; With
Be used for making the device of the intrinsic parasitic capacitance discharge of described first electronic switch, it comprises:
Second inductance device, it is connected between described first inductance device and described first rectifying device; With
Second electronic switch, it links to each other with described second inductance device, when described first electronic switch is in nonconducting state, described second electronic switch switches to conducting state periodically, thereby make forward current branch to described second inductance device from described first rectifying device, described parasitic capacitance is by flowing to the current discharge of described second inductance device from described first electronic switch.
2. step-up converter circuit as claimed in claim 1, it is characterized in that, when the described second electronic switch conducting, described second inductance device limits forward current that flows out from described first inductance device and the reverse recovery current that flows out from described first rectifying device.
3. step-up converter circuit as claimed in claim 1 is characterized in that, described first inductance device has than the bigger inductance of described second inductance device.
4. step-up converter circuit as claimed in claim 1 is characterized in that described second inductance device comprises the tandem compound of an inductor and a saturated inductor.
5. step-up converter circuit as claimed in claim 1 is characterized in that, the described device that is used to discharge further comprises:
Second rectifying device, it is connected between described second inductance device and described second electronic switch;
The 3rd rectifying device, it links to each other with described first rectifying device with described second inductance device; With
The 4th rectifying device, it links to each other with the 3rd rectifying device with described second.
6. step-up converter circuit as claimed in claim 5 is characterized in that,
Described first and second electronic switches all comprise a MOSFET power transistor;
The described first, second, third and the 4th rectifying device all comprises a diode; And
Described first capacitive means comprises a capacitor.
7. step-up converter circuit as claimed in claim 1 is characterized in that, the described device that is used to discharge also comprises:
Second capacitive means, it links to each other with described second inductance device;
Second rectifying device, it is connected between described second inductance device and described second electronic switch;
The 3rd rectifying device, it is connected between described second inductance device and described second capacitive means;
The 4th rectifying device, it links to each other with the 3rd rectifying device with described second; With
The 5th rectifying device, it is connected between described first rectifying device and second capacitive means.
8. step-up converter circuit as claimed in claim 7 is characterized in that,
Described first and second electronic switches all comprise a MOSFET power transistor;
The described first, second, third, fourth and the 5th rectifying device all comprises a diode; And
Described first and second capacitive means comprise a capacitor.
9. method of in the step-up converter circuit that comprises first inductance element and first electronic switch, using, wherein said first electronic switch is used to control forward current and flows to first rectifier cell from described first inductance element, described first switch has parasitic capacitance and switches on and off periodically, described method is used to make the parasitic capacitance discharge of described first switch, simultaneously the connection loss in described first switch and from the reverse recovery current of described first rectifier cell caused loss reduce to minimum, it is characterized in that described method comprises the following steps:
When described first switch disconnects, will cause to second inductance element from the forward current of described first inductance element with from the reverse recovery current of described first rectifier cell, thereby make described parasitic capacitance discharge; With
When described first switch connection, will guide to second rectifier cell from the electric current of described second inductance element.
10. method as claimed in claim 9, it is characterized in that, connect second electronic switch, causing described second inductance element from the forward current of described first inductance element with from the reverse recovery current of described first rectifier cell, and disconnect second electronic switch, will cause described second rectifier cell from the electric current of described second inductance element.
11. method as claimed in claim 10 is characterized in that, described second inductance element comprises the tandem compound of an inductor and a saturated inductor.
12. method as claimed in claim 10 is characterized in that, described first inductance element has than the bigger inductance of described second inductance element.
13. method as claimed in claim 10 is characterized in that, described second rectifier cell links to each other with first capacity cell, and when each described electronic switch disconnected respectively, the restriction of described first capacity cell was across described first and the voltage of described second electronic switch.
14. method as claimed in claim 13 is characterized in that, the 3rd rectifier cell is connected between described first rectifier cell and described first capacity cell.
15. method as claimed in claim 14 is characterized in that, described first links to each other with second capacity cell with the output of the 3rd rectifier cell at described circuit.
16. method as claimed in claim 10, it is characterized in that, described the 3rd rectifier cell links to each other with described second inductance element with described second rectifier cell, when described second switch disconnects, described the 3rd rectifier cell causes described second inductance element with electric current from ground, in order to the decline of compensation from the reverse recovery current of described second rectifier cell.
17. a step-up converter circuit is characterized in that, comprising:
First inductor, it has first end and second end, and described first end links to each other with the input of described circuit;
First switch, it has conducting state and nonconducting state, when described first switch is in conducting state, second end of described first inductor and ground is connected, and when described first switch was in nonconducting state, it disconnected second end of described first inductor and ground;
First rectifier, it has first end and second end, and first end of described first rectifier links to each other with second end of described first inductor, and second end of described first rectifier links to each other with the output of described circuit;
One capacitor, it has first end and second end, and first end of described first capacitor links to each other with the output of described circuit, and the second end ground connection of described capacitor;
Second inductor, it has first end and second end, and first end of described second inductor links to each other with second end of described first inductor;
The 3rd inductor, it has first end and second end, and first end of described the 3rd inductor links to each other with second end of described second inductor;
Second rectifier, it has first end and second end, and first end of described second rectifier links to each other with second end of described the 3rd inductor, and second end of described second rectifier links to each other with the output of described circuit;
The 3rd rectifier, it has first end and second end, and first end of described the 3rd rectifier links to each other with second end of described the 3rd inductor;
Second switch, it has conducting state and nonconducting state, when described second switch is in conducting state, second end of described the 3rd rectifier and ground is connected, and when described second switch was in to conducting state, it disconnected second end of described the 3rd rectifier and ground;
The 4th rectifier, it has first end and second end, and second end of described the 4th rectifier links to each other with first end of described the 3rd rectifier; With
One resistor, it has first end and second end, and first end of described resistor links to each other with first end of described the 4th rectifier, and the second end ground connection of described resistor.
18. a step-up converter circuit is characterized in that, comprising:
First inductor, it has first end and second end, and described first end links to each other with the input of described circuit;
First switch, it has conducting state and nonconducting state, when described first switch is in conducting state, second end of described first inductor and ground is connected, and when described first switch was in nonconducting state, it disconnected second end of described first inductor and ground;
First rectifier, it has first end and second end, and first end of described first rectifier links to each other with second end of described first inductor, and second end of described first rectifier links to each other with the output of described circuit;
First capacitor, it has first end and second end, and first end of described capacitor links to each other with the output of described circuit, and the second end ground connection of described first capacitor;
Second inductor, it has first end and second end, and first end of described second inductor links to each other with second end of described first inductor;
The 3rd inductor, it has first end and second end, and first end of described the 3rd inductor links to each other with second end of described second inductor;
Second rectifier, it has first end and second end, and first end of described second rectifier links to each other with second end of described the 3rd inductor;
The 3rd rectifier, it has first end and second end, and first end of described the 3rd rectifier links to each other with second end of described the 3rd inductor;
Second switch, it has conducting state and nonconducting state, when described second switch is in conducting state, second end of described the 3rd rectifier and ground is connected, and when described second switch was in nonconducting state, it disconnected second end of described the 3rd rectifier and ground;
The 4th rectifier, it has first end and second end, and second end of described the 4th rectifier links to each other with first end of described second rectifier; With
One resistor, it has first end and second end, and first end of described resistor links to each other with first end of described the 4th rectifier, and the second end ground connection of described resistor.
Second capacitor, it has first end and second end, and first end of described second capacitor links to each other with first end of described first rectifier, and second end of described second capacitor links to each other with second end of described second rectifier; With
The 5th rectifier, it has first end and second end, and first end of described the 5th rectifier links to each other with second end of described second capacitor, and second end of described the 5th rectifier links to each other with the output of described circuit.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| US08/255,380 US5543704A (en) | 1994-06-08 | 1994-06-08 | Pulse width modulated DC-to-DC boost converter |
| US08/255,380 | 1994-06-08 |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CN1129497A true CN1129497A (en) | 1996-08-21 |
| CN1041984C CN1041984C (en) | 1999-02-03 |
Family
ID=22968066
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CN95190525A Expired - Fee Related CN1041984C (en) | 1994-06-08 | 1995-06-07 | Pulse width modulated dc-to-dc boost converter |
Country Status (12)
| Country | Link |
|---|---|
| US (1) | US5543704A (en) |
| EP (1) | EP0712546B1 (en) |
| JP (1) | JPH09504160A (en) |
| CN (1) | CN1041984C (en) |
| AU (1) | AU704193B2 (en) |
| BR (1) | BR9506002A (en) |
| CA (1) | CA2169160A1 (en) |
| DE (1) | DE69522169T2 (en) |
| ES (1) | ES2161899T3 (en) |
| FI (1) | FI960569A0 (en) |
| NO (1) | NO960493L (en) |
| WO (1) | WO1995034120A1 (en) |
Cited By (5)
| Publication number | Priority date | Publication date | Assignee | Title |
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| CN1578081B (en) * | 2003-07-29 | 2010-06-02 | 因芬尼昂技术股份公司 | Device and method for converting DC input voltage to multiple DC output voltages |
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Families Citing this family (35)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| SE511059C2 (en) * | 1997-01-24 | 1999-07-26 | Ericsson Telefon Ab L M | Method and device for power conversion. |
| EP0910158A4 (en) * | 1997-02-10 | 2000-05-10 | Tdk Corp | Step-up switching power supply |
| US5914587A (en) * | 1997-08-05 | 1999-06-22 | Lucent Technologies Inc. | Circuit for reducing switching losses of a power converter and method of operation thereof |
| US5841268A (en) * | 1997-09-29 | 1998-11-24 | Power Architects Corporation | Multi-resonant soft switching snubber network for DC-to-DC converter |
| DE19808637A1 (en) * | 1998-02-28 | 1999-09-09 | Bosch Gmbh Robert | DC / DC converter with a transformer and a choke |
| US6118673A (en) * | 1998-06-01 | 2000-09-12 | Virginia Power Technologies, Inc. | Single-stage AC/DC converters with saturable conductor PFC |
| US5991174A (en) * | 1998-08-28 | 1999-11-23 | Lucent Technologies Inc. | Snubber circuit for a rectifier, method of operation thereof and power converter employing the same |
| US6028418A (en) * | 1999-02-11 | 2000-02-22 | Delta Electronics, Inc. | Boost converter with minimum-component-count active snubber |
| US6002603A (en) * | 1999-02-25 | 1999-12-14 | Elliott Energy Systems, Inc. | Balanced boost/buck DC to DC converter |
| US6128206A (en) * | 1999-03-12 | 2000-10-03 | Ericsson, Inc. | Clamping circuit and method for synchronous rectification |
| CA2377503A1 (en) | 1999-06-30 | 2001-01-11 | Peco Ii, Inc. | Diode recovery current suppression circuits |
| JP3495660B2 (en) * | 1999-09-30 | 2004-02-09 | 三洋電機株式会社 | DC-DC converter circuit |
| AU2001250144A1 (en) * | 2000-04-12 | 2001-10-30 | Wolfgang Croce | Circuit for reducing switching losses in electronic valves |
| US6236191B1 (en) * | 2000-06-02 | 2001-05-22 | Astec International Limited | Zero voltage switching boost topology |
| KR100377133B1 (en) | 2000-08-29 | 2003-03-19 | 페어차일드코리아반도체 주식회사 | A motor driving circuit using a pwm input signal |
| JP3425418B2 (en) * | 2000-09-20 | 2003-07-14 | ティーディーケイ株式会社 | Step-up switching power supply |
| US6434029B1 (en) | 2001-10-17 | 2002-08-13 | Astec International Limited | Boost topology having an auxiliary winding on the snubber inductor |
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| US9855620B2 (en) | 2005-02-07 | 2018-01-02 | Lincoln Global, Inc. | Welding system and method of welding |
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| JP2006324534A (en) * | 2005-05-20 | 2006-11-30 | Seiko Instruments Inc | Light emitting diode driving circuit |
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| EP2230754B1 (en) | 2009-03-18 | 2015-04-29 | STMicroelectronics (Tours) SAS | Switching mode power supply |
| TWI418130B (en) * | 2011-05-19 | 2013-12-01 | Univ Nat Taipei Technology | Step-up conversion circuit |
| CN105529925B (en) * | 2016-02-01 | 2019-04-09 | 浙江艾罗网络能源技术有限公司 | Boost based on switched inductors |
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Family Cites Families (17)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US4184197A (en) * | 1977-09-28 | 1980-01-15 | California Institute Of Technology | DC-to-DC switching converter |
| US4654769A (en) * | 1984-11-02 | 1987-03-31 | California Institute Of Technology | Transformerless dc-to-dc converters with large conversion ratios |
| DE3606896A1 (en) * | 1985-12-03 | 1987-06-04 | Zdzislaw Gulczynski | SWITCHING POWER SUPPLY |
| US4720668A (en) * | 1986-06-20 | 1988-01-19 | Lee Fred C | Zero-voltage switching quasi-resonant converters |
| US4841220A (en) * | 1987-09-23 | 1989-06-20 | Tabisz Wojciech A | Dc-to-Dc converters using multi-resonant switches |
| GB8816774D0 (en) * | 1988-07-14 | 1988-08-17 | Bsr Int Plc | Power supplies |
| US4903182A (en) * | 1989-03-20 | 1990-02-20 | American Telephone And Telegraph Company, At&T Bell Laboratories | Self-oscillating converter with light load stabilizer |
| US4940929A (en) * | 1989-06-23 | 1990-07-10 | Apollo Computer, Inc. | AC to DC converter with unity power factor |
| US4959764A (en) * | 1989-11-14 | 1990-09-25 | Computer Products, Inc. | DC/DC converter switching at zero voltage |
| US5180964A (en) * | 1990-03-28 | 1993-01-19 | Ewing Gerald D | Zero-voltage switched FM-PWM converter |
| FR2671922B1 (en) * | 1991-01-22 | 1994-02-11 | Agence Spatiale Europeenne | METHOD FOR REDUCING THE SWITCHING LOSSES CREATED BY A POWER SWITCH. |
| US5418704A (en) * | 1992-06-12 | 1995-05-23 | Center For Innovative Technology | Zero-voltage-transition pulse-width-modulated converters |
| SE500589C2 (en) * | 1992-10-22 | 1994-07-18 | Ericsson Telefon Ab L M | Low-loss boost converter through limited back current in the main diode |
| US5307005A (en) * | 1992-12-23 | 1994-04-26 | International Business Machines Corporation | Zero current switching reverse recovery circuit |
| US5313382A (en) * | 1993-05-18 | 1994-05-17 | At&T Bell Laboratories | Reduced voltage/zero current transition boost power converter |
| US5457379A (en) * | 1993-10-15 | 1995-10-10 | At&T Ipm Corp. | High efficiency switch mode regulator |
| US5434767A (en) * | 1994-01-10 | 1995-07-18 | University Of Central Florida | Power converter possessing zero-voltage switching and output isolation |
-
1994
- 1994-06-08 US US08/255,380 patent/US5543704A/en not_active Expired - Fee Related
-
1995
- 1995-06-07 CN CN95190525A patent/CN1041984C/en not_active Expired - Fee Related
- 1995-06-07 EP EP95922063A patent/EP0712546B1/en not_active Expired - Lifetime
- 1995-06-07 AU AU26882/95A patent/AU704193B2/en not_active Ceased
- 1995-06-07 ES ES95922063T patent/ES2161899T3/en not_active Expired - Lifetime
- 1995-06-07 WO PCT/SE1995/000665 patent/WO1995034120A1/en active IP Right Grant
- 1995-06-07 BR BR9506002A patent/BR9506002A/en not_active Application Discontinuation
- 1995-06-07 JP JP8500772A patent/JPH09504160A/en active Pending
- 1995-06-07 CA CA002169160A patent/CA2169160A1/en not_active Abandoned
- 1995-06-07 DE DE69522169T patent/DE69522169T2/en not_active Expired - Fee Related
-
1996
- 1996-02-07 NO NO960493A patent/NO960493L/en not_active Application Discontinuation
- 1996-02-07 FI FI960569A patent/FI960569A0/en unknown
Cited By (6)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN1055804C (en) * | 1998-04-27 | 2000-08-23 | 深圳市华为电气股份有限公司 | Topological circuit for soft switch |
| CN1320751C (en) * | 2002-11-14 | 2007-06-06 | 国际整流器公司 | Circuit for providing resistance to single event upset to pulse width modulator integrated circuit |
| CN1578081B (en) * | 2003-07-29 | 2010-06-02 | 因芬尼昂技术股份公司 | Device and method for converting DC input voltage to multiple DC output voltages |
| CN101345489B (en) * | 2008-03-06 | 2010-12-08 | 上海海事大学 | Current transformer for limiting reverse recovery current |
| CN112671239A (en) * | 2019-10-15 | 2021-04-16 | 立锜科技股份有限公司 | Flyback power supply circuit and secondary side control circuit and control method thereof |
| CN112671239B (en) * | 2019-10-15 | 2024-02-27 | 立锜科技股份有限公司 | Flyback power supply circuit, secondary side control circuit and control method thereof |
Also Published As
| Publication number | Publication date |
|---|---|
| FI960569A (en) | 1996-02-07 |
| US5543704A (en) | 1996-08-06 |
| AU2688295A (en) | 1996-01-04 |
| AU704193B2 (en) | 1999-04-15 |
| CN1041984C (en) | 1999-02-03 |
| ES2161899T3 (en) | 2001-12-16 |
| NO960493L (en) | 1996-02-28 |
| JPH09504160A (en) | 1997-04-22 |
| NO960493D0 (en) | 1996-02-07 |
| DE69522169T2 (en) | 2001-11-29 |
| EP0712546A1 (en) | 1996-05-22 |
| FI960569A0 (en) | 1996-02-07 |
| CA2169160A1 (en) | 1995-12-14 |
| BR9506002A (en) | 1997-08-19 |
| EP0712546B1 (en) | 2001-08-16 |
| WO1995034120A1 (en) | 1995-12-14 |
| DE69522169D1 (en) | 2001-09-20 |
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