CN1129497A - Pulse Width Modulation DC-DC Boost Converter - Google Patents

Abstract

A pulse width modulated DC-DC boost converter circuit (300) that minimizes switching losses. The circuit operates at a constant frequency and in continuous mode (constant current). Turn-on losses are minimized by discharging the inherent parasitic capacitance of the main switching device (302) before the switching device switches to the on-state and reducing the reverse recovery current of the output rectifier (320) in the circuit when the output rectifier is turned off. lowest. Turn-off losses in the switching device are minimized by creating a zero voltage condition across the switching device at turn-off.

Description

Pulse Width Modulation DC-DC Boost Converter

field of invention

The present invention relates to DC-DC voltage converters, in particular to those DC-DC boost converters which use zero-voltage switching technology to minimize switching losses in converter semiconductor devices.

Background technique

A DC-DC boost converter is often chosen as the front-end power stage in an AC/DC AC/DC converter. There are two types of DC-DC converters: pulse width modulation (PWM) converters and resonant converters. A PWM converter blocks power flow and controls the duty cycle in order to handle the power. A resonant converter handles the power supply sinusoidally. PWM converters operate at a constant frequency and variable pulse width, while resonant converters operate at a constant pulse width with varying frequency. PWM converters are mainly used today because of their circuit simplicity and ease of control.

In a PWM boost converter circuit, a switch is turned on quickly to generate a high voltage across an inductor. When the switch is turned off, the inductor current through a diode charges an output capacitor and produces a higher voltage at the output than was originally supplied. FIG. 1 shows a basic (PWM) boost converter circuit 100 consisting of a metal oxide semiconductor field effect transistor (MOSFET) power transistor (MOSFET) 102 , an inductor 104 , a diode 106 and a capacitor 108 . The gate terminal "g" of MOSFET 102 is connected to an external pulse-switched voltage source (Vswitch) 116. The drain terminal "d" of MOSFET 102 is connected to inductor 104 and diode 106. The source terminal "s" of MOSFET 102 is grounded. FIG. 1 shows a voltage source (Vin) 112 connected to the input of circuit 100 and a load 114 connected in parallel with capacitor 108 at the output of circuit 100 . Fig. 1 also shows current markers such as the current IL104 flowing through the inductor 104, the current ID106 flowing through the diode 106, and the current IDS102 flowing through the MOSFET 102; Voltage marks such as the voltage VC108 across the two ends and the voltage VL across the load 114 .

MOSFET 102 acts as an electronic switch to control current IL 104 flowing through inductor 104 . Vswitch 116 applies a pulsed voltage to the gate of MOSFET 102 during a converter switching cycle. The pulsed voltage cycles MOSFET 102 between an "on" (conducting) state and an "off" (non-conducting) state. When MOSFET 102 is on and conducting, the drain-to-source voltage VDS102 across MOSFET 102 is zero and drives current IL104 from Vin through inductor 104 and MOSFET 102 to ground. At this time, the current IL104 in the inductor 104 is the same as the current IDS102 in the MOSFET 102. During this time of the switching cycle, the voltage VC108 charged by capacitor 108 in the previous cycle is applied to the load as VL. Diode 106 blocks reverse current flow from capacitor 108 into MOSFET 102 and ground.

The current IDS102 prevented from flowing through MOSFET 102 generates a higher voltage across inductor 104 when MOSFET 102 is turned off. At the moment of turn-off, the voltage across the inductor 104 changes polarity momentarily and rises to the difference between Vin and VL. Diode 106 is now forward biased so that the energy stored in inductor 104 is transferred by current ID 106 through diode 106 to capacitor 108 and the load. The current flowing through the inductor 104 decreases, and the voltage VC108 across the capacitor 108 increases.

The switching cycle of switching MOSFET 102 on and off as described above is repeated. After a set period of time, MOSFET 102 is turned on again. The converter is automatically controlled so that the average current in the inductor is equal to the load current. Current is again driven from Vin through inductor 104 and MOSFET 102 to ground, while capacitor 108 supplies load 114 with the charging energy stored in the previous cycle. The average voltage across capacitor 108 depends on the pulse width output by Vswitch 116 . The cycle of switching MOSFET 102 between on and off repeats at a very high rate. The frequency of the pulsed voltage applied by Vswitch 116 may typically be 30-50 kilohertz. High switching frequencies are desirable for converters since higher frequencies allow the use of smaller value and smaller inductors. Therefore, the volume of the converter can be made smaller and lighter in weight.

However, one disadvantage of operating the converter at higher switching frequencies is that switching power losses increase as the switching frequency increases. In fact, these switching losses are the limiting factor in choosing the switching frequency. It has been a goal of converter design to operate the converter at high switching frequency and to minimize the switching losses in the converter's switching elements.

In the boost circuit 100 of FIG. 1, the MOSFET 102 generates losses both during turn-on and turn-off. A MOSFET such as MOSFET 102 has an inherent parasitic capacitance, which is actually a capacitance across drain-source. This drain-to-source capacitance causes MOSFET 102 to be inductively turned off and capacitively turned on. During turn-off, voltage spikes caused by leakage inductance generate noise and voltage stress. During turn-on, the energy stored in the drain-source capacitance of MOSFET 102 is internally dissipated. Turn-on losses depend on the switching frequency and the energy stored in the drain-source capacitance.

Another source of switching losses in the boost circuit 100 is the turn-on loss in the switching transistor MOS-FET 102 due to the reverse recovery current in the diode 106 before the diode 106 turns off. When MOSFET 102 is turned on, a finite period of time is required to recombine the charge in diode 106. A negative-going spike in reverse recovery current will be generated in diode 106 until these charges in diode 106 recombine. Energy from this current spike is dissipated in MOSFET 102.

Turn-on switching losses due to the drain-source capacitance of MOSFET 102 and the reverse recovery current of diode 106 are shown in FIGS. 2A-2D , which depict current and voltage waveforms during the turn-on phase of MOSFET 102 during a switching cycle. FIG. 2A shows the waveform of drain-source voltage VDS102 on MOSFET102. FIG. 2B shows the waveform of current ID106 flowing through diode 106 . FIG. 2C shows the waveform of drain-source current IDS102 flowing through MOSFET 102. FIG. 2D shows the waveform of the gate-source voltage Vswitch applied to MOSFET 102. As seen in FIGS. 2A-2D , the conduction phase of the MOSFET 102 switching cycle can be divided into five periods I-V.

During period I, Vswitch is zero and MOSFET 102 is off. VDS 102 is the sum of the output voltage level and the voltage drop across diode 106 . When period I begins, as Vswitch is pulsed high, it begins to rise to turn on MOSFET 102. During period II, Vswitch is below the threshold voltage required to turn on MOSFET 102 . During period III, when Vswitch rises above the turn-on threshold, MOSFET 102 is turned on. As the drain-source capacitance of MOSFET 102 discharges, VDS 102 drops and diode 106 becomes reverse biased and begins to turn off. Current ID106 becomes negative due to the large pulsed reverse recovery current in diode 106 . Since there is no current limiting resistor in series with MOSFET 102 and diode 106, current ID 106 is quite large. Because voltage VDS 102 is still high, a large power loss occurs in MOSFET 102. In period IV, diode 106 has been switched off. As VDS 102 falls to zero, more losses occur in MOSFET 102. During period V, Vswitch rises and saturates MOSFET 102, fully on.

The losses in MOSFET 102 during periods III and IV of the on-cycle can be limited by minimizing the voltage VDS102 across MOSFET 102 and the inverse To the recovery current ID106. The ideal turn-on condition is to set VDS 102 across MOSFET 102 to zero voltage. The zero voltage on VDS 102 during turn-on is utilized so that the product of voltage times current in MOSFET 102 and therefore zero power loss. This goal can be achieved by applying known zero-voltage switching techniques to the basic boost converter circuit of Figure 1.

ZVS is the technique of discharging the drain-source capacitance of MOSFET 102 in a quasi-sine wave to switch it when the voltage across the device is zero. Traditionally, a PWM converter has been converted into a hybrid of a PWM converter and a resonant converter using zero-voltage switching techniques. These hybrids are called quasi-resonant converters. Although these quasi-resonant converters reduce switching power losses, they do not operate like true PWM converters. In a quasi-resonant converter, the voltage across the switching device can be as high as twice the output voltage. Therefore, a quasi-resonant converter requires the switching device to withstand more than twice the output voltage, whereas a PWM converter requires only the switching device to withstand the output voltage. Quasi-resonant converters also operate at variable frequencies. But it is desirable to obtain a PWM boost converter that can operate at a constant frequency in continuous mode (constant current) with minimal switching losses.

The present invention provides a PWM converter device capable of zero-voltage switching at turn-on with the lowest switching losses. The PWM boost converter device operates like a real PWM converter. Plus, it works at higher frequencies and is smaller and lighter. Also, since it operates at a constant frequency, the converter circuit of the present invention uses simpler input and output filters. Contents of the invention

In one aspect, the present invention provides a boost converter circuit comprising first inductive means for receiving forward current supplied to an input terminal of the circuit, and a first electronic switch coupled to the first inductive means. The first electronic switch having inherent parasitic capacitance is periodically switched between a conduction state and a non-conduction state, in the conduction state, a forward current flows from the first inductive means through the first electronic switch, and in the non-conduction state , forward current flows from the first inductive means to the output of the circuit. The circuit also includes first capacitive means coupled to the output, and first rectifying means connected between the first inductive means and the first capacitive means. The first capacitive means provides an output voltage and is charged when the first electronic switch is in a non-conductive state. When the first electronic switch is in a non-conducting state, the first rectifying means allows forward current to flow to the first capacitive means, and when the switch is in a conducting state, the first rectifying means prevents reverse current from flowing from the first capacitive means .

The circuit of the present invention also includes means for discharging the parasitic capacitance of the first electronic switch, said means comprising second inductive means connected between the first inductive means and the first rectifying means and a second inductive means coupled to the second inductive means. electronic switch. When the first electronic switch is in a non-conductive state, the second electronic switch is periodically switched to a conductive state so as to send forward current from the first rectifying means to the second inductive means. The parasitic capacitance is then discharged by the flow of current from the first electronic switch to the second inductive means. When the second electronic switch is turned on, the second inductive means limits the forward current from the first inductive means and the reverse recovery current from the first rectifying means.

In another aspect, the present invention provides a method for use in a boost converter circuit comprising a first inductive element and a first electronic switch. A first switch having a parasitic capacitance is periodically turned on and off to control the flow of forward current to the first rectifying element. The method of the present invention discharges the parasitic capacitance of the first switch while minimizing turn-on losses in the first switch and losses due to reverse recovery current from the first rectifying element. The method includes the step of directing forward current from the first inductive element and reverse recovery current from the rectifying element to the second inductive element when the first switch is turned off to allow the parasitic capacitance to discharge. The method also includes the step of directing current from the second inductive means to the second rectifying means when the first switch is turned on. Turning on the second electronic switch leads the forward current from the first inductive element and the reverse recovery current from the rectifying element to the second inductive device, and turning off the second electronic switch directs the current from the second inductive element lead to the second rectifying element.

Figure overview

For a deeper understanding of the present invention and for other objects and advantages of the present invention, reference is now made to the following description in conjunction with the accompanying drawings, wherein:

Fig. 1 is a schematic circuit diagram of a boost converter circuit of the prior art;

2A-2D are voltage and current waveforms depicting turn-on power losses in the circuit of FIG. 1;

Fig. 3 is a schematic circuit diagram of a first boost converter circuit incorporating the principles of the present invention;

4A-4F depict voltage and current waveforms for switching cycles in the circuit of FIG. 3;

5 is a schematic circuit diagram of a second boost converter circuit incorporating the principles of the present invention;

6A-6F are voltage and current waveforms depicting switching cycles in the circuit of FIG. 5 .

BEST MODE FOR CARRYING OUT THE INVENTION

Referring first to FIG. 3, there is shown a boost converter circuit 300, a first embodiment constructed in accordance with the principles of the present invention. Circuit 300 includes MOSFET power transistor (MOSFET) 302, inductor 304, diode 306, capacitor 308, MOSFET power transistor (MOSFET) 310, inductor 312, saturable inductor 314, diode 316, diode 318, diode 320, and resistor 322 . A voltage source (Vin) 324 is connected to the input of the circuit 300 and a load 326 is connected to the output of the circuit 300 . The gate of MOSFET 302 is connected to a pulsed switch voltage source (Vswitch) 328 and the gate of MOSFET 310 is connected to an auxiliary switch voltage source (Vswitch) 330 . Vswitch 328 applies a pulsed voltage to the gate of MOSFET 302, turning MOSFET 302 on and off. Vswitch 330 applies a pulse voltage to the gate of MOSFET 310, turning MOSFET 310 on and off. 3 also shows current IL304 flowing through inductor 304, current ID306 flowing through diode 306, current IL312 flowing through inductor 312, current ID320 flowing through diode 320, current IR322 flowing through resistor 322, etc. and a voltage VDS302 across the drain-source of the MOS-FET 302, a voltage VDS310 across the drain-source of the MOSFET 310, a voltage VC308 across the capacitor 308, a voltage VD316 across the diode 316, and a voltage across the load 326 Voltage mark such as voltage VL.

In operation, circuit 300 acts like a conventional boost converter while minimizing losses due to reverse recovery current in diode 306 and turn-on losses in MOSFET 302. Circuit 300 is also designed such that losses due to reverse recovery current in diode 320 and turn-on losses in MOSFET 310 are minimized.

Circuit 300 uses a resonant switching technique whereby the drain-to-source capacitance of MOSFET 302 is discharged in resonant mode through MOSFET 310 before MOSFET 302 is turned on. Then, MOSFET 302 is turned on when VDS 302 is equal to zero. Resonant switching is only used during the turn-on phase of the MOSFET 302 switching cycle. Turn-on losses in MOSFET 310 are minimized by utilizing inductor 314 to limit the rise of current in MOSFET 310 at turn-on.

The operation of circuit 300 can be better understood with reference to the switching cycle waveforms shown in FIGS. 4A-4F. FIG. 4A shows the waveform of drain-source voltage VDS302 on MOSFET 302. FIG. 4B shows the waveform of the sum of the drain-source voltage on the MOSFET 310 and the voltage across the auxiliary diode 316 VDS310+VD316. FIG. 4C shows the waveform of current ID 306 flowing through diode 306 . FIG. 4D shows the waveform of the current IL312 flowing through the auxiliary inductor 312 . FIG. 4E shows the waveform of the voltage Vswitch 328 applied to the gate of MOSFET 302. FIG. 4F shows the waveform of the voltage Vswitch 330 applied to the gate of MOSFET 310. In FIGS. 4A-4F , it can be seen that the switching cycle of the circuit 300 is divided into six periods I-VI.

3-4, during period I, both Vswitch 328 and Vswitch 330 are zero, and MOSFET 302 and MOSFET 310 are off. Voltages VDS302 and VDS310+VD316 are the sum of the load voltage and the voltage drop across diode 306 . At this time, current IL304 in inductor 304 is equal to current ID306 in diode 306 and flows to capacitor 308 and load 326 . When period II begins, Vswitch 330 is pulsed to turn on MOSFET 310. VDS310+VD316 falls to zero, current IL312 in inductor 312 rises, and current ID306 in diode 306 falls. Since the inductance of the inductor 304 is relatively large, for example, on the order of 1 millihenry, the current IL 304 will remain constant during the time the MOSFET 310 is on. Therefore, as current IL312 in inductor 312 rises, current ID306 in diode 306 will fall at the same rate. Saturable inductor 314 initially limits the rise of current IL 312 in inductor 312, which flows through MOSFET 310 to ground. This limits the turn-on losses of MOSFET 310. Eventually, saturable inductor 314 saturates, and the rate of rise of current IL 312 in inductor 312 will depend on the inductance value of inductor 312 . When current IL312 in inductor 312 equals current IL304 through inductor 304, current ID306 becomes negative due to the reverse recovery current in diode 306 when diode 306 was turned off. The reverse recovery current is limited by the rate at which current ID306 falls. The rate at which ID 306 falls is determined by the inductance value of inductor 312 . This negative spike in ID 306 will raise current IL 312 in inductor 312 to a value higher than the current in inductor 304 until the diode is completely disconnected.

Period III begins when diode 306 is turned off. The current IL 312 in the inductor 312 is now greater than the current flowing through the inductor 304 . The portion of the current IL312 in the inductor 312 that exceeds the current IL304 flowing through the inductor 304 flows out of the MOSFET 302 through the inductor 312 so that the current IL312 flowing through the inductor 312 remains constant. This excess current discharges the drain-source capacitance of MOSFET 302. When the drain-source capacitance is discharged, the diode of the MOS-FET 302 will conduct the excess current. When VDS 302 goes to zero, Vswitch 328 is pulsed to turn on MOSFET 302. Since the drain-source capacitance is already discharged, there will be no drain-source capacitance loss on turn-on.

At the beginning of period IV, MOSFET 310 is turned off. Current IL 312 in inductor 312 is commutated to diode 320 and capacitor 308 . The current IL 312 in the inductor 312 will drop at a certain rate depending on the output voltage and the inductance value of the inductor 312 . At the same time, the current in MOSFET 302 will increase so that the total current remains constant, equal to the current IL304 in inductor 304. When current IL312 in inductor 312 approaches zero, inductor 314 does not saturate and the rate of fall of IL312 slows further. Due to the reverse recovery current in diode 320, IL 312 exhibits a negative current spike when diode 320 is reverse biased and begins to turn off. This reverse recovery current will be limited by inductor 314 .

Period V begins when diode 320 is turned off. At this point, negative current ID 320 commutates and flows from ground as current IR 322 through resistor 322 , diode 318 and inductors 312 and 314 . Diode 316 prevents current from flowing into inductor 312 from ground through the body diode of MOSFET 310. The current IL 312 in the inductor 312 now falls at a rate dependent on the resistance value of the resistor 322 . The resistive damping effect of resistor 322 prevents excessive ringing in inductor 312 , inductor 314 , diode 318 , and diode 306 . This prevents excessive voltage across diode 316 and diode 320 when diode 320 is turned off.

The cycle ends in period VI when Vswitch 328 is set to zero to turn MOSFET 302 off. The current in MOSFET 302 is commutated to diode 306, which then conducts and current ID 306 in diode 306 rises. The next switching cycle then begins at period I and continues as described above.

Figure 5 shows a second embodiment, a boost converter circuit 500, constructed in accordance with the present invention. Circuit 500 includes MOSFET power transistor (MOSFET) 502, inductor 504, diode 506, capacitor 508, switching MOSFET power transistor (MOSFET) 510, inductor 512, saturable inductor 514, diode 516, diode 518, diode 520, resistor 522 , diode 534 and capacitor 536 . A voltage source (Vin) 524 is connected to the input of the circuit 500 and a load 526 is connected to the output of the circuit 500 . Vswitch 528 applies a pulse voltage to the gate of MOSFET 502 to switch MOSFET 502 on and off, and Vswitch 530 applies a pulse voltage to the gate of MOS-FET 510 to switch MOSFET 510 on and off.

5 also shows current IL504 flowing through inductor 504, current ID506 flowing through diode 506, current IL512 flowing through inductor 512, current ID520 flowing through diode 520, current IR522 flowing through resistor 522, and current Current marks such as current ID534 through diode 534; and voltage VDS502 across MOSFET 502 drain-source, voltage VC508 across capacitor 508, voltage VDS510 across MOSFET 510 drain-source, voltage VC536 across capacitor 536 , the voltage VD516 across the diode 516 and the voltage VL across the load 526 and other voltage marks.

In operation, circuit 500 acts like a conventional boost converter while minimizing losses due to reverse recovery current in diode 506 and turn-on and turn-off losses in MOSFET 502 in accordance with the present invention. Circuit 500 is also designed such that losses due to reverse recovery current in diode 520 and turn-on and turn-off losses in MOSFET 510 are minimized.

Circuit 500 uses a resonant switching technique by which the drain-to-source capacitance of MOSFET 502 is discharged in resonant mode through MOSFET 510 before MOSFET 502 is turned on. Resonant switching is only used during the MOSFET 502 turn-on phase of the switching cycle. During the time MOSFET 502 is off, the voltage across MOSFET 502 is minimized by voltage VC536 across capacitor 536. Turn-on losses in MOSFET 510 are minimized by utilizing inductor 514 to limit the current flowing through MOSFET 510 at turn-on. The turn-off losses of MOSFET 510 are minimized with voltage VC536 across capacitor 536.

FIG. 6A shows the waveform of drain-source voltage VDS502 on MOSFET 502. FIG. 6B shows the waveform of the sum of the drain-source voltage on MOSFET 510 and the voltage across auxiliary diode 516 VDS510+VD516. FIG. 6C shows the waveform of current ID506 flowing through diode 506 . FIG. 6D shows the waveform of the current IL512 flowing through the auxiliary inductor 512 . FIG. 6E shows the waveform of the voltage Vswitch 528 applied to the gate of MOSFET 502. FIG. 6F shows the waveform of the voltage Vswitch 530 applied to the gate of MOSFET 510. The switching cycle of the circuit in Fig. 5 can be divided into six periods I-VI.

During periods I-III, elements 502-522 in circuit 500 of FIG. 5 substantially function the same as elements 302-322 in FIG. 3, respectively. During periods IV-VI, the operation of circuit 500 is essentially the same as the corresponding operation of circuit 300 described above, except that the disconnection in MOSFETs 502 and 510 is due to the voltage conditions present on VC 536 when either MOSFET 502 or MOSFET 510 is disconnected. The loss is minimal. The operation of the circuit 500 during the period IV-VI is described below.

Referring to Figures 5-6, at the beginning of period IV, Vswitch 530 switches to zero and MOSFET 510 turns off. Current IL512 in inductor 512 now charges capacitor 536 through diode 520 . Capacitor 536 begins to discharge and VC536 is zero. When capacitor 536 is charging, the voltage VDS510 across MOSFET 510 is equal to the voltage VC536 across capacitor 536. Turn-off losses in MOSFET 510 are minimized since VDS510 and VD516 are zero when off. This makes the rising rate of VDS510 and VD516 during period IV in FIG. 6 slower than that of VDS310 and VD316 during period IV in FIG. 4 . When capacitor 536 charges to the output voltage, current IL512 in inductor 512 flows into diode 534 as current ID534, and diode 534 becomes forward biased and turns on. Now the current IL512 in the inductor 512 drops at a certain rate according to the output voltage VL and the inductance of the inductor 512 . When the current IL512 in the inductor 512 approaches zero, the inductor 514 no longer saturates and further slows down the rate of decline of IL512. Due to the reverse recovery current in diode 520 and diode 534, current IL512 exhibits a negative current spike when diode 520 and diode 534 become reverse biased and begin to turn off. The reverse recovery current will then be limited by the inductor 514 .

At the beginning of period V, diode 520 and diode 534 are disconnected, and a small reverse current IL512 in inductor 512 commutates and flows as current IR522 from ground, through resistor 522, diode 518 and inductors 514 and 512, and finally into MOSFET502. Diode 516 prevents current from flowing into inductor 512 from ground through the body diode of MOSFET 510. Current IL512 falls at a rate of fall that depends on the resistance value of resistor 522 . The damping effect of resistor 522 prevents excessive voltage across diode 516 and diode 520 when diode 520 is off. Current now flows through inductor 504 and MOSFET 502, and the voltage VC536 across capacitor 536 is equal to the output voltage.

At the beginning of period VI, Vswitch 528 goes to zero and MOSFET 502 turns off. VDS502 is now equal to VL minus the voltage VC536 across capacitor 536 and is close to zero. Turn-off losses in MOSFET 502 are thus minimized. Current IL504 in inductor 504 discharges capacitor 536 to zero through diode 534 . This makes the rising rate of VDS502 in period VI in FIG. 6 slightly slower than the rising rate of VDS302 in period VI in FIG. 4 . When capacitor 536 is fully discharged, diode 506 will turn on and diode 534 will turn off. The switching cycle is now complete.

The following is a table illustrating typical industry standard components and circuit parameters that can be used to build and operate the boost converter circuits of Figures 3 and 5.

MOSFETs 302, 502 IRF460 MOSFETs 310, 510 IRF840 Diodes 306, 506 APT30D60B Diodes 316, 516   Philips BYM26C Diodes 318, 518   Philips BYM26C Diode 320, 520   Philips BYM26C Diode 534   Philips BYM26C Resistor 322, 522 20 ohms Inductor 304, 504 1 millihenry Inductor 312, 512 4 microhenries Inductor 314, 514 6 turns wound on Toshiba SA14×8×4.5 Capacitor 536 6.8 nanofarads Capacitor 308, 508 C = 1 millifarad (variable) Vout 400 volts Vin                                   On-off level 50 kHz

Those skilled in the art will appreciate that these component values are exemplary and that the circuits of FIGS. 3 and 5 can be implemented with many different component values and circuit parameters. It will also be apparent that various changes may be made in the structural matter shown in and discussed in connection with the drawings without departing from the spirit and scope of the invention. Accordingly, the invention should not be considered limited to what is shown and described.

Claims (18)

1. a step-up converter circuit is characterized in that, comprising:

First inductance device is used to receive the forward current that offers described circuit input end;

First electronic switch, it links to each other with described first inductance device, described first electronic switch has intrinsic parasitic capacitance and the cycle of doing between conducting state and nonconducting state switches, wherein under conducting state, forward current flows through described switch from described first inductance device, under nonconducting state, forward current flow to the output of described circuit from described first inductance device;

First capacitive means, it links to each other with the described output that output voltage is provided, and when described first electronic switch was in nonconducting state, described capacitive means was recharged;

First rectifying device, it is connected between described first inductance device and described first capacitive means, when described first electronic switch is in nonconducting state, described first rectifying device makes forward current flow to described first capacitive means, and when described switch was in conducting state, described first rectifying device stoped reverse current to flow out from described first capacitive means; With

Be used for making the device of the intrinsic parasitic capacitance discharge of described first electronic switch, it comprises:

Second inductance device, it is connected between described first inductance device and described first rectifying device; With

Second electronic switch, it links to each other with described second inductance device, when described first electronic switch is in nonconducting state, described second electronic switch switches to conducting state periodically, thereby make forward current branch to described second inductance device from described first rectifying device, described parasitic capacitance is by flowing to the current discharge of described second inductance device from described first electronic switch.

2. step-up converter circuit as claimed in claim 1, it is characterized in that, when the described second electronic switch conducting, described second inductance device limits forward current that flows out from described first inductance device and the reverse recovery current that flows out from described first rectifying device.

3. step-up converter circuit as claimed in claim 1 is characterized in that, described first inductance device has than the bigger inductance of described second inductance device.

4. step-up converter circuit as claimed in claim 1 is characterized in that described second inductance device comprises the tandem compound of an inductor and a saturated inductor.

5. step-up converter circuit as claimed in claim 1 is characterized in that, the described device that is used to discharge further comprises:

Second rectifying device, it is connected between described second inductance device and described second electronic switch;

The 3rd rectifying device, it links to each other with described first rectifying device with described second inductance device; With

The 4th rectifying device, it links to each other with the 3rd rectifying device with described second.

6. step-up converter circuit as claimed in claim 5 is characterized in that,

Described first and second electronic switches all comprise a MOSFET power transistor;

The described first, second, third and the 4th rectifying device all comprises a diode; And

Described first capacitive means comprises a capacitor.

7. step-up converter circuit as claimed in claim 1 is characterized in that, the described device that is used to discharge also comprises:

Second capacitive means, it links to each other with described second inductance device;

Second rectifying device, it is connected between described second inductance device and described second electronic switch;

The 3rd rectifying device, it is connected between described second inductance device and described second capacitive means;

The 4th rectifying device, it links to each other with the 3rd rectifying device with described second; With

The 5th rectifying device, it is connected between described first rectifying device and second capacitive means.

8. step-up converter circuit as claimed in claim 7 is characterized in that,

Described first and second electronic switches all comprise a MOSFET power transistor;

The described first, second, third, fourth and the 5th rectifying device all comprises a diode; And

Described first and second capacitive means comprise a capacitor.

9. method of in the step-up converter circuit that comprises first inductance element and first electronic switch, using, wherein said first electronic switch is used to control forward current and flows to first rectifier cell from described first inductance element, described first switch has parasitic capacitance and switches on and off periodically, described method is used to make the parasitic capacitance discharge of described first switch, simultaneously the connection loss in described first switch and from the reverse recovery current of described first rectifier cell caused loss reduce to minimum, it is characterized in that described method comprises the following steps:

When described first switch disconnects, will cause to second inductance element from the forward current of described first inductance element with from the reverse recovery current of described first rectifier cell, thereby make described parasitic capacitance discharge; With

When described first switch connection, will guide to second rectifier cell from the electric current of described second inductance element.

10. method as claimed in claim 9, it is characterized in that, connect second electronic switch, causing described second inductance element from the forward current of described first inductance element with from the reverse recovery current of described first rectifier cell, and disconnect second electronic switch, will cause described second rectifier cell from the electric current of described second inductance element.

11. method as claimed in claim 10 is characterized in that, described second inductance element comprises the tandem compound of an inductor and a saturated inductor.

12. method as claimed in claim 10 is characterized in that, described first inductance element has than the bigger inductance of described second inductance element.

13. method as claimed in claim 10 is characterized in that, described second rectifier cell links to each other with first capacity cell, and when each described electronic switch disconnected respectively, the restriction of described first capacity cell was across described first and the voltage of described second electronic switch.

14. method as claimed in claim 13 is characterized in that, the 3rd rectifier cell is connected between described first rectifier cell and described first capacity cell.

15. method as claimed in claim 14 is characterized in that, described first links to each other with second capacity cell with the output of the 3rd rectifier cell at described circuit.

16. method as claimed in claim 10, it is characterized in that, described the 3rd rectifier cell links to each other with described second inductance element with described second rectifier cell, when described second switch disconnects, described the 3rd rectifier cell causes described second inductance element with electric current from ground, in order to the decline of compensation from the reverse recovery current of described second rectifier cell.

17. a step-up converter circuit is characterized in that, comprising:

First inductor, it has first end and second end, and described first end links to each other with the input of described circuit;

First switch, it has conducting state and nonconducting state, when described first switch is in conducting state, second end of described first inductor and ground is connected, and when described first switch was in nonconducting state, it disconnected second end of described first inductor and ground;

First rectifier, it has first end and second end, and first end of described first rectifier links to each other with second end of described first inductor, and second end of described first rectifier links to each other with the output of described circuit;

One capacitor, it has first end and second end, and first end of described first capacitor links to each other with the output of described circuit, and the second end ground connection of described capacitor;

Second inductor, it has first end and second end, and first end of described second inductor links to each other with second end of described first inductor;

The 3rd inductor, it has first end and second end, and first end of described the 3rd inductor links to each other with second end of described second inductor;

Second rectifier, it has first end and second end, and first end of described second rectifier links to each other with second end of described the 3rd inductor, and second end of described second rectifier links to each other with the output of described circuit;

The 3rd rectifier, it has first end and second end, and first end of described the 3rd rectifier links to each other with second end of described the 3rd inductor;

Second switch, it has conducting state and nonconducting state, when described second switch is in conducting state, second end of described the 3rd rectifier and ground is connected, and when described second switch was in to conducting state, it disconnected second end of described the 3rd rectifier and ground;

The 4th rectifier, it has first end and second end, and second end of described the 4th rectifier links to each other with first end of described the 3rd rectifier; With

One resistor, it has first end and second end, and first end of described resistor links to each other with first end of described the 4th rectifier, and the second end ground connection of described resistor.

18. a step-up converter circuit is characterized in that, comprising:

First inductor, it has first end and second end, and described first end links to each other with the input of described circuit;

First switch, it has conducting state and nonconducting state, when described first switch is in conducting state, second end of described first inductor and ground is connected, and when described first switch was in nonconducting state, it disconnected second end of described first inductor and ground;

First rectifier, it has first end and second end, and first end of described first rectifier links to each other with second end of described first inductor, and second end of described first rectifier links to each other with the output of described circuit;

First capacitor, it has first end and second end, and first end of described capacitor links to each other with the output of described circuit, and the second end ground connection of described first capacitor;

Second inductor, it has first end and second end, and first end of described second inductor links to each other with second end of described first inductor;

The 3rd inductor, it has first end and second end, and first end of described the 3rd inductor links to each other with second end of described second inductor;

Second rectifier, it has first end and second end, and first end of described second rectifier links to each other with second end of described the 3rd inductor;

The 3rd rectifier, it has first end and second end, and first end of described the 3rd rectifier links to each other with second end of described the 3rd inductor;

Second switch, it has conducting state and nonconducting state, when described second switch is in conducting state, second end of described the 3rd rectifier and ground is connected, and when described second switch was in nonconducting state, it disconnected second end of described the 3rd rectifier and ground;

The 4th rectifier, it has first end and second end, and second end of described the 4th rectifier links to each other with first end of described second rectifier; With

One resistor, it has first end and second end, and first end of described resistor links to each other with first end of described the 4th rectifier, and the second end ground connection of described resistor.

Second capacitor, it has first end and second end, and first end of described second capacitor links to each other with first end of described first rectifier, and second end of described second capacitor links to each other with second end of described second rectifier; With

The 5th rectifier, it has first end and second end, and first end of described the 5th rectifier links to each other with second end of described second capacitor, and second end of described the 5th rectifier links to each other with the output of described circuit.

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