CN111384477B - Broadband phase shifter and phase array module using same - Google Patents

Broadband phase shifter and phase array module using same Download PDF

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CN111384477B
CN111384477B CN201811613923.9A CN201811613923A CN111384477B CN 111384477 B CN111384477 B CN 111384477B CN 201811613923 A CN201811613923 A CN 201811613923A CN 111384477 B CN111384477 B CN 111384477B
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switch
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array
phase
capacitor
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CN111384477A (en
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林彦亨
蔡作敏
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Industrial Technology Research Institute ITRI
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    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P1/00—Auxiliary devices
    • H01P1/18—Phase-shifters
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q3/00—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system
    • H01Q3/26—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture
    • H01Q3/30—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture varying the relative phase between the radiating elements of an array
    • H01Q3/34—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture varying the relative phase between the radiating elements of an array by electrical means
    • H01Q3/36—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture varying the relative phase between the radiating elements of an array by electrical means with variable phase-shifters

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Abstract

The invention discloses a broadband phase shifter and a phase array module using the same. The phase shifter includes at least one phase shifting unit. The phase shift unit comprises a first switch, a first capacitor, a second capacitor, a first inductor, a second switch, a second inductor, a first resistor and a second resistor. The first switch has a first end, a second end and a third end. The first capacitor is connected between the first inductor and the second end of the first switch. The second capacitor is connected between the first inductor and the third end of the first switch. The second switch has a first end, a second end and a third end. The second end of the second switch is connected to a ground terminal. Two ends of the second inductor are respectively connected to a grounding end and a third end of the second switch. The first inductor is connected between the first capacitor and the third end of the second switch.

Description

Broadband phase shifter and phase array module using same
Technical Field
The disclosure relates to a broadband phase shifter and a phase array module using the same.
Background
With the development of communication technology, various antenna modules have been developed. The phased array module used by the antenna module contains a phase shifter. The frequencies suitable for 5G communication are, for example, 28GHz and 39GHz, and the frequency band in which the phase shifter is applicable must be wide enough for the antenna module to be applicable at both frequencies.
To achieve a sufficiently wide frequency band, it is necessary to maintain a sufficiently low root mean square phase error (RMS phase error) and root mean square gain error (RMS gain error) throughout the frequency band. The rms phase error is the average of the phase errors of the phase shifters. The rms gain error is the average error value of the phase shifter feed loss. For example, the rms phase error is calculated according to the following equation (1), and the rms gain error is calculated according to the following equation (2).
Figure BDA0001925423750000011
Figure BDA0001925423750000012
Where N is the number of states, i is the number of states, θΔiError of i-th state from ideal state, AiIs the feed loss of the i-th state, AAvgThe average value of the feed loss for all states.
The conventional switched 5-bit phase shifter requires 5 phase shift units of different angles, and 32 different phase states can be formed by switching on (on) state/off (off) state of each phase shift unit.
However, in the conventional phase shifter, it is a difficult challenge to maintain a sufficiently small rms phase error and a sufficiently small rms gain error in the whole frequency band.
Therefore, how to design a phase shifter applicable to a wide frequency band has become a main research direction for future development.
Disclosure of Invention
The disclosure relates to a broadband phase shifter and a phase array module using the same.
According to an embodiment of the present disclosure, a phase shifter is provided. The phase shifter includes at least one phase shifting unit. The phase shift unit comprises a first switch, a first capacitor, a second capacitor, a first inductor, a second switch, a second inductor, a first resistor and a second resistor. The first switch has a first end, a second end and a third end. The first capacitor is connected between the first inductor and the second end of the first switch. The second capacitor is connected between the first inductor and the third end of the first switch. The second switch has a first end, a second end and a third end. The second terminal of the second switch is connected to a ground terminal. Two ends of the second inductor are respectively connected to a grounding end and a third end of the second switch. The first inductor is connected between the first capacitor and the third end of the second switch. The first resistor is connected between the first end of the first switch and a first control voltage. The second resistor is connected between the first end of the second switch and a second control voltage. The first control voltage is a reverse voltage of the second control voltage.
According to an embodiment of the present disclosure, a phase array module is provided. The phased array module includes a power divider, a tunable attenuator array, a first switch array, a power amplifier array, a low noise amplifier array, a second switch array, and a phase shift array. The adjustable attenuator array is connected with the power divider. The phase shift array is connected to the adjustable attenuator array. The phase shift array includes a plurality of phase shifters. One of the phase shifters includes at least one phase shifting unit. The phase shift unit comprises a first switch, a first capacitor, a second capacitor, a first inductor, a second switch, a second inductor, a first resistor and a second resistor. The first switch has a first end, a second end and a third end. The first capacitor is connected between the first inductor and the second end of the first switch. The second capacitor is connected between the first inductor and the third end of the first switch. The second switch has a first end, a second end and a third end. The second terminal of the second switch is connected to a ground terminal. Two ends of the second inductor are respectively connected to a grounding end and a third end of the second switch. The first inductor is connected between the first capacitor and the third end of the second switch. The first resistor is connected between the first end of the first switch and a first control voltage. The second resistor is connected between the first end of the second switch and a second control voltage. The first control voltage is a reverse voltage of the second control voltage. The first switch array is connected to the phase shift array. The power amplifier array is connected to the first switch array. The low noise amplifier array is connected to the first switch array. The second switch array is connected to the power amplifier array and the low noise amplifier array.
For a better understanding of the above and other aspects of the disclosure, reference should be made to the following detailed description of the embodiments, which is to be read in connection with the accompanying drawings:
drawings
Fig. 1 is a schematic diagram of a phased array module according to an embodiment.
Fig. 2 shows a schematic diagram of a phase shifter according to an embodiment.
Fig. 3 illustrates a relative phase curve for a 22.5 degree phase shift unit.
Fig. 4 illustrates the feed loss curves of a 22.5 degree phase shift cell at turn-on and turn-off, respectively.
FIG. 5 is a schematic diagram of a phase shift unit according to an embodiment.
Fig. 6 is an equivalent circuit diagram of the phase shift unit with the first control voltage at a low voltage level and the second control voltage at a high voltage level.
Fig. 7 is an equivalent circuit diagram of the phase shift unit with the first control voltage at a high voltage level and the second control voltage at a low voltage level.
FIG. 8 shows the relative phase curve of the 11.25 degree phase shift unit of FIG. 5 and the relative phase curve of the conventional 11.25 degree phase shift unit.
FIG. 9A shows the feed loss curve of the 11.25 degree phase shift unit of FIG. 5 and the feed loss curve of the conventional 11.25 degree phase shift unit.
FIG. 9B shows a difference curve of the feeding loss of the 11.25 degree phase shift unit of FIG. 5 and a difference curve of the feeding loss of the conventional 11.25 degree phase shift unit.
FIG. 10 shows the relative phase curve of the 22.5 degree phase shift unit of FIG. 5 and the relative phase curve of the conventional 22.5 degree phase shift unit.
FIG. 11A shows the feeding loss curve of the 22.5 degree phase shift unit of FIG. 5 and the feeding loss curve of the conventional 22.5 degree phase shift unit.
Fig. 11B shows a difference curve of the feed loss of the 22.5 degree phase shifter according to fig. 5 and a difference curve of the feed loss of the conventional 22.5 degree phase shifter.
FIG. 12 is a diagram of a substrate terminal connection resistor of the transistor.
Fig. 13 is a graph of the feed loss versus frequency for the transistor of fig. 12 versus the feed loss versus frequency for a conventional transistor.
Fig. 14 is a graph of the feed loss versus transistor width for the transistor of fig. 12 versus the feed loss versus transistor width for a conventional transistor.
FIG. 15 shows a schematic diagram of a phase shift unit employing the design of FIG. 12.
Fig. 16 shows a feed loss curve of the 22.5 degree phase shift unit according to fig. 5 and a feed loss curve of the 22.5 degree phase shift unit according to fig. 15.
FIG. 17 shows a schematic diagram of a phase shifter according to an embodiment.
Fig. 18 shows the average feed loss curve and the rms gain error curve of the phase shifter.
FIG. 19 shows the relative phase curve and the RMS phase error curve of the phase shifter.
[ notation ] to show
100: power divider
200: phase shift array
210. 710: phase shifter
211. 211', 711, 712, 713, 714, 715: phase shift unit
300: antenna array
310: antenna with a shield
400: adjustable attenuator array
410: adjustable attenuator
500: power amplifier array
510: power amplifier
600: low noise amplifier array
610: low noise amplifier
700: first switch array
710: first switch
800: second switch array
810: second switch
1000: phased array module
b1, b 2: fourth terminal
b 12: base end
C1: first capacitor
C2: second capacitor
C3, C81, C82, C101, C102: relative phase curve
C41, C42, C91, C92, C93, C94, C111, C112, C113, C114, C161, C162: feed loss curve
C95, C96, C115, C116: feed loss difference curve
C131, C132: feed loss versus frequency curve
C141, C142: feed-in loss versus transistor width
C181: average feed loss curve
C182: root mean square gain error curve
C191: relative phase curve
C192: root mean square phase error curve
d1, d 2: third terminal
g1, g 2: first end
GN1, GN 2: grounding point
L1: first inductor
L2: second inductor
M1: first switch
M2: second switch
M12: transistor with a metal gate electrode
PH51, PH 52: current path
R1: a first resistor
R2: second resistance
R3: third resistance
R4: fourth resistor
R12: resistance (RC)
s1, s 2: second end
VC: a first control voltage
VC': second control voltage
Detailed Description
Referring to fig. 1, a schematic diagram of a phased array module 1000 according to an embodiment is shown. The phased array module 1000 comprises a power divider 100, a variable attenuator array 400, a phase shift array 200, a first switch array 700, a Power Amplifier (PA) array 500, a Low Noise Amplifier (LNA) array 600, and a second switch array 800.
The adjustable attenuator array 400 is connected to the power divider 100. The adjustable attenuator array 400 is composed of a plurality of adjustable attenuators 410. The phase shift array 200 is connected to the adjustable attenuator array 400. For example, the phase shift array 200 is composed of a plurality of phase shifters 210. The power divider 100 has a plurality of output terminals, and each of the adjustable attenuator 410 and the phase shifter 210 is connected to one of the output terminals of the power divider 100. The first switch array 700 is connected to the phase shift array 200. The first switch array 710 is composed of a plurality of first switches 710. Each first switch 710 is connected to one phase shifter 210. The power amplifier array 500 is connected to a first switch array 700. The low noise amplifier array 600 is connected to the first switch array 700. The power amplifier array 500 is composed of a plurality of power amplifiers 510. The lna array 600 is composed of several lnas 610. Each power amplifier 510 and each low noise amplifier 610 is connected to a first switch 710. The second switch array 800 is connected to the power amplifier array 500 and the lna array 600. The second switch array 800 is composed of a plurality of second switches 810. Each second switch 810 is connected to one power amplifier 510 and one low noise amplifier 610. The antenna array 300 is connected to the second switch array 800. The antenna array 300 is comprised of a plurality of antennas 310. Each antenna 310 is connected to a second switch 810.
Referring to fig. 2, a schematic diagram of a phase shifter 210 according to an embodiment is shown. The phase shifter 210 is composed of a plurality of phase shift units 211, for example. The phase shift angle of the phase shift unit 211 may be different, for example, 11.25 degrees, 22.5 degrees, 45 degrees, 90 degrees, 180 degrees. Each phase shifting unit 211 may be individually turned on or off to constitute various phase shifting angles. In order to increase the frequency band to which the phase shifter 210 can be applied, the relative phase error of each phase shift unit 211 within the frequency band and the feed loss difference of the state switching become important.
As shown in fig. 3, which illustrates a relative phase curve C3 for a 22.5 degree phase shift unit 211. The relative phase refers to a phase in which the on state of the phase shift unit 211 corresponds to a phase shift of a different frequency, which is relative to the off state. As the relative phase error of the phase shift unit 211 can be controlled smaller (the relative phase curve C3 is more gradual), the root mean square phase error (RMS phase error) of the overall phase shifter 210 will also be smaller.
As shown in fig. 4, the feeding loss curves C41 and C42 of the 22.5 degree phase shift unit 211 in the on state and the off state are illustrated. The feed loss difference refers to the difference of the feed loss curves C41, C42. As the feed loss difference of the phase shift unit 211 can be controlled smaller, the root mean square gain error (RMS gain error) of the whole phase shifter 210 will also be smaller.
The following disclosure proposes the design of the phase shift unit 211 to achieve the goal of maintaining a lower relative phase error and a lower feeding loss difference in a wider bandwidth, so that the phase shifter 210 maintains a low rms phase error and a low rms gain error in a wider bandwidth, thereby being suitable for more frequency bands.
Referring to fig. 5, a schematic diagram of a phase shift unit 211 according to an embodiment is shown. The phase shift unit 211 is, for example, a switched millimeter wave phase shift architecture. The phase shift unit 211 includes a first switch M1, a first capacitor C1, a second capacitor C2, a second switch M2, a first inductor L1, a second inductor L2, a first resistor R1, and a second resistor R2. The first switch M1 has a first terminal g1, a second terminal s1 and a third terminal d 1. The first switch M1 is, for example, an N-type metal oxide semiconductor field effect transistor (NMOS). The first terminal g1, the second terminal s1 and the third terminal d1 are, for example, a gate, a source and a drain.
The first capacitor C1 is connected between the first inductor L1 and the second terminal s1 of the first switch M1. The second capacitor C2 is connected between the first inductor L1 and the third terminal d1 of the first switch M1.
The second switch M2 has a first terminal g2, a second terminal s2 and a third terminal d 2. The second terminal s2 of the second switch M2 is connected to a ground terminal. The second switch M2 is, for example, an N-type metal oxide semiconductor field effect transistor (NMOS). The first terminal g2, the second terminal s2 and the third terminal d2 are, for example, a gate, a source and a drain.
Two ends of the second inductor L2 are connected to a ground terminal and the third end d2 of the second switch M2, respectively, and the first inductor L1 is connected between the first capacitor C1 (or the second capacitor C2) and the third end d2 of the second switch M2.
As shown in fig. 5, the first resistor R1 is connected between the first terminal g1 of the first switch M1 and a first control voltage VC. The second resistor R2 is connected between the first terminal g2 of the second switch M2 and a second control voltage VC'. The first control voltage VC is a reverse voltage of the second control voltage VC'. That is, when the first control voltage VC is at a low voltage level (lower than a threshold voltage of the first switch M1), the second control voltage VC' is at a high voltage level (higher than a threshold voltage of the second switch M2). When the first control voltage VC is at a high voltage level (higher than the threshold voltage of the first switch M1), the second control voltage VC' is at a low voltage level (lower than the threshold voltage of the second switch M2).
Through the control of the first control voltage VC and the second control voltage VC', two different phase states can be achieved. Referring to fig. 5 to 7, fig. 6 is an equivalent circuit diagram of the phase shift unit 211 in which the first control voltage VC is at a low level and the second control voltage VC 'is at a high level, and fig. 7 is an equivalent circuit diagram of the phase shift unit 211 in which the first control voltage VC is at a high level and the second control voltage VC' is at a low level. As shown in fig. 5 and 6, when the first control voltage VC is at a low voltage level, the first switch M1 is not turned on. When the second control voltage VC' is at a high voltage level, the second switch M2 is turned on, so that a current path PH52 is formed. The second terminal s1 and the third terminal d1 of the first switch M1 are disconnected, and the second switch M2 is equivalent to a small resistor, so that the on state shown in fig. 6 is formed (high pass T type circuit).
As shown in fig. 5 and 7, when the first control voltage VC is at a high voltage level, the first switch M1 is turned on, so that the current path PH51 is formed, and the first switch M1 is equivalent to a small resistor. When the second control voltage VC' is at a low voltage level, the second switch M2 is equivalent to a capacitor, and the capacitor resonates with the second inductor L2 and is regarded as an open circuit at the target frequency. The second terminal s1 and the third terminal d1 of the first switch M1 form a closed circuit therebetween, and the second terminal s2 and the third terminal d2 of the second switch M2 form an open circuit therebetween, thereby forming the closed state shown in fig. 7.
Referring to FIG. 8, a relative phase curve C81 of the 11.25 degree phase shift unit 211 of FIG. 5 and a relative phase curve C82 of the conventional 11.25 degree phase shift unit are shown. As shown in fig. 8, the relative phase curve C81 of the present disclosure is relatively flat with a relatively low relative phase error.
Referring to fig. 9A and 9B, fig. 9A illustrates the feeding loss curves C91 and C92 of the 11.25 degree phase shift unit 211 of fig. 5 and the feeding loss curves C93 and C94 of the conventional 11.25 degree phase shift unit, and fig. 9B illustrates the feeding loss difference curve C95 of the 11.25 degree phase shift unit 211 of fig. 5 and the feeding loss difference curve C96 of the conventional 11.25 degree phase shift unit. The feed loss curves C91 and C93 are the measured feed losses for the on state. The feed loss curves C92 and C94 are the measured feed losses for the off state. The difference between the feed loss curve C91 and the feed loss curve C92 is the feed loss difference curve C95; the difference between the feed loss curve C93 and the feed loss curve C94 is the feed loss difference curve C96. As shown in fig. 9B, the feed loss difference curve C95 is close to 0 over a wide frequency range, and has a relatively low feed loss difference.
Referring to fig. 10, a relative phase curve C101 of the 22.5 degree phase shift unit 211 of fig. 5 and a relative phase curve C102 of the conventional 22.5 degree phase shift unit are shown. As shown in fig. 10, the relative phase curve C101 of the present disclosure is relatively flat and has a relatively low relative phase error.
Referring to fig. 11A and 11B, fig. 11A shows the feeding loss curves C111 and C112 of the 22.5 degree phase shift unit 211 of fig. 5 and the feeding loss curves C113 and C114 of the conventional 22.5 degree phase shift unit, and fig. 11B shows the feeding loss difference curve C115 of the 22.5 degree phase shifter 210 of fig. 5 and the feeding loss difference curve C116 of the conventional 22.5 degree phase shifter. The feed loss curves C111 and C113 are the measured feed losses in the on state. The feed loss curves C112 and C114 are the measured feed losses for the off state. The difference between the feed loss curve C111 and the feed loss curve C112 is a feed loss difference curve C115; the difference between the feeding loss curve C113 and the feeding loss curve C114 is a feeding loss difference curve C116. As shown in fig. 11B, the feed loss difference curve C115 is close to 0 over a wide frequency range, and has a relatively low feed loss difference.
Referring to fig. 12, a schematic diagram of the substrate terminal b12 of the transistor M12 connected to the resistor R12 is shown. One end of the resistor R12 is connected to a ground terminal. When the substrate of the conventional transistor is directly grounded, the parasitic capacitance at the substrate affects the impedance of the transistor, and thus affects the feed-in loss of high frequency signals through the transistor. The resistor R12 is connected to the substrate terminal b12 of the transistor M12, and for high frequency signals, the resistor R12 provides an open circuit, so that the parasitic capacitance effect of the substrate terminal b12 is reduced, and the feeding loss is further reduced (hereinafter referred to as "substrate-side floating technology"). Referring to fig. 13, a curve C131 of the feeding loss versus frequency of the transistor M12 of fig. 12 and a curve C132 of the feeding loss versus frequency of the conventional transistor are shown. The substrate terminal of a conventional transistor is directly grounded. It can be seen from fig. 13 that the substrate-end floating technique can make the transistor M12 have lower feeding loss, and the higher the frequency, the better the effect.
Referring to fig. 14, a curve C141 of the transistor M12 of fig. 12 showing the relationship between the feeding loss and the transistor width at 39GHz and a curve C142 of the conventional transistor showing the relationship between the feeding loss and the transistor width at 39GHz are shown. It can be seen from fig. 14 that the larger the transistor width, the better the parasitic capacitance effect is reduced by using the coupling resistor at the base end.
Referring to fig. 15, a schematic diagram of the phase shift unit 211' designed in fig. 12 is shown. The first switch M1 has a fourth terminal b1, and the second switch M2 also has a fourth terminal b 2. The fourth terminal b1 of the first switch M1 is a base terminal, and the fourth terminal b2 of the second switch M2 is a base terminal. Compared to the phase shift unit 211 of FIG. 5, the phase shift unit 211' of FIG. 15 further includes a third resistor R3 and a fourth resistor R4. The third resistor R3 is connected between the fourth terminal b1 of the first switch M1 and a ground GN1, and the fourth resistor R4 is connected between the fourth terminal b2 of the second switch M2 and a ground GN 2.
Referring to fig. 16, the feeding loss curves C111 and C112 of the 22.5 degree phase shift unit 211 shown in fig. 5 and the feeding loss curves C161 and C162 of the 22.5 degree phase shift unit 211' shown in fig. 15 are shown. It can be seen from fig. 16 that the feed loss of the 22.5 degree phase shift unit 211' can be reduced by about 1dB, which proves effective in the technique of using coupling resistors at the base end to reduce the parasitic capacitance effect therein. Through the experiments of researchers, the phase shift units 211 'with different angles have different effects, for example, the phase shift units 211' at 45 degrees can be reduced by 1.5 dB. Also, the amount of feed loss reduction is also related to the transistor width.
The various phase shifting units 211, 211' proposed in the present disclosure above may be combined with each other into a multi-bit phase shifter. Alternatively, the various phase shifting units 211, 211' proposed above may also be combined with other kinds of phase shifting units into multi-bit phase shifters. Referring to fig. 17, a schematic diagram of a phase shifter 710 according to an embodiment is shown. The phase shifter 710 includes a 180-degree phase shift unit 711, an 11.25-degree phase shift unit 712, a 22.5-degree phase shift unit 713, a 45-degree phase shift unit 714, and a 90-degree phase shift unit 715. The 11.25 degree phase shift unit 712, the 22.5 degree phase shift unit 713, and the 45 degree phase shift unit 714 are, for example, the phase shift unit 211' of fig. 15.
Referring to fig. 18, an average feed-in loss curve C181 and an rms gain error curve C182 of the phase shifter 710 are shown. The 5-bit phase shifter 710 has 32 combination states, so that 32 sets of feed loss can be measured. The average feed loss curve C181 is the average of the feed losses for the 32 states. The rms gain error curve C182 of the phase shifter 710 can be obtained by calculating the feed loss and the average feed loss curve C181 for the 32 states. As can be seen from fig. 18, the rms gain error curve C182 can be maintained at a low value over a wide frequency range, so that the phase shifter 710 can be used in a wide frequency band.
Referring to fig. 19, a relative phase curve C191 and a root mean square phase error curve C192 of the phase shifter 710 are shown. The 5-bit phase shifter 710 has 32 combination states, so 32 relative phase curves C191 can be measured. From the calculation of the relative phase curve C191, a root mean square phase error curve C192 can be obtained. As can be seen from fig. 19, the rms phase error curve C192 of the phase shifter 710 has a relatively low value, so that the phase shifter 710 can be used in a wider frequency band.
In summary, although the present disclosure has been described with reference to the above embodiments, the disclosure is not limited thereto. Various modifications and alterations may be made by those skilled in the art without departing from the spirit and scope of the disclosure. Therefore, the protection scope of the present disclosure should be subject to the definition of the appended claims.

Claims (8)

1. A phase shifter, comprising at least one phase shifting unit, the phase shifting unit comprising:
the first switch is provided with a first end, a second end and a third end, wherein the first end, the second end and the third end of the first switch are respectively a grid end, a source end and a drain end;
a first capacitor;
a second capacitor;
the first capacitor is connected between the first inductor and the second end of the first switch, and the second capacitor is connected between the first inductor and the third end of the first switch;
the second switch is provided with a first end, a second end and a third end, wherein the first end, the second end and the third end of the second switch are respectively a grid end, a source end and a drain end, and the second end of the second switch is connected with a grounding end;
the two ends of the second inductor are respectively connected with a grounding end and the third end of the second switch, and the first inductor is connected between the first capacitor and the third end of the second switch;
a first resistor connected between the first end of the first switch and a first control voltage; and
the second resistor is connected between the first end of the second switch and a second control voltage, and the first control voltage is the reverse voltage of the second control voltage;
wherein one end of the first inductor is connected to the connection line between the first capacitor and the second capacitor, and the other end of the first inductor is connected to the third end of the second switch.
2. The phase shifter of claim 1, wherein the first switch is a field effect transistor, the first switch further having a fourth terminal, the second switch is a field effect transistor, the second switch further having a fourth terminal, the phase shift unit further comprising:
a third resistor connected to the fourth terminal of the first switch; and
and the fourth resistor is connected with the fourth end of the second switch.
3. The phase shifter of claim 2, wherein the third resistor is connected between the fourth terminal of the first switch and ground, and the fourth resistor is connected between the fourth terminal of the second switch and ground.
4. The phase shifter of claim 2, wherein the fourth terminal of the first switch is a base terminal and the fourth terminal of the second switch is a base terminal.
5. A phased array module, comprising:
a power divider;
the adjustable attenuator array is connected with the power divider;
a phase shift array connected to the adjustable attenuator array, the phase shift array including a plurality of phase shifters, one of the phase shifters including at least one phase shift unit, the phase shift unit including:
the first switch is provided with a first end, a second end and a third end, wherein the first end, the second end and the third end of the first switch are respectively a grid end, a source end and a drain end;
a first capacitor;
a second capacitor;
the first capacitor is connected between the first inductor and the second end of the first switch, and the second capacitor is connected between the first inductor and the third end of the first switch;
the second switch is provided with a first end, a second end and a third end, wherein the first end, the second end and the third end of the second switch are respectively a grid end, a source end and a drain end, and the second end of the second switch is connected with a grounding end;
the two ends of the second inductor are respectively connected with a grounding end and the third end of the second switch, and the first inductor is connected between the first capacitor and the third end of the second switch;
a first resistor connected between the first end of the first switch and a first control voltage; and
the second resistor is connected between the first end of the second switch and a second control voltage, and the first control voltage is the reverse voltage of the second control voltage;
a first switch array connected to the phase shift array;
a power amplifier array connected to the first switch array;
a low noise amplifier array connected to the first switch array; and
a second switch array connected to the power amplifier array and the low noise amplifier array;
wherein one end of the first inductor is connected to the connection line between the first capacitor and the second capacitor, and the other end of the first inductor is connected to the third end of the second switch.
6. The phased array module of claim 5, wherein the first switch is a field effect transistor, the first switch further having a fourth terminal, the second switch is a field effect transistor, the second switch further having a fourth terminal, the phase shift unit further comprising:
a third resistor connected to the fourth terminal of the first switch; and
and the fourth resistor is connected with the fourth end of the second switch.
7. The phased array module of claim 6, wherein the third resistor is connected between the open fourth terminal of the first switch and ground, and the fourth resistor is connected between the fourth terminal of the second switch and ground.
8. The phased array module of claim 6, wherein the fourth terminal of the first switch is a base terminal and the fourth terminal of the second switch is a base terminal.
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