CN111194077A - Timing synchronization method under low sampling rate - Google Patents
Timing synchronization method under low sampling rate Download PDFInfo
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- CN111194077A CN111194077A CN201911302690.5A CN201911302690A CN111194077A CN 111194077 A CN111194077 A CN 111194077A CN 201911302690 A CN201911302690 A CN 201911302690A CN 111194077 A CN111194077 A CN 111194077A
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04W—WIRELESS COMMUNICATION NETWORKS
- H04W56/00—Synchronisation arrangements
- H04W56/0055—Synchronisation arrangements determining timing error of reception due to propagation delay
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L7/00—Arrangements for synchronising receiver with transmitter
- H04L7/0016—Arrangements for synchronising receiver with transmitter correction of synchronization errors
- H04L7/002—Arrangements for synchronising receiver with transmitter correction of synchronization errors correction by interpolation
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L7/00—Arrangements for synchronising receiver with transmitter
- H04L7/02—Speed or phase control by the received code signals, the signals containing no special synchronisation information
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04W—WIRELESS COMMUNICATION NETWORKS
- H04W56/00—Synchronisation arrangements
- H04W56/0035—Synchronisation arrangements detecting errors in frequency or phase
Abstract
The invention discloses a timing synchronization method under a low sampling rate, and relates to the technical field of timing synchronization of wireless communication receivers. The method aims at the problems that the prior art has relatively high requirements on the sampling rate, obvious distortion can be introduced under the condition of meeting the low sampling rate of Nyquist distortion-free sampling, and great challenges exist in millimeter wave emerging communication with the bandwidth of up to several GHz. The invention arranges the interpolation inside the matched filter, and realizes the integration of the timing interpolation and the matched filtering by dynamically adjusting the filter coefficient of the matched filter. Wherein, the timing error extractor, the loop filter and the NCO still adopt the basic structure of the prior Gardner timing synchronization. According to experimental results, the interpolation process is integrated into matched filtering, so that distortion caused by interpolation filtering is avoided, and the method can theoretically work under the lowest Nyquist sampling rate, namely the sampling rate is equal to the signal bandwidth; the invention avoids the time delay caused by interpolation filtering and saves the corresponding multiplier resource.
Description
Technical Field
The invention belongs to the receiver timing synchronization technology in the field of wireless communication, and particularly relates to a timing synchronization method suitable for a receiver under the condition of low sampling rate.
Background
With the advent of the 5G era, wireless communication will be increasingly widely applied, not only relating to traditional mobile communication, but also playing a great role in emerging fields such as integrated traffic wireless interconnection, Internet of Things (IoT) and the like. Since the wireless communication transceiving baseband is completed in the Digital domain, after the signal received by the receiver is down-converted to the baseband, the signal needs to be discretized and converted to the Digital domain by sampling through an Analog-to-Digital Converter (ADC). However, due to the non-ideal characteristics of the ADC and the fact that the operating characteristics are susceptible to external factors such as temperature, a certain deviation, i.e., a timing error, occurs between the sampling rate and the ideal required value.
Timing synchronization, i.e. recovering the clock information synchronized with the received symbols for timing errors in the sampling process. Correct clock synchronization is the basis for correct operation of a wireless communication receiving end, is an important factor influencing the error rate of a system, and reliable data transmission cannot be carried out without an accurate timing synchronization algorithm. Algorithms for realizing timing synchronization are various and can be divided into an analog mode, a semi-digital mode and a full-digital mode. The full digital mode is to extract timing error from the digital signal sampled by the ADC and compensate, the process is relatively simple, high-speed digital signal processing is facilitated, and digital integration is facilitated.
Existing classical all-digital timing synchronization schemes are typically implemented based on a Gardner timing error recovery algorithm that utilizes an interpolation filter to achieve sampling at the optimal sampling point of a symbol, depending on the way the input signal is re-sampled. Fig. 1 shows a block diagram of a prior art, where timing synchronization needs to work under a condition of oversampling at twice a symbol rate, and mainly includes a timing error extractor, a loop filter, a Numerically Controlled Oscillator (NCO), an interpolation filter coefficient lookup table, an interpolation filter, and a matched filter.
1) Timing error extractor
The key to Gardner timing synchronization is timing error extraction, which is performed at twice the symbol rate over-sampling, the main principle being based on the symmetry of the symbol waveform. The symbol waveform needs to ensure a higher Signal-to-Noise Ratio (SNR) by the action of matched filtering, so that before timing error extraction, the sequence sampled by the ADC needs to be interpolated into a sequence with twice the symbol rate, and then the symbol waveform under twice oversampling is obtained by matched filtering at twice the symbol rate. Generally, the shaping filter at the transmitting end and the matched filter at the receiving end are both root raised cosine filters, so that the symbol waveform after matched filtering is a raised cosine waveform. The raised cosine waveform has a peak value at the optimum sampling time of the symbol and zero values at the optimum sampling time of other symbols, i.e., no intersymbol interference (ISI) occurs between symbols at the ideal symbol rate sampling.
If there is a sampling deviation, it cannot be determined whether the sampling point is early or late only by the main sampling point z (n) at the peak of the symbol. However, if the sampling deviation direction is judged by the sub-sampling points z (n-1/2) and z (n +1/2) at twice oversampling. The Gardner timing error extraction and calculation formula is
Wherein the content of the first and second substances,
representing the real part, and complex conjugate operator. Fig. 2 shows sampling diagrams in different cases by taking positive real symbols (the optimal sampling points are positive real numbers) as an example. When there is no sampling deviation, τ (n) is 0; when the sampling is advanced, τ (n)<0; when the sampling is lagging, τ (n)>0. The timing error extraction works with symbol enable since it is only calculated once per symbol.
2) Loop filter
In fig. 2, only a single symbol is used for illustrating the timing error extraction principle, and when a plurality of symbols actually exist, the timing error extraction calculation result of each symbol cannot be guaranteed to be correct because waveforms between the symbols are mutually overlapped. By adding loop filteringThe filter can smooth out this effect. A conventional loop filter is shown in fig. 3, which is substantially a PI control filter, including a P branch sensitive to input errors and an I branch insensitive to input errors. Wherein, TsRepresenting the system clock period, which can be simply considered as the ADC sampling period, the module in which one sampling point, k, is delayedi＜＜kp. Similar to timing error extraction, loop filtering also works with symbol enable.
3)NCO
The NCO is used to calculate the effective position of the interpolation point. On one hand, the NCO needs to provide an enabling signal for each symbol; on the other hand, an interpolation phase is calculated. If the accumulated phase of the n-1 th symbol main sampling point under the system clock is mn-1+μn-1Wherein m isn-1Is an integer part, mun-1Is a fractional part, i.e. has (n-1) Ti＝(mn-1+μn-1)TsWherein T isiThe accumulated phase of the nth symbol main sampling point is
Where η (n) is the output of the loop filter, according to the integer part mnKnowing the enabling position of the nth symbol interpolation; fraction μ according to fractionnThe corresponding interpolated phase can be obtained.
Note: the accumulated phase in the present invention refers to the number of sampling point cycles of the system clock corresponding to the time, and the phase refers to the fraction of the number of sampling point cycles of the time without specific description.
4) Interpolation filter coefficient lookup table and interpolation filter
The interpolation filter being dependent on the input signal x (nT)s) The interpolation is used to obtain primary sampling point data y (n) and secondary sampling point data y (n +1/2) at symbol level
InterpolationThe essence of filtering is the process of resampling the signal at a given time instant. Wherein I1、I2Determining the number of taps for interpolation filtering, ideally I1＝+∞，I2Infinity, and h (i, μ)n) Is a Sinc function at a phase of i + munThe time-of-day value taking point, which cannot be achieved in practical situations. In practical implementation, lagrange polynomial interpolation is usually adopted, and only x (m) is utilizednTs) A few sample points in the vicinity can achieve the resampling at the desired moment. Wherein the interpolation filter coefficients h (i, mu) are calculated in real time for simplicityn) The interpolation filter coefficients are typically stored as a look-up table in terms of μnThe interpolated filter coefficients are read directly from the look-up table. Lookup table size is given bynQuantization accuracy requirement and number of interpolation filter taps I1+I2+1 is jointly determined.
The interpolation of each symbol needs to obtain a main sampling point and a sub-sampling point at the same time, and the realization process of the sub-sampling point is
Wherein m isn+1/2Is an integer part, mun+1/2Are fractional parts. If m isn+1/2＝mnIf not, the input sample point needs to be correspondingly adjusted (or the interpolation coefficient is filled with zero). Here, the primary sample point and the secondary sample point may share one interpolation filter coefficient lookup table.
5) Matched filtering
Matched filtering is performed on the data at twice the oversampling level to achieve maximum SNR, which can act to suppress out-of-band noise. The specific process is
Wherein h isMF(i/2) tap coefficients at twice oversampling for the matched filter.
In fig. 1, matched filtering may be placed before interpolation filtering, however, since the ADC sampling rate is not necessarily twice the symbol rate, an extra interpolation filter outside the loop is usually added before matched filtering to convert the sampling rate to twice the symbol rate.
The main problem of the existing technical scheme is that the sampling rate is relatively high, obvious distortion may be introduced under the condition of meeting the low sampling rate of nyquist distortion-free sampling, and great challenge is existed in millimeter wave emerging communication with the bandwidth of up to several GHz.
The nyquist distortion-free sampling theorem states that the original analog signal can be strictly recovered by discrete sampling points as long as the sampling rate is greater than or equal to the signal bandwidth. In other words, in nyquist distortion-free sampling, the resampling by the interpolation filter can obtain distortion-free results at corresponding time, and from the perspective of the frequency domain, the amplitude response of the interpolation filter in the signal bandwidth is a horizontal straight line, and the phase response is linear. However, in the limit, if the sampling rate is equal to the signal bandwidth, only strict interpolation of the Sinc function can ensure that the frequency domain response in the signal bandwidth meets the above requirement, and Sinc function interpolation filtering cannot be realized in a practical system. If lagrange polynomial interpolation is used, frequency distortion occurs at the band edge portion. Considering interpolation filtering of 10 sample points, the frequency domain amplitude response for different interpolation phases is shown in fig. 4.
As in fig. 4, only at the interpolated phase μnWhen the sampling bandwidth is equal to 0, no distortion, mu, can be ensured in the sampling bandwidthnDistortion is most pronounced at 0.5. If distortion is not guaranteed within the signal bandwidth, the sampling bandwidth, i.e., the ADC sampling rate, needs to be significantly higher than the signal bandwidth, i.e., the nyquist minimum sampling rate, so that the distorted portion of the edge within the sampling bandwidth does not fall within the signal bandwidth. The higher the ADC sampling rate is, the higher the cost is, and in order to reduce the implementation cost, the requirement of reducing the ADC sampling rate to the theoretical minimum value is very urgent, especially for millimeter wave large-bandwidth communication.
In addition, the interpolation filtering and the matched filtering are convolution operations, and the prior art scheme shown in fig. 1 separately executes the interpolation filtering and the matched filtering, which consumes more multiplier resources and larger filtering delay.
Disclosure of Invention
Aiming at the defects of the prior art, the method arranges the interpolation inside the matched filter, and realizes the fusion of the timing interpolation and the matched filtering by dynamically adjusting the filter coefficient of the matched filter. Wherein, the timing error extractor, the loop filter and the NCO still adopt the basic structure of the prior Gardner timing synchronization.
The invention firstly discloses a timing synchronization method under low sampling rate, which comprises the following steps:
1) extracting a timing error according to the output result of the (n-1) th symbol,
2) the timing error is filtered by a loop and NCO to obtain the accumulated phase of a main sampling point and a secondary sampling point of timing n symbols, and the accumulated phase is determined according to the integer part m of the accumulated phasenAnd mn+1/2Generating a matched filtering enable signal;
3) accumulating fractional part mu of phase according to nth symbol primary and secondary sampling pointnAnd mun+1/2Combining the matched filter coefficient lookup table to obtain matched filter coefficients of a main sampling point and a secondary sampling point of the nth symbol;
4) and performing matched filtering on the nth symbol to obtain a corresponding main sampling point and a corresponding sub sampling point value.
As a preferred embodiment of the present invention, the integer part m of the phase is accumulated when the main sample point and the sub-sample pointnAnd mn+1/2When the sampling time is not equal, different matched filtering enables are respectively given to enable the main sampling point and the secondary sampling point to be output under different clocks.
As a preferred embodiment of the present invention, the integer part m of the phase is accumulated when the main sample point and the sub-sample pointnAnd mn+1/2When the sampling times are not equal, the access mode of a shift register in the matched filter is changed, so that the main sampling point and the secondary sampling point are output under the same clock.
As a preferred embodiment of the present invention, h is assumedMF(T) is the symbol period T in timeiThe normalized standard matched filter has a wave form with an accurate matched filter coefficient of h under the condition of zero phasea(i,0)＝hMF(iTs/Ti) When the phase is munThe exact matched filter coefficient is ha(i,μn)＝hMF((i+μn)Ts/Ti) wherein-L1≤i≤L2，μnTo interpolate phase, L1、L2Is a matched filter length parameter.
As a preferred embodiment of the present invention, the method for constructing the matched filter coefficient lookup table includes: and quantizing the interpolation phase within the interval of [0, 1), wherein the quantization precision needs to be selected by combining the actual performance requirement and the size of a lookup table. For each quantized phase, a set of matched filter coefficients is calculated as described in claim 4.
As a preferred embodiment of the present invention, the matched filtering of the primary sampling point and the secondary sampling point may share a matched filtering coefficient lookup table.
As a preferred embodiment of the present invention, the matched filter coefficient obtaining process includes:
according to the preset quantization phase in the matched filter coefficient lookup table, the interpolation phase mu of the main sampling point and the sub-sampling pointnAnd mun+1/2And respectively quantizing the values into quantized phase values which are closest to the quantized phase values, and reading the matched filter coefficients of the main sampling point and the secondary sampling point from the matched filter coefficient lookup table according to the quantization result.
As a preferred embodiment of the present invention, the matched filtering process is: and loading the input data through a shift register, and multiplying and adding the input data by the matched filter coefficients of the main sampling point and the sub sampling point respectively to obtain the results of the main sampling point and the sub sampling point of the corresponding symbol respectively.
The invention also discloses a timing synchronization device under low sampling rate, which comprises:
a timing error extractor for extracting a timing error using a Gardner algorithm;
the loop filter is used for filtering noise and high-frequency components of the timing error extraction calculation result;
a numerically controlled oscillator for calculating the effective position of the interpolation point, providing an enabling signal for each symbol on the one hand, and calculating the interpolation phase on the other hand;
a matched filter coefficient lookup table for accumulating fractional part mu of phase according to the nth symbol primary and secondary sampling pointnAnd mun+1/2Obtaining the matched filter coefficients of the main sampling point and the sub sampling point of the nth symbol;
and the matched filter loads input data through a shift register, multiplies and adds the matched filter coefficients of the main sampling point and the sub sampling point respectively to obtain the results of the main sampling point and the sub sampling point of the corresponding symbol respectively.
The technical effects of the invention are mainly embodied in the following two aspects:
1) because the interpolation process is fused into the matched filtering, the distortion caused by the interpolation filtering is avoided, and theoretically, the method can work under the lowest sampling rate of Nyquist, namely the sampling rate is equal to the signal bandwidth;
2) the delay caused by interpolation filtering is avoided, and the corresponding multiplier resource is saved.
Drawings
FIG. 1 is a block diagram of a prior art timing synchronization implementation;
FIG. 2 is a schematic diagram of sampling in different cases;
FIG. 3 is an exemplary timing synchronization loop filter structure;
FIG. 4 is a frequency domain amplitude response at different interpolation phases;
FIG. 5 is a block diagram of a timing synchronization implementation of the present invention;
FIG. 6 shows a root-raised cosine filter waveform with a roll-off coefficient of 0.2;
FIG. 7 is a block diagram of a matched filter implementation;
fig. 8 is a diagram comparing MSE performance after timing synchronization according to the present invention and the prior art.
Detailed Description
The invention will be further illustrated and described with reference to specific embodiments. The technical features of the embodiments of the present invention can be combined correspondingly without mutual conflict.
As shown in fig. 5, which is a block diagram of the implementation scheme of timing synchronization of the present invention, the present invention places interpolation inside a matched filter, and realizes the fusion of timing interpolation and matched filtering by dynamically adjusting the filter coefficient of the matched filter. Wherein, the timing error extractor, the loop filter and the NCO still adopt the basic structure of the prior Gardner timing synchronization, and are not described again here.
1. Matched filter coefficient lookup table construction
Conventional matched filtering schemes operate at twice the symbol rate, and at such integer symbol rates, the phase of matched filtering is fixed for each data, and the matched filter coefficients are therefore unchanged. The matched filtering works under the ADC sampling rate, the ADC sampling rate cannot be guaranteed to be an integral multiple of the symbol rate, and the timing phase deviation needs to be adjusted continuously, so that the phase of the matched filtering is changed continuously, and the coefficient of the matched filter needs to be updated continuously.
Suppose hMF(T) is the symbol period T in timeiNormalized standard matched filter waveform (common h)MF(t) is a root-raised cosine filter, and fig. 6 shows the root-raised cosine filter waveform when the roll-off coefficient is 0.2), and the matched filter coefficient is h under the condition of zero phasea(i,0)＝hMF(iTs/Ti) When the phase is munWhen the matched filter coefficient is ha(i,μn)＝hMF((i+μn)Ts/Ti) wherein-L1≤i≤L2. Interpolated phase munQuantization precision of (1), matched filter length parameter L1、L2Determined by the system design requirements (the higher the system performance requirements, μnThe higher the quantization accuracy requirement of, L1、L2The larger the requirement).
The matched filter coefficient lookup table of the technical scheme of the invention is comparable to the interpolation filter coefficient lookup table of the prior technical scheme, and the matched filters of the main sampling point and the secondary sampling point can share one coefficient lookup table.
2. Matched filter
The implementation of the matched filter is shown in fig. 7. And loading the input data through a shift register, and multiplying and adding the input data by the matched filter coefficients of the main sampling point and the sub sampling point respectively to obtain the results of the main sampling point and the sub sampling point of the corresponding symbol respectively. In the present invention, the sub-sample is half a symbol period later than the main sample corresponding to the symbol.
Compared with the prior art, the matched filter coefficients of the invention are dynamically changed, and the matched filter coefficients of the main sampling point and the secondary sampling point are different. In fig. 7, since the multiplication operation is only enabled at the symbol level, the multiplier resources of the primary and secondary sampling points can be multiplexed when the system clock is higher than twice the symbol clock.
The specific implementation process of the invention is as follows:
1) extracting a timing error by using a Gardner algorithm according to an output result of the (n-1) th symbol, wherein a sub-sampling point of the (n-2) th symbol, a main sampling point of the (n-1) th symbol and a sub-sampling point are required;
2) the timing error is filtered by a loop and NCO to obtain the accumulated phase of a main sampling point and a secondary sampling point of timing n symbols, and the accumulated phase is determined according to the integer part m of the accumulated phasenAnd mn+1/2Generating a matched filtering enable signal;
3) accumulating fractional part mu of phase according to nth symbol primary and secondary sampling pointnAnd mun+1/2Combining the matched filter coefficient lookup table to obtain matched filter coefficients of a main sampling point and a secondary sampling point of the nth symbol;
4) and performing matched filtering on the nth symbol to obtain a corresponding main sampling point and a corresponding sub sampling point value.
Integral part m of accumulated phase when main sampling point and sub sampling pointnAnd mn+1/2When the sampling time is not equal to the preset sampling time, different matched filtering enabling is respectively given to enable the main sampling point and the sub sampling point to be output under different clocks, or the number taking mode in the shift register in the figure 7 is changed to enable the main sampling point and the sub sampling point to be output under the same clock.
In order to compare the performance of the solution of the present invention with the prior art solution, an example of Matlab simulation is given here. Wherein the symbol rate is set to 1Gsps, and the shaping filter and the matched filter both adopt root raised cosine filters with roll-off coefficients of 0.2, i.e. signalsThe bandwidth is 1.2GHz, the SNR is set to be 50dB, the sampling deviation phase is quantized to 1024 values, the sampling error is set to be 10ppm, and the length parameter L of the matched filter is1＝L 215, the channel environment is a gaussian white noise channel. For the prior art scheme, the sampling value of the required time point, namely I, is obtained by resampling the surrounding 10 points based on the Lagrange polynomial1＝5，I2＝4。
Fig. 8 shows the performance result of Mean Square Error (MSE), where MSE is ideally inversely proportional to SNR, i.e. MSE and SNR are opposite numbers in dB. As shown in fig. 8, by using the technical solution of the present invention, as long as the sampling rate reaches the nyquist sampling requirement (greater than or equal to 1.2GHz), the MSE approaches the ideal-50 dB. Under the prior art, when the sampling rate is close to the lowest sampling rate of the Nyquist distortion-free, the MSE deterioration is very obvious, the deterioration amount even exceeds 25dB, and the MSE performance of the prior art can reach the level of the technical scheme of the invention only when the sampling rate reaches 2GHz, namely 67.67% higher than the lowest sampling rate of the Nyquist distortion-free.
The above-mentioned embodiments only express several embodiments of the present invention, and the description thereof is more specific and detailed, but not construed as limiting the scope of the present invention. It should be noted that, for a person skilled in the art, several variations and modifications can be made without departing from the inventive concept, which falls within the scope of the present invention. Therefore, the protection scope of the present patent shall be subject to the appended claims.
Claims (9)
1. A timing synchronization method under low sampling rate is characterized by comprising the following steps:
1) extracting a timing error according to the output result of the (n-1) th symbol,
2) the timing error is filtered by a loop and NCO to obtain the accumulated phase of a main sampling point and a secondary sampling point of timing n symbols, and the accumulated phase is determined according to the integer part m of the accumulated phasenAnd mn+1/2Generating a matched filtering enable signal;
3) according to the n thFractional part mu of accumulated phase of symbol primary and secondary sampling pointsnAnd mun+1/2Combining the matched filter coefficient lookup table to obtain matched filter coefficients of a main sampling point and a secondary sampling point of the nth symbol;
4) and performing matched filtering on the nth symbol to obtain a corresponding main sampling point and a corresponding sub sampling point value.
2. The method for timing synchronization at low sampling rate according to claim 1, wherein: integral part m of accumulated phase when main sampling point and sub sampling pointnAnd mn+1/2When the sampling time is not equal, different matched filtering enables are respectively given to enable the main sampling point and the secondary sampling point to be output under different clocks.
3. The method for timing synchronization at low sampling rate according to claim 1, wherein: integral part m of accumulated phase when main sampling point and sub sampling pointnAnd mn+1/2When the sampling times are not equal, the access mode of a shift register in the matched filter is changed, so that the main sampling point and the secondary sampling point are output under the same clock.
4. The method for timing synchronization at low sampling rate according to claim 1, wherein: suppose hMF(T) is the symbol period T in timeiThe normalized standard matched filter has a wave form with an accurate matched filter coefficient of h under the condition of zero phasea(i，0)＝hMF(iTs/Ti) When the phase is munThe exact matched filter coefficient is ha(i，μn)＝hMF((i+μn)Ts/Ti) wherein-L1≤i≤L2，μnTo interpolate phase, L1、L2Is a matched filter length parameter.
5. The method for timing synchronization at low sampling rate of claim 4, wherein: the construction method of the matched filter coefficient lookup table comprises the following steps: the interpolated phases are quantized in the interval [0, 1), and for each quantized phase a set of matched filter coefficients is calculated according to the method of claim 4.
6. The method for timing synchronization at low sampling rate according to claim 1, wherein: the matched filtering of the primary and secondary samples may share a matched filter coefficient look-up table.
7. The method for timing synchronization at low sampling rate according to claim 1, wherein: the matched filter coefficient obtaining process comprises the following steps:
according to the preset quantization phase in the matched filter coefficient lookup table, the interpolation phase mu of the main sampling point and the sub-sampling pointnAnd mun+1/2And respectively quantizing the values into quantized phase values which are closest to the quantized phase values, and reading the matched filter coefficients of the main sampling point and the secondary sampling point from the matched filter coefficient lookup table according to the quantization result.
8. The method for timing synchronization at low sampling rate according to claim 1, wherein: the matched filtering process comprises the following steps: and loading the input data through a shift register, and multiplying and adding the input data by the matched filter coefficients of the main sampling point and the sub sampling point respectively to obtain the results of the main sampling point and the sub sampling point of the corresponding symbol respectively.
9. A timing synchronization apparatus at a low sampling rate, comprising:
a timing error extractor for extracting a timing error using a Gardner algorithm;
the loop filter is used for filtering noise and high-frequency components of the timing error extraction calculation result;
a numerically controlled oscillator for calculating the effective position of the interpolation point, providing an enabling signal for each symbol on the one hand, and calculating the interpolation phase on the other hand;
a matched filter coefficient lookup table for accumulating fractional part mu of phase according to the nth symbol primary and secondary sampling pointnAnd mun+1/2To obtain the nth symbolMatching filter coefficients of the main sampling point and the secondary sampling point of the sign;
and the matched filter loads input data through a shift register, multiplies and adds the matched filter coefficients of the main sampling point and the sub sampling point respectively to obtain the results of the main sampling point and the sub sampling point of the corresponding symbol respectively.
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| CN114845376A (en) * | 2022-04-24 | 2022-08-02 | 之江实验室 | High-speed parallel timing synchronization method based on FPGA |
| CN115001645A (en) * | 2022-06-13 | 2022-09-02 | 北京邮电大学 | Clock recovery method and device, electronic equipment and computer storage medium |
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