CN110912405B - Four-switch buck-boost converter based on voltage mode control - Google Patents

Four-switch buck-boost converter based on voltage mode control Download PDF

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CN110912405B
CN110912405B CN201910991786.0A CN201910991786A CN110912405B CN 110912405 B CN110912405 B CN 110912405B CN 201910991786 A CN201910991786 A CN 201910991786A CN 110912405 B CN110912405 B CN 110912405B
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buck
boost
voltage
module
mode
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CN110912405A (en
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赵亦林
胡峻
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Hangzhou Yugu Technology Co ltd
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Hangzhou Dongyi Technology Co ltd
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H02M3/1582—Buck-boost converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0016—Control circuits providing compensation of output voltage deviations using feedforward of disturbance parameters
    • H02M1/0022—Control circuits providing compensation of output voltage deviations using feedforward of disturbance parameters the disturbance parameters being input voltage fluctuations
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Abstract

The invention discloses a four-switch buck-boost converter based on voltage mode control, which comprises a buck module, a boost module and an inductor L connected with the buck module and the boost module, wherein the buck module is connected with a current input end; the modulation signal generation module outputs two modulation signals V c_Buck 、V c_Boost (ii) a The PWM modulator combines one or both of the two modulated signals with a carrier signal V ramp After the comparison, the duty control signal d1 or the duty control signal d2 is output or the duty control signals d1 and d2 are output simultaneously. The converter realizes smooth switching of the working mode of the converter by a dual modulation-single carrier three-mode modulation mode; the compensator is arranged to supplement the converter, so that smooth mode switching and stable work of the converter are realized, the reliability is higher, and the service life is longer.

Description

Four-switch buck-boost converter based on voltage mode control
Technical Field
The invention belongs to the technical field of power supply voltage conversion, and particularly relates to a four-switch buck-boost converter based on voltage mode control.
Background
In modern society, as technology is continuously developed, electronic devices are applied to various fields such as life, office, medical treatment, production and the like, and especially, portable electronic devices such as mobile phones, notebooks, mobile hard disks, hair dryers, various small household appliances and the like are widely used by people because the use of the portable electronic devices is not limited by regions and the portable electronic devices are easy to carry about. The wider and wider use of portable electronic devices has led to higher and higher requirements for power management systems of electronic devices. Due to the wide range of the output voltage of the external charging power supply and the requirements of power energy conversion and use among various devices, most of the portable electronic devices can be charged after a buck converter or a boost converter is used as a conversion circuit of a battery charging system. The buck-boost converter with the single-inductor four-switch has wider input and output voltage range, and compared with other buck-boost converters such as the conventional buck-boost converter, cuk converter, zeta converter and SEPIC converter, the buck-boost converter has the advantages of same input and output polarities, simple circuit, few devices and high efficiency, and is widely applied to portable electronic equipment, communication power supplies, storage battery power supply systems, power factor correction power supplies and the like.
Fig. 1 is a schematic diagram of a single-inductor four-switch synchronous buck boost converter, wherein Q1 and Q3 are main power transistors, and Q2 and Q4 are synchronous rectifier transistors. Compared with the traditional control mode that Q1 and Q3 are simultaneously switched on and off, the control strategy for respectively controlling the Q1 and the Q3 to be switched on can effectively reduce the conduction loss and the switching loss of the converter and improve the efficiency. When the input voltage is higher than the output voltage, the converter works in buck mode; when the input voltage is lower than the output voltage, the converter operates in the boost mode.
However, in the process of directly switching from the buck mode to the boost mode or from the boost mode to the buck mode, the direct switching of the two modes makes the voltage signal change greatly, which is easy to cause disturbance, and is not beneficial to the stable and smooth change of the voltage, so that the switching process is more abrupt, the switching loss is large, and the mode conversion effect and the service life of the whole converter are affected.
Therefore, there is a need for an improved four-switch buck-boost converter, which can effectively alleviate the switching between modes and achieve a smooth switching.
Disclosure of Invention
The invention aims to solve the problems and provides a four-switch buck-boost converter based on voltage mode control.
In order to achieve the purpose of the invention, the invention adopts the following technical scheme:
a four-switch buck-boost converter based on voltage mode control comprises a buck module, a boost module and an inductor L connected with the buck module and the boost module, wherein the buck module is connected with a current input end, and the boost module is provided with a current output end;
the modulation signal generation module outputs two modulation signals V c_Buck 、V c_Boost ；
The PWM modulator will have twoOne or both of the modulated signals being simultaneously associated with a carrier signal V ramp After the comparison, the duty control signal d1 or the duty control signal d2 is output or the duty control signals d1 and d2 are output simultaneously.
The converter of this scheme of adoption through the setting of comparator and modulation signal generation module, PWM modulator, has increased the mode of converter, and this scheme of adoption promptly has three kinds of mode: buck mode, buck-boost mode, when the input voltage changes, the control signal d 1 And d 2 The three-phase inverter can be rapidly changed to inhibit the influence of input voltage disturbance, and the stable switching of three working modes is realized, so that the inverter is more reliable.
Further, the modulation signal generation module comprises an error amplifier, a resistor Z1 and a bias voltage V bia The current output end, the error amplifier and the resistor Z 1 The output end of the error amplifier is divided into two paths, one path is directly connected with the PWM modulator, and the other path is connected with the bias voltage V bia And then connected with the PWM modulator. The feedback voltage V is output after the output voltage of the current output end is connected with the high voltage FB For the error amplifier, the error amplifier will V FB After comparing with the reference voltage, the error amplifying signal is output, and the error amplifying signal is output into 2 paths of modulation signals through the bias voltage.
Furthermore, the modulation signal generation module further comprises a high-voltage module, and the current output end, the high-voltage module and the error amplifier are connected in sequence.
Further, the buck module includes a switch tube Q1 and a switch tube Q2, a drain of the switch tube Q1 is connected to the power input terminal VIN, a gate of the switch tube Q1 is connected to the duty ratio control signal d1 and a gate of the switch tube Q2, and a drain and a source of the switch tube Q2 are connected to the inductor and the boost module, respectively.
Furthermore, the boost module comprises a switching tube Q3 and a switching tube Q4, and the switching tube Q4 and the switching tube Q2 are synchronous rectifier tubes;
the source electrode of the switch tube Q4 is connected with the inductor, the grid electrode of the switch tube Q4 is respectively connected with the duty ratio control signal d2 and the grid electrode of the rectifier tube Q3, and the source electrode and the drain electrode of the switch tube Q3 are respectively connected with the source electrode of the switch tube Q2 in the inductor and the buck module.
Furthermore, the converter is provided with a lowest voltage VL and a highest voltage VH, V of the control signal bias Is the offset between the two control signals, VM is the amplitude of the carrier, where V c_Buck ＝V c_Boos t+V bias ,V bias <VM；
When VL<V c_Buck <VH,V c_Boost <In VL, the converter operates in buck mode, V c_Buck Comparing with the carrier signal to generate a duty ratio d1 signal required by the buck working mode, wherein the switching tube Q3 is open-circuited, and the switching tube Q4 is closed;
when VL is present<V c_Buck <VH,VL<V c_Boost <At VH, the converter operates in buck-boost mode, V c_Buck And V c_Boost Comparing the signal with a carrier wave to generate needed duty ratio d1 and d2 signals;
when VH<V c_Buck ,VL<V c_Boost <In VH, the converter works in boost mode, the switching tube Q1 is closed, the switching tube Q2 is open, and V is c_Boost The comparison with the carrier generates the required duty cycle d2 signal.
And the input end of the comparator is connected with the output end of the modulation signal generation module, and the output end of the comparator is connected with the PWM modulator.
Furthermore, the voltage regulator also comprises a III type compensator, wherein one input end of the III type compensator is respectively connected with a reference voltage Vref input end of the error amplifier, and the other input end of the III type compensator is connected with a feedback voltage V of the error amplifier FB The output end of the III-type compensator is connected with the modulation signal end.
Furthermore, the III type compensator comprises an operational amplifier, a resistor R1, a resistor R2, a resistor R3, a resistor Rz and a capacitor C ff A capacitor Cz and a capacitor Cp, wherein the non-inverting input end of the operational amplifier is connected with the reference voltage V of the error amplifier ref Input terminal, inverse of said operational amplifierThe input end of the voltage regulator is connected with one end of the resistor R1, one end of the resistor R2 and the feedback voltage V of the error amplifier FB The input ends are respectively connected;
the capacitor C ff The two ends of the branch circuit after the resistor R3 is connected in series are respectively connected with the two ends of the resistor R1;
the capacitor Cz and the resistor Rz are connected in series and then are connected in parallel with the capacitor Cp to form a branch, and two ends of the branch are respectively connected with the output end and the inverted input end of the operational amplifier;
the other end of the R2 is connected with a voltage output end.
Compared with the prior art, the invention has the beneficial effects that:
1, by adopting the converter, the smooth switching of the working mode of the converter is realized by a dual-modulation-single-carrier three-mode modulation mode;
2. the method comprises the steps that feedforward voltage mode control is adopted, and in order to restrain the influence of input voltage disturbance on output voltage, input voltage is introduced into a control loop in a carrier signal construction mode through the input voltage;
3. when the converter is in light load, a skip cycle control mode is adopted to improve the light load efficiency of the converter;
4. the compensator is arranged to supplement the converter, so that the mode smooth switching and stable work of the converter are realized;
5. by switching the Buck mode, the Buck-Boost mode and the Boost mode, the mode switching is smoother and more reliable, the conduction loss is smaller, the reliability of the converter is higher, and the service life is longer.
Drawings
FIG. 1 is a prior art single inductor four switch synchronous buck boost converter;
FIG. 2 is a schematic diagram of the operation of dual modulation-single carrier tri-mode switching;
FIG. 3 (a) is a simulation waveform diagram of Buck operation mode among three operation modes;
FIG. 3 (b) is a simulation waveform diagram of Buck-Boost operation mode among three operation modes;
FIG. 3 (c) is a simulation waveform diagram of the Boost operation mode of the three operation modes;
FIG. 4 is a schematic diagram of a single-inductor four-switch Buck-Boost converter of one configuration of the present invention;
FIG. 5 is a block diagram of the four switch buck-boost converter control of the present invention;
FIG. 6 is a schematic diagram of a type III compensator;
FIG. 7 (a) is a Burde plot of a loop modeled and simulated in buck mode;
FIG. 7 (b) is a loop bode plot for modeling and simulation in Boost mode;
FIG. 8 (a) is a closed loop transfer function bode plot of buck mode input voltage to output voltage;
FIG. 8 (b) is a closed loop transfer function boost plot of input voltage to output voltage for boost mode;
FIG. 9 (a) is a diagram of the system dynamic response of a low-to-high transition waveform between input voltages 8V-24V;
FIG. 9 (b) is a diagram of the system dynamic response of the high-to-low transition waveform between input voltages 8V-24V;
FIG. 10 (a) is a waveform diagram of Buck mode experiment;
FIG. 10 (b) is a Buck-Boost mode experimental waveform diagram;
fig. 10 (c) is a Boost mode experimental waveform diagram;
FIG. 11 (a) is a waveform diagram of a load jump experiment;
fig. 11 (b) is an enlarged waveform diagram.
Detailed Description
The technical scheme of the invention is further described and illustrated by specific embodiments below, so that the technical scheme is clearer and more obvious.
As shown in fig. 4, the present embodiment discloses a four-switch buck-boost converter capable of implementing mode smooth switching, which mainly includes:
the direct current power supply comprises a buck module, a boost module and an inductor L for connecting the buck module with the boost module, wherein the buck module is provided with a current input end, the boost module is connected with the current input end, and the direct current power supply further comprises a modulation signal generation module, a PWM modulator, a current output end, a modulation signal generation module and the PWM modulator which are sequentially connected;
the modulation signal generationThe module outputs two modulation signals V c_Buck 、V c_Boost ；
The PWM modulator combines one or both of two modulation signals with a carrier signal V ramp After the comparison, the duty control signal d1 or the duty control signal d2 is output or the duty control signals d1 and d2 are output simultaneously.
As a preferred embodiment in this embodiment, in the present converter:
the buck module is mainly provided with a switching tube Q1 and a switching tube Q2, wherein the switching tube Q1 is a main power tube, and the switching tube Q2 is a rectifying tube;
the boost module is mainly provided with a switching tube Q3 and a switching tube Q4, wherein the switching tube Q3 is a main power tube, and the switching tube Q4 and a switching tube Q2 are synchronous rectifier tubes.
Specifically analyzing a preferred structure of the buck module, in this embodiment, the buck module includes a switching tube Q1 and a switching tube Q2, a drain of the switching tube Q1 is connected to the power input terminal VIN, a gate of the switching tube Q1 is connected to the duty ratio control signal d1 and a gate of the switching tube Q2, and a drain and a source of the switching tube Q2 are connected to the inductor and the boost module, respectively.
Specifically analyzing a preferred structure of the boost module, in this embodiment, the boost module includes a switching tube Q3 and a switching tube Q4, and the switching tube Q4 and the switching tube Q2 are synchronous rectifier tubes;
the source electrode of the switching tube Q4 is connected with the inductor, the grid electrode of the switching tube Q4 is respectively connected with the duty ratio control signal d2 and the grid electrode of the rectifying tube Q3, and the source electrode and the drain electrode of the switching tube Q3 are respectively connected with the inductor and the source electrode of the switching tube Q2 in the buck module.
Fig. 2 is a schematic diagram illustrating the operation of dual modulation-single carrier three-mode switching. Vc _ Buck is the control signal for Buck mode, vc _ Boost is the control signal for Boost mode, vbias is the offset between the two control signals, and VM is the amplitude of the carrier. Wherein Vc _ Buck = Vc _ Boost + Vbias, vbias < VM. When VL < Vc _ Buck < VH and Vc _ Boost < VL, the converter works in a Buck mode, vc _ Buck is compared with a carrier signal to generate a duty ratio d1 signal required by the Buck working mode, a switching tube Q3 is open, and a switching tube Q4 is closed; when VL < Vc _ Buck < VH and VL < Vc _ Boost < VH, the converter works in Buck-Boost mode, and Vc _ Buck and Vc _ Boost signals are compared with a carrier to generate required duty ratio d1 and d2 signals; when in use
When VH < Vc _ Buck and VL < Vc _ Boost < VH, the converter works in a Boost mode, at the moment, the switching tube Q1 is closed, the switching tube Q2 is open, and the Vc _ Boost is compared with the carrier to generate a required duty ratio d2 signal.
The simulation waveforms of the three operation modes are shown in the pairs of fig. 3 (a), 3 (b) and 3 (c), wherein G1 is the driving signal of the Q1 transistor, G3 is the driving signal of the Q3 transistor, and IL is the inductor current waveform.
When Vin =20v, vout =12v, and iout =4a, the simulated waveform of the converter is as shown in fig. 3 (a), and the converter operates in buck mode.
When Vin =11v, vout =12v, iout =4a, the converter simulation waveform is as shown in fig. 3 (b), and the converter operates in buck-boost mode.
When Vin =8v, vout =12v, iout =4a, the simulated waveform of the converter is as shown in fig. 3 (c), and the converter operates in the boost mode.
From the simulation waveforms, compared with the traditional control mode that Q1 and Q3 are simultaneously turned on and off, the three-mode switching working mode has the advantages that when the converter works in a Buck or Boost mode, the switching loss is reduced, and when the converter works in a Buck-Boost mode, the effective value of the current flowing through the power tube is reduced, so the conduction loss is smaller.
As a preferred implementation manner of this embodiment, the modulation signal generating module includes an error amplifier, a resistor Z 1 Bias voltage V bia The current output end, the error amplifier and the resistor Z 1 Sequentially connected, the output end of the error amplifier is divided into two paths, one path is directly connected with the PWM modulator, and the other path is connected with the bias voltage V bia And then connected with the PWM modulator.
As a preferred implementation manner of this embodiment, the modulation signal generation module further includes a high voltage module, and the current output terminal, the high voltage module, and the error amplifier are connected in sequence.
As a preferred implementation manner of this embodiment, the PWM controller further includes a comparator, an input terminal of the comparator is connected to the output terminal of the modulation signal generation module, and an output terminal of the comparator is connected to the PWM modulator.
As can be seen from FIG. 4, the output of the error amplifier generates two modulated signals V by means of a bias voltage c_buck And V c_boost The two modulated signals are compared with the same carrier signal to generate the required drive signal (as shown in fig. 2). In order to suppress the influence of the disturbance of the input voltage on the output voltage, a feedforward control is introduced, and the input voltage is passed through an integrating circuit to form a carrier signal. When the input voltage varies, the control signals d1 and d2 can be rapidly varied to suppress the influence of the input voltage disturbance.
In the present embodiment, the expression of the carrier signal is as shown in equation (1):
Figure BDA0002238526890000081
when the input voltage is unchanged, the carrier signal Vramp is a group of periodic sawtooth wave signals, and the modulation signals Vc _ Buck and Vc _ Boost and the carrier signal Vramp generate duty ratio control signals d1 and d2 through a PWM modulator. The expression of the PWM modulator is shown in equation (2):
Figure BDA0002238526890000082
wherein vc (t) is a modulation signal,
Figure BDA0002238526890000083
is the carrier slope. As can be seen from equation (2), the output d signal of the PWM modulator is a function d = f (v) of vin and vc c ，v in ) So the gain expression of the PWM modulator in the complex frequency domain is:
Figure BDA0002238526890000084
at the time of the steady-state,
Figure BDA0002238526890000085
where VL is the low clamp voltage value of Vc.
Substituting formula (4) into formula (3) yields:
Figure BDA0002238526890000091
FIG. 5 is a block diagram of the voltage closed-loop control of the converter, where Gvd(s), gvg(s), and Zout(s) are the duty cycle d ^ and the transfer function and output impedance of the input voltage g v ^ to the output voltage o v ^, respectively, H(s) is the sampling function of the output voltage, gv(s) is the compensation function of the voltage loop, fm is the PWM modulator gain, and k is the gain of the feedforward input voltage.
The values of the PWM modulator gain Fm and the feedforward voltage gain k can be derived from equation (5).
Figure BDA0002238526890000092
Figure BDA0002238526890000093
When the converter is operating in buck mode, the duty cycle to output transfer function is:
Figure BDA0002238526890000094
the transfer function of the input voltage to the output voltage is:
Figure BDA0002238526890000095
wherein the content of the first and second substances,
Figure BDA0002238526890000096
when the converter is operating in boost mode, the duty cycle to output transfer function is:
Figure BDA0002238526890000101
the transfer function of the input voltage to the output voltage is:
Figure BDA0002238526890000102
wherein the content of the first and second substances,
Figure BDA0002238526890000103
since the buck-boost operating mode is only a transition state between the buck mode and the boost mode, and the boost mode and the buck-boost mode have similarity on a control loop, we put the buck-boost operating mode into the boost mode for analysis.
No matter the converter works in a Buck mode or a Boost mode, the system is a second-order system, and in order to enable the converter to work stably, a III-type compensator is adopted to compensate the converter. A schematic diagram of a type iii compensator is shown in fig. 6. One input end of the III type compensator is respectively connected with a reference voltage Vref input end of the error amplifier, the other input end of the III type compensator is connected with a feedback voltage VFB input end and a voltage output end of the error amplifier, and the output end of the III type compensator is connected with the modulation signal end.
As a preferred implementation, the iii-type compensator of this embodiment includes an operational amplifier, a resistor R1, a resistor R2, a resistor R3, a resistor Rz, and a capacitor C ff A capacitor Cz and a capacitor Cp, wherein the non-inverting input end of the operational amplifier is connected with the reference voltage V of the error amplifier ref The input end, the inverting input end of the operational amplifier is connected with one end of the resistor R1, one end of the resistor R2 and the error amplifierFeedback voltage V of amplifier FB The input ends are respectively connected; the capacitor C ff The two ends of the branch circuit after the resistor R3 is connected in series are respectively connected with the two ends of the resistor R1; the capacitor Cz is connected with the resistor Rz in series and then connected with the capacitor Cp in parallel to form a branch, and two ends of the branch are respectively connected with the output end and the inverting input end of the operational amplifier; and the other end of the R2 is connected with a voltage output end.
The complex frequency domain expression of the type III compensator is as follows:
Figure BDA0002238526890000111
the pole at the origin of the compensator improves the direct current gain of the system and reduces the steady-state error of the output voltage of the system. Two zero points are designed before the LC resonance frequency, the zero points weaken the influence of the LC pole pairs of the system, and the phase margin of the system is improved. After the two poles are designed at 1/2 of the switching frequency, the two high-frequency poles enable the high-frequency gain to be rapidly reduced, and high-frequency noise is suppressed.
In addition to requiring good compensation, we also need to reduce the Q of buck and boost modes to stabilize them. As can be seen from the above equations (8) and (10), to reduce the Q value, we need to reduce the output capacitance or increase the inductance, but for the boost mode, increasing the inductance also causes the zero point of the right half-plane to move to a low frequency, so that the phase margin of the boost mode is reduced, and the boost mode is unstable, so the value of the inductance needs to be moderate.
As shown in fig. 7 (a), a loop bode diagram of modeling and simulation in buck mode is shown, and it can be seen from the diagram that the built buck mode small signal model is quite consistent with the simulation, which shows that the built small signal model is accurate. At this time, the crossover frequency f kHz c =22, phase margin o =53 mp.
As shown in fig. 7 (b), which is a boost-mode modeled and simulated loop boost diagram, it can be seen that the right half-plane zero suppresses the phase margin increasing effect caused by the feedforward capacitance compensation, resulting in the reduction of the phase margin of the boost mode. At this time, the crossover frequency f kHz c =21, the phase margin o =28 mp.
As can be seen from fig. 7 (a) and 7 (b), the crossing frequency of the system is low and the dynamic response of the input voltage of the system is poor both in buck mode and boost mode, so we need to introduce the input voltage into the feedback loop to improve the dynamic response of the input voltage. As can be seen from the control block diagram of the buck-boost converter of fig. 5, the open-loop gain of the voltage loop is:
T v (s)＝H(s)G v (s)F m G vd (s) (13)
when input voltage feed-forward is not employed, the closed-loop transfer function of the system input voltage to output voltage is:
Figure BDA0002238526890000121
when input voltage feed-forward is employed, the closed-loop transfer function of the system input voltage to output voltage is:
Figure BDA0002238526890000122
from formula (15) and formula (14):
Figure BDA0002238526890000123
when the buck-boost converter operates in buck mode:
Figure BDA0002238526890000124
when the buck-boost converter operates in boost mode:
Figure BDA0002238526890000125
in design, the low-clamp value of the control signal Vc is usually small, and as can be seen from equations (17) and (18), after the voltage feedforward is added, the closed-loop gain from the input to the output of the buck-boost converter is reduced, which shows that the input voltage feedforward control suppresses the influence of the input voltage disturbance on the output voltage.
As shown in fig. 8 (a) and fig. 8 (b), which are closed loop transfer function bode plots of input voltage to output voltage in buck mode and boost mode, it can be seen from the plots that, after the voltage feedforward is added, the gain is obviously reduced, and the system can effectively suppress the influence of input voltage disturbance on the output voltage.
As shown in fig. 9, is the system dynamic response for the transition between input voltages 8V-24V. It can be seen from the figure that the system smoothly switches between the three modes of operation when the input voltage jumps. When the input voltage jumps from 8V to 24V, the converter passes through buck-boost mode from buck mode and finally works in boost mode. When the input voltage jumps from 24V to 8V, the converter passes through the buck-boost mode from the boost mode and finally works in the buck mode. When the input voltage jumps, the change of the output voltage is within +/-8 percent, and the system requirement is met.
Finally, for the converter of the present embodiment, validity verification is performed by an experimental prototype. Specifically, an experimental prototype is built under laboratory conditions to verify the working performance of the prototype under the conditions of the proposed control strategy and compensation parameters. The main parameters of the prototype: vin =3V-36v, vout =12v, l =4.4uh, cout =44uf, rz =50k Ω, cz =680pf, cff =470pf, cp =5p.
Fig. 10 shows the steady state experimental waveforms in the different operating modes. Fig. 10 (b) shows an experimental waveform in buck-boost mode, in which the inductor current waveform is a trapezoidal wave, and the effective value of the inductor current is reduced and the system conduction loss is reduced compared with a triangular wave of the same load. Fig. 10 (c) shows experimental waveforms in the boost mode, and since the system operates in the skip cycle operation mode to reduce system loss during light load, the system may slightly oscillate due to the mutual influence of the skip cycle loop and the continuous conduction control loop during critical conduction.
Since the buck-boost mode is a transition state between the buck mode and the boost mode and is also a region of system instability, the test of the load jump response of the buck-boost mode is shown in fig. 11. It can be seen from the figure that when the load jumps between 0 and 5A, the output voltage is stabilized within ± 5%, which indicates that the bandwidth of the system is reasonably designed, and meanwhile, the system does not have unstable oscillation, which indicates that the system has sufficient phase margin, and at this time, the system has better suppression on the input disturbance.
In summary, for a four-switch buck-boost converter working in three modes, different working modes are modeled, and the accuracy of the model is verified through loop simulation. Because the voltage mode control has weak ability of suppressing the input voltage disturbance, in order to solve the problem, the input voltage is introduced into a feedback loop by constructing a carrier signal mode by using the input voltage to suppress the disturbance of the input voltage, and simulation shows that the phase margin of a system is improved and the stability is enhanced by reasonable compensation parameter design and input voltage feedforward. Finally, an experimental prototype with input voltage of 3V-36V and input voltage of 12V is designed, and tests show that the system can well realize smooth switching of three modes, and meanwhile, the compensation parameters of the system are reasonable, the bandwidth of the system is high, and sufficient phase margin is provided.
The above is the preferred embodiment of the present invention, and does not limit the scope of the present invention, and the modifications and improvements made by those skilled in the art according to the design idea of the present invention should be considered as within the scope of the present invention.
Although the invention has been described herein with reference to the examples and drawings, it should be understood that numerous other modifications and embodiments can be devised by those skilled in the art that will fall within the scope and spirit of the principles of this disclosure. More specifically, various variations and modifications are possible in the component parts and/or arrangements of the subject combination arrangement within the scope of the disclosure, the drawings and the appended claims. In addition to variations and modifications in the component parts and/or arrangements, other uses will also be apparent to those skilled in the art.

Claims (8)

1. A four-switch buck-boost converter based on voltage mode control comprises a buck module, a boost module and an inductor L connected with the buck module and the boost module, wherein the buck module is connected with a current input end, and the boost module is provided with a current output end;
the modulation signal generation module outputs two modulation signals V c_Buck 、V c_Boost (ii) a The PWM modulator combines one or two of the two modulation signals with a carrier signal V ramp After comparison, outputting a duty ratio control signal d1 or a duty ratio control signal d2 or simultaneously outputting the duty ratio control signals d1 and d2; the converter has minimum voltage VL and maximum voltage VH, V of control signal bias Is the offset between two control signals, VM is the amplitude of the carrier, where V c_Buck ＝V c_Boos t+V bias ,V bias <VM; when VL is present<V c_Buck <VH,V c_Boost <In VL, the converter operates in buck mode, V c_Buck Comparing with the carrier signal to generate a duty ratio d1 signal required by the buck working mode, wherein the switching tube Q3 is open-circuited, and the switching tube Q4 is closed;
when VL<V c_Buck <VH,VL<V c_Boost <At VH, the converter operates in buck-boost mode, V c_Buck And V c_Boost Comparing the signal with a carrier to generate signals with required duty ratios d1 and d2;
when VH<V c_Buck ,VL<V c_Boost <In VH, the converter works in boost mode, the switching tube Q1 is closed, the switching tube Q2 is open, and V is c_Boost The comparison with the carrier generates the required duty cycle d2 signal.
2. The four-switch buck-boost converter based on voltage mode control according to claim 1, wherein the modulation signal generation module comprises an error amplifier, a resistor Z1 and a bias voltage V bia Error amplification at the current output terminalDevice and resistor Z 1 Sequentially connected, the output end of the error amplifier is divided into two paths, one path is directly connected with the PWM modulator, and the other path is connected with the bias voltage V bia And then connected with the PWM modulator.
3. The four-switch buck-boost converter based on voltage mode control according to claim 2, wherein the modulation signal generating module further comprises a high voltage module, and the current output terminal, the high voltage module and the error amplifier are connected in sequence.
4. The four-switch buck-boost converter based on voltage mode control according to any one of claims 1 to 3, wherein the buck module comprises a switch tube Q1 and a switch tube Q2, and the drain of the switch tube Q1
And the grid electrode of the switching tube Q1 is connected with the duty ratio control signal d1 and the grid electrode of the switching tube Q2 respectively, and the drain electrode and the source electrode of the switching tube Q2 are connected with the inductor and the boost module respectively.
5. A four-switch buck-boost converter based on voltage mode control according to claim 4, wherein said boost module comprises a switch tube Q3 and a switch tube Q4, and said switch tube Q4 and said switch tube Q2 are synchronous rectifier tubes;
the source electrode of the switching tube Q4 is connected with the inductor, the grid electrode of the switching tube Q4 is respectively connected with the duty ratio control signal d2 and the grid electrode of the rectifying tube Q3, and the source electrode and the drain electrode of the switching tube Q3 are respectively connected with the inductor and the source electrode of the switching tube Q2 in the buck module.
6. The four-switch buck-boost converter based on voltage mode control according to claim 1, further comprising a comparator, wherein an input terminal of the comparator is connected to the output terminal of the modulation signal generation module, and an output terminal of the comparator is connected to the PWM modulator.
7. A method according to any one of claims 1 to 3, based onThe four-switch buck-boost converter controlled in the voltage mode is characterized by further comprising a III type compensator, wherein one input end of the III type compensator is connected with a reference voltage Vref input end of the error amplifier, and the other input end of the III type compensator is connected with a feedback voltage V of the error amplifier FB The output end of the III-type compensator is connected with the modulation signal end.
8. The four-switch buck-boost converter based on voltage mode control according to claim 7, wherein the type III compensator comprises an operational amplifier, a resistor R1, a resistor R2, a resistor R3, a resistor Rz and a capacitor C ff A capacitor Cz and a capacitor Cp, wherein the non-inverting input end of the operational amplifier is connected with the reference voltage V of the error amplifier ref An input end, an inverting input end of the operational amplifier, one end of the resistor R1, one end of the resistor R2 and a feedback voltage V of the error amplifier FB The input ends are respectively connected;
the capacitor C ff The two ends of the branch circuit after the resistor R3 is connected in series are respectively connected with the two ends of the resistor R1;
the capacitor Cz and the resistor Rz are connected in series and then are connected in parallel with the capacitor Cp to form a branch, and two ends of the branch are respectively connected with the output end and the inverted input end of the operational amplifier;
and the other end of the R2 is connected with a voltage output end.
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