CN109857186B - Source follower with negative feedback and filter structure - Google Patents
Source follower with negative feedback and filter structure Download PDFInfo
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- CN109857186B CN109857186B CN201811633468.9A CN201811633468A CN109857186B CN 109857186 B CN109857186 B CN 109857186B CN 201811633468 A CN201811633468 A CN 201811633468A CN 109857186 B CN109857186 B CN 109857186B
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Abstract
The invention discloses a source follower with negative feedback and a filter structure, which can realize complex filtering functions by using few circuits, are suitable for high-speed application occasions and low-power consumption occasions, and can work in a differential mode and a single-ended mode.
Description
Technical Field
The invention relates to the technical field of filters, in particular to a source follower with negative feedback and a series of filter structures.
Background
Filters are used in a wide variety of electrical systems to alter the frequency characteristics of analog electrical signals. The filter infrastructure is first order filtering, second order filtering, and third order filtering, and higher order filters can be constructed using the above infrastructure. There are a number of structures and methods to construct a filter. One type of filter uses an operational amplifier and closed loop feedback properties like a Tow-Thomas filter, sallen-Key filter. Such filters, because of the use of closed loop operational amplifiers, are limited in application to low speed and high power consumption applications. Another widely used class of filters uses an open loop Gm-C architecture, mainly for high speed and high performance applications.
Referring to fig. 1, a conventional source follower circuit schematic diagram includes a first transistor M1 and a current mirror including a third transistor M3 and a fourth transistor M4 that are cascode. The gate of the first transistor M1 serves as the input node vg of the source follower, the drain thereof is connected to the power supply VCC, and the source thereof serves as the output node vs of the source follower. The drain electrode of the fourth transistor M4 is connected to the output node vs as the output end of the current mirror, and the source electrode thereof is grounded, and the fourth transistor M4 is the current load of the first transistor M1. The drain electrode of the third transistor M3 is shorted to the gate electrode thereof and then connected to a current source It as the input end of the current mirror, while the source electrode thereof is grounded. The current source It is connected to the power supply VCC. The source follower has very low output impedance, is particularly suitable for driving heavy loads (low-impedance loads) such as motors and speakers, has high common power and has good thermal breakdown resistance.
Recently, there is a new class of simple source follower based filters. One of them is a source follower based filter proposed by Bailey et al (patent application number: US7196573B 1), as shown in fig. 2. The other is a source follower filter based on cross-coupled positive feedback proposed by Conta et al (patent application number: US7659774B 2), as shown in FIG. 3. Such filters have the advantage of being suitable for high speed and low power applications because such structures can use very few circuits to implement complex filtering functions. However, a common disadvantage of both types of filters is that they can only operate in differential mode and not in single-ended mode.
Disclosure of Invention
The invention aims to solve the technical problems in the prior art and provide a source follower with negative feedback and a filter structure, which can realize complex filtering functions by using few circuits, are suitable for high-speed application occasions and low-power consumption occasions, and can work in a differential mode and a single-ended mode.
In order to achieve the above object, the present invention provides a source follower with negative feedback, the source follower including a first transistor, a second transistor, and a current mirror; the grid electrode of the first transistor is used as an input node of the source electrode follower, the drain electrode of the first transistor is connected with a first current source, and the source electrode of the first transistor is used as an output node of the source electrode follower; the second transistor is a folded common gate transistor of the first transistor, the grid electrode of the second transistor is used for receiving a bias voltage, the source electrode of the second transistor is connected with the drain electrode of the first transistor, and the drain electrode of the second transistor is connected with the input end of the current mirror; and the output end of the current mirror is connected to the output node.
In order to achieve the above object, the present invention further provides a filter structure, which includes a source follower and a first capacitor module; the source follower comprises a first transistor and a current mirror; the grid electrode of the first transistor is used as an input node of the source electrode follower, the drain electrode of the first transistor is connected with a first current source, and the source electrode of the first transistor is used as an output node of the source electrode follower; the current mirror comprises a third transistor and a fourth transistor which are in common gate and common source, wherein a common end, which is connected with a grid electrode of the third transistor, is used as a grid electrode node of the current mirror, a drain electrode of the third transistor is short-circuited with the grid electrode node and then is used as an input end of the current mirror to be coupled to the first current source, a source electrode of the third transistor is grounded, a drain electrode of the fourth transistor is used as an output end of the current mirror to be connected to the output node, a source electrode of the fourth transistor is grounded, and the fourth transistor is used as a current load of the first transistor; the first capacitor module is connected to an output node of the source follower to form a first-order low-pass filter, and the bandwidth of the first-order low-pass filter is adjusted through a current mirror image proportion factor of the current mirror.
The invention has the advantages that: the filter structure of the invention can realize complex filtering functions by using few circuits, is suitable for high-speed application occasions and low-power consumption occasions, and can work in differential mode and single-ended mode.
Drawings
FIG. 1 is a schematic diagram of a prior art source follower circuit;
FIG. 2 is a schematic diagram of a prior art simple source follower based filter circuit;
FIG. 3 is a schematic diagram of a prior art source follower filter circuit based on cross-coupled positive feedback;
FIG. 4 is a schematic diagram of a source follower circuit with negative feedback of the present invention;
FIG. 5 is a schematic circuit diagram of an embodiment of a first order low pass filter of the present invention;
FIG. 6 is a schematic circuit diagram of an embodiment of a second-order low-pass filter of the present invention;
FIG. 7 is a schematic circuit diagram of an embodiment of a second-order low-pass filter without parasitic zero of the present invention;
FIG. 8 is a diagram of a second order low pass filter architecture of the present invention;
fig. 9 is a schematic diagram of a third order low pass filter circuit of the present invention.
Detailed Description
Embodiments of the present invention are described in detail below, examples of which are illustrated in the accompanying drawings, wherein the same or similar reference numerals refer to the same or similar elements or elements having the same or similar functions throughout. The embodiments described below by referring to the drawings are exemplary only for explaining the present invention and are not to be construed as limiting the present invention. Furthermore, the present invention in various examples repeat reference numerals and/or letters in the various examples, which are for the purpose of brevity and clarity, does not in itself dictate a relationship between the various embodiments and/or configurations discussed.
Referring to fig. 4, a schematic diagram of a source follower circuit with negative feedback in accordance with the present invention. The source follower includes a first transistor M1, a second transistor M2, and a current mirror. The gate of the first transistor M1 is used as the input node vg of the source follower, the drain of the first transistor M is connected to a first current source It, and the source of the first transistor M is used as the output node vs of the source follower. The second transistor M2 is a folded cascode transistor of the first transistor M1, and a gate of the second transistor M2 is configured to receive a bias voltage vb, a source thereof is connected to a drain of the first transistor M1, and a drain thereof is connected to an input terminal of the current mirror. And the output end of the current mirror is connected to the output node vs. The common terminal of the drain electrode of the first transistor M1 connected to the source electrode of the second transistor M2 is used as the source follower source node vd to access the first current source It.
Specifically, the current mirror includes a third transistor M3 and a fourth transistor M4 that are co-gated and co-sourced, and a common terminal of the gate of the third transistor M3 and the gate of the fourth transistor M4 is used as the gate node vg3 of the current mirror. The drain electrode of the third transistor M3 is shorted to the gate node vg3 and then serves as an input terminal of the current mirror, while the source electrode thereof is grounded, and the third transistor M3 is a current load of the second transistor M2. The drain of the fourth transistor M4 is used as the output terminal of the current mirror, and the source thereof is grounded, and the fourth transistor M4 is the current load of the first transistor M1.
The source follower is a novel source follower structure with negative feedback. The above transistors are all NMOS transistors, and in other embodiments, the above transistors may be PMOS transistors as well.
The invention provides a new type of filter structure based on a source follower, wherein a capacitance module is added on an output node of the source follower, and meanwhile, a current mirror image proportion factor X of a current mirror of the source follower is taken as a design parameter, so that a novel first-order low-pass filter can be formed; a capacitor module is added on an output node, a capacitor module is added on a grid connection node of a transistor of a common gate and common source of a current mirror, and a current mirror proportion factor X of the current mirror is taken as a design parameter, so that a novel second-order low-pass filter can be formed; the output node is added with a capacitor module, the grid connection node of the transistor of the common grid and the common source of the current mirror is added with a capacitor module, the source of the folded common grid transistor of the input transistor of the source follower is added with a capacitor module, and meanwhile, the current mirror image proportion factor X of the current mirror is used as a design parameter, so that the compact three-order low-pass filter can be formed. Based on the first-order, second-order and third-order low-pass filters, the high-order filter can be easily realized, for example, the series connection of the second-order low-pass filters can generate even-order filters, and the addition of the additional first-order or third-order low-pass filters can generate different combinations of odd-order filters. The first and second order low pass filter structures are connected in series and parallel, level shifted blocks can be added to the filter circuit. The filter structure of the invention can realize complex filtering functions by using few circuits, is suitable for high-speed application occasions and low-power consumption occasions, and can work in differential mode and single-ended mode.
Referring to fig. 5, a schematic circuit diagram of an embodiment of a first order low pass filter of the present invention is shown. The first-order low-pass filter comprises a source follower and a first capacitor C1, the source follower comprises a first transistor M1, a second transistor M2 and a current mirror, and the bandwidth of the first-order low-pass filter is adjusted through a current mirror proportion factor X of the current mirror.
The gate of the first transistor M1 is used as the input node vg of the source follower, the drain of the first transistor M is connected to a first current source It, and the source of the first transistor M is used as the output node vs of the source follower. The first current source It is connected to the power supply VCC.
The second transistor M2 is a folded common-gate transistor of the first transistor M1, the gate of the second transistor M2 is configured to receive a bias voltage vb, the source of the second transistor M2 is shorted to the drain of the first transistor M1 and then connected to the first current source It, and the drain of the second transistor M2 is connected to the input end of the current mirror. Wherein the second transistor M2 is an optional component, and a similar first-order low-pass filter can be obtained by removing the second transistor M2 or by connecting one or more transistors sharing the gate with the second transistor M2 in series with the second transistor M2.
The current mirror includes a third transistor M3 and a fourth transistor M4 that are cascode. The common terminal of the gate of the third transistor M3 and the gate of the fourth transistor M4 serves as the gate node vg3 of the current mirror. The third transistor M3 is a diode-connected transistor, the drain of which is shorted to the gate node vg3 and is electrically connected to the drain of the second transistor M2 as the input terminal of the current mirror, and the source of which is grounded, and the third transistor M3 is used as the current load of the second transistor M2. The drain of the fourth transistor M4 is connected to the output node vs as the output terminal of the current mirror, and the source thereof is grounded, and the fourth transistor M4 is used as the current load of the first transistor M1. Optionally, one or more transistors connected in series with the second transistor M2 and co-gated with the second transistor M2 (i.e. one or more transistors connected in series with the second transistor M2 and co-gated with the second transistor M2) are provided between the first current source It and the drain of the third transistor M3, and a similar first order low pass filter can be obtained.
The above transistors are all NMOS transistors, and in other embodiments, the above transistors may be PMOS transistors, bipolar transistors (NPN, PNP), BICMOS, etc.
The output node vs of the first capacitor C1 connected to the source follower forms the first-order low-pass filter, and the other end of the first capacitor C1 is grounded. In this embodiment, the first capacitor module is a first capacitor C1, and in other embodiments, the first capacitor module may be a capacitor array, so that the first-order low-pass filter is a tunable filter.
The characteristics and advantages of the first-order low-pass filter of the invention are further described below through theoretical derivation of the small signal.
First, gm1 and gds1 are defined as the small signal transconductance and the output conductance of the first transistor M1 (i.e., the input transistor of the source follower), respectively. Next, gm4 and gds4 are defined as the small signal transconductance and the output conductance of the fourth transistor M4, respectively. Then, gm3 and gds3 are defined as the small signal transconductance and the output conductance of the third transistor M3, respectively, the small signal transconductance gds3 can be ignored first, since it will have little effect. In small signal analysis, it is necessary to derive the small signal transfer function from the input node vg to the output node vs.
By small signal analysis, the small signal transfer function of the first order low pass filter of the present invention from the input node vg to the output node vs can be expressed as:
in the formula (1), X is a current mirror scaling factor, gm1 is a small signal transconductance of the first transistor M1, gds1 is an output conductance of the first transistor M1, gm4 is a small signal transconductance of the fourth transistor M4, gds4 is an output conductance of the fourth transistor M4, C1 is a capacitance value of the first capacitor, and S is a transfer function.
The DC gain g0 and 3dB bandwidth ω3dB of the transfer function of equation (1) may be expressed as:
ω3dB＝((1+X)gm1+gds4)/C1 (3)
it will be appreciated that the circuit of figure 5 forms a novel first order low pass filter. The dc gain and 3dB bandwidth of this novel first order low pass filter is very different from the existing source follower shown in fig. 3.
According to the existing small signal transfer function of the source follower from the input node vg to the output node vs, the dc gain g0 and the 3dB bandwidth ω3dB can be expressed as:
ω3dB＝(gm1+gds4+gds4)/C1 (5)
assume that the first transistor M1 in fig. 5 and 3 have the same transconductance gm1. As can be seen from comparing the formula (3) with the formula (5), the 3dB bandwidth of the novel first-order low-pass filter provided by the invention is (1+X) times that of the existing source follower. Meanwhile, as can be seen from comparing the formula (3) with the formula (5), for a reasonable X value, the DC gain attenuation of the novel first-order low-pass filter provided by the invention is smaller than that of the conventional source follower.
In order to make the performance comparison of the first order low pass filter of the present invention and the existing source follower two structures more reasonable, we limit the total current of the two structures to be the same (the value is) and the first transistor M1 to be the same.
For the first order low pass filter of the present invention, considering that the folded cascode transistor M2 requires a part of the current, the load current of the first transistor M1 can be expressed as:
the small signal transconductance gm1 of the first transistor M1 of the first order low pass filter of the present invention can be expressed as:
the 3dB bandwidth ω3dB of the first order low pass filter of the present invention can therefore be expressed as:
for the existing source follower, the small signal transconductance gm1 of the first transistor M1 can be expressed as
The 3dB bandwidth ω3dB of an existing source follower can be expressed as:
as can be seen from the equation (8) and the equation (10), for a given total current It and the same size of the first transistor M1, the ratio of the 3dB bandwidth of the first-order low-pass filter of the present invention to the 3dB bandwidth of the existing source follower is as follows:
thus, the current mirror scaling factor X of the current mirror can be sized to optimize the bandwidth of the first order low pass filter of the present invention under the same conditions of the same total current It and the same first transistor M1. For example, when x=1, the first order low pass filter bandwidth of the present invention increases by 41.4%; when x=2, the bandwidth of the first order low pass filter of the present invention increases by 145%. From the above analysis, it can be seen that the bandwidth of the first order low pass filter of the present invention is improved over the prior art source follower shown in fig. 3, as long as X > 0.618. Meanwhile, the DC gain attenuation performance of the filter structure is improved. Although the above analysis is based on CMOS, it is within the scope of the invention that it may be implemented in a similar manner using other techniques.
Referring to fig. 6, a schematic circuit diagram of an embodiment of a second-order low-pass filter according to the present invention is shown. The difference from the embodiment shown in fig. 5 is that in this embodiment, a first capacitor C1 is connected to an input node vg of a source follower, and a second capacitor C3 is connected to a gate node of a current mirror of the source follower, so as to form the second-order low-pass filter, the other end of the first capacitor C1 is grounded, the other end of the second capacitor C3 is grounded, and the natural frequency and the quality factor of the second-order low-pass filter are adjusted by a current mirror scaling factor X of the current mirror. As in the embodiment shown in fig. 5, the first transistor M1 is an input transistor of a source follower and the second transistor M2 is a folded cascode transistor of the first transistor M1; the third transistor M3 and the fourth transistor M4 of the common gate common source constitute a current mirror, the gate node vg3 of the current mirror being the common terminal of the gate of the third transistor M3 and the gate of the fourth transistor M4, the third transistor M3 being a diode connected transistor and being the current load of the second transistor M2, the fourth transistor M4 being a mirror image of the third transistor M3 and being the current load of the first transistor M1. As can be seen from fig. 6, the second-order low-pass filter of the present invention is based on the first-order low-pass filter of the present invention, and a second capacitor C3 is added to the gate node vg3 of the current mirror.
The characteristics and advantages of the second-order low-pass filter of the invention are further described below through theoretical derivation of small signals.
Gm1 and gds1 are defined as the small signal transconductance and the output conductance of the first transistor M1 (i.e., the input transistor of the source follower), respectively. Next, gm4 and gds4 are defined as the small signal transconductance and the output conductance of the fourth transistor M4, respectively. Then, gm3 and gds3 are defined as the small signal transconductance and the output conductance of the third transistor M3, respectively, the small signal transconductance gds3 can be ignored first, since it will have little effect. In small signal analysis, it is necessary to derive the small signal transfer function from the input node vg to the output node vs.
By small signal analysis, the small signal transfer function of the second-order low-pass filter from the input node vg to the output node vs can be expressed as:
in the formula (12), X is a current mirror scaling factor, gm1 is a small signal transconductance of the first transistor M1, gm3 is a small signal transconductance of the third transistor M3, C1 is a capacitance value of the first capacitor, C3 is a capacitance value of the second capacitor, and S is a transfer function.
From equation (12), the circuit shown in FIG. 6 forms a novel second order low pass filter. The natural pole frequency ωn and the quality factor Q corresponding to the second order transfer function of this novel second order low pass filter can be expressed as:
the natural zero frequency ωz corresponding to the second order transfer function of this novel second order low pass filter can be expressed as:
similar to the analysis of the first order low pass filter shown in FIG. 5, the natural pole frequency of the novel second order low pass filter shown in FIG. 6 is the natural pole frequency of the second order filter formed by the series connection of source followers shown in FIG. 3Multiple times. As long as X is greater than 0.25, the natural pole frequency of the novel second order low pass filter shown in fig. 6 is greater than the natural pole frequency of the second order filter formed by the series connection of source followers shown in fig. 3. The natural zero frequency of the novel second order low pass filter shown in fig. 6 can be eliminated in series (see specifically fig. 7 and its description). Quality factor of second-order low-pass filtersub-Q is a very important parameter affecting the second order system, requiring good control. As shown in equation (14), the quality factor Q of the novel second-order low-pass filter shown in fig. 6 is determined based on the ratio between the capacitances and between the transconductances of the transistors, i.e. the quality factor Q is related only to the ratio of the capacitances and the transconductances of the transistors and is therefore a constant that is weakly related to the production process and can be adjusted by the current mirror scaling factor X of the current mirror and can thus be very well controlled, which is very important for obtaining a high-performance filter.
If we limit c1=c3, then equation (14) can be expressed as:
if gm1 = gm3 is further defined, the quality factor Q can be reduced to:
the quality factor of the second order low pass filter depends only on the current mirror scaling factor X under the specific conditions of gm1=gm3.
In the usual case where gm1 is not equal to gm3, we further derive a simplified formula for the quality factor Q. As defined above, the current mirror scaling factor X is the load current ratio of the first transistor M1 to the third transistor M3, and thus the ratio of the small signal transconductance of the first transistor M1 to the small signal transconductance of the third transistor M3 can be simplified as follows:
the quality factor Q of equation (16) may be reduced as follows:
that is, when c1=c3, the quality factor Q is related only to the current mirror scaling factor X, independent of other parameters of the filter. When X is greater than 0.1, the quality factor Q can be further simplified as follows:
in combination with the above analysis of the quality factor Q, the quality factor Q of the second-order low-pass filter of the present invention is related to the current mirror scaling factor X only, on the premise that c1=c3. As long as X >0.1, the quality factor Q of the second-order low-pass filter of the present invention has a similar expression even if gm1 is not equal to gm2, as shown in formula (20).
Alternatively, in the second-order low-pass filter shown in fig. 6, the second-order band-pass filter is formed with the input node vg as an input terminal and the gate node vg3 (instead of the output node vs) of the second-order low-pass filter as an output terminal. By a similar small signal analysis, the transfer function from the input vg to the output vg3 can be expressed as:
in the formula (21), X is a current mirror scaling factor, gm1 is a small signal transconductance of the first transistor M1, gm3 is a small signal transconductance of the third transistor M3, C1 is a capacitance value of the first capacitor, C3 is a capacitance value of the second capacitor, and S is a transfer function.
From equation (21), this is the transfer function of a second order bandpass filter, i.e., the above structure forms a novel second order bandpass filter. It has the same natural frequency and quality factor as the second order low pass filter (second order low pass filter) analyzed above, and the performance is also similar to the second order low pass filter analyzed above.
Fig. 7 is a schematic circuit diagram of an embodiment of a second-order low-pass filter without parasitic zero according to the present invention. The difference from the embodiment shown in fig. 6 is that in this embodiment, a source follower with negative feedback is connected in series to the output node vs of the second-order low-pass filter. Since the second order low pass filter shown in fig. 6 has one zero, as shown in equation (12) and equation (15). If the zero point of the left half plane needs to be removed, one method is to simply stack (connect in series) a first-order low-pass filter or a source follower (which can be the existing source follower or the source follower with negative feedback) on the basis of the second-order low-pass filter shown in fig. 6, so as to form a second-order low-pass filter without parasitic zero point.
Referring to fig. 8, the second order low pass filter architecture of the present invention is shown. The second-order low-pass filter 81 and the second-order band-pass filter 82 are summed to form a high-frequency gain controllable second-order low-pass filter. The second-order low-pass filter 81 receives the low-voltage Vlp, the second-order band-pass filter 82 receives the band-pass voltage Vbp, and the sum of the two voltages obtains the high-frequency gain voltage Vboost. The second-order band-pass filter 82 is configured as a second-order band-pass filter formed by taking the input node vg as an input end and the gate node vg3 as an output end in the second-order low-pass filter shown in fig. 6. The second-order low-pass filter 81 may be configured as the second-order low-pass filter shown in fig. 6. The second-order low-pass filter 81 may be configured by connecting the second-order low-pass filter of the present invention and the first-order low-pass filter of the present invention in series (or connecting the existing source follower or the source follower with negative feedback of the present invention) as shown in fig. 7.
The same second-order low-pass filter can be shared, a second-order band-pass filter is obtained by outputting the grid node vg3 of the second-order low-pass filter, a second-order low-pass filter with parasitic zero is obtained by outputting the grid node vg3 of the second-order low-pass filter, and the second-order low-pass filter is connected with a first-order low-pass filter or the existing source follower or the source follower with negative feedback in series, so that the second-order low-pass filter without the parasitic zero is obtained after the zero is eliminated; and the second-order band-pass filter performs summation operation with a second-order low-pass filter without parasitic zero. The structure can realize a high-frequency gain controllable second-order low-pass filter which is very compact and can accurately control the high-frequency gain. Wherein the control of the high frequency gain can be expressed as:
in the formula (22), alpf is the high-frequency gain of the second-order low-pass filter without parasitic zero, abpf is the high-frequency gain of the second-order band-pass filter, and X is the current load image ratio of the filter. By designing these three parameters, the high frequency gain can be precisely controlled. For example, if x=3, alpf=1, abpf=3, boost (ωn) =4.4 dB.
Referring to fig. 9, a schematic diagram of a third order low pass filter circuit of the present invention. The difference from the embodiment shown in fig. 6 is that in this embodiment, the input node vg of the source follower is connected to the first capacitor C1, the gate node of the current mirror of the source follower is connected to the second capacitor C3, and the source of the folded common-gate transistor is connected to the third capacitor C2, so as to form the third-order low-pass filter, the other end of the first capacitor C1 is grounded, the other end of the second capacitor C3 is grounded, and the other end of the third capacitor C2 is grounded. As in the embodiment shown in fig. 6, the first transistor M1 is an input transistor of a source follower, and the second transistor M2 is a folded cascode transistor of the first transistor M1; the third transistor M3 and the fourth transistor M4 of the common gate common source constitute a current mirror, the current mirror has a current mirror scaling factor X, the common terminal of the gate of the third transistor M3 and the gate of the fourth transistor M4 serves as the gate node vg3 of the current mirror, the third transistor M3 is a diode connected transistor and is the current load of the second transistor M2, and the fourth transistor M4 is a mirror of the third transistor M3 and is the current load of the first transistor M1. As can be seen from fig. 9, the third-order low-pass filter of the present invention is based on the second-order low-pass filter of the present invention, and a third capacitor C2 is connected to the source of the folded common-gate transistor.
The characteristics and advantages of the third-order low-pass filter of the invention are further described below through theoretical derivation of small signals.
Gm1 and gds1 are defined as the small signal transconductance and the output conductance of the first transistor M1 (i.e., the input transistor of the source follower), respectively. Next, gm4 and gds4 are defined as the small signal transconductance and the output conductance of the fourth transistor M4, respectively. Then, gm3 and gds3 are defined as the small signal transconductance and the output conductance of the third transistor M3, respectively, the small signal transconductance gds3 can be ignored first, since it will have little effect. In small signal analysis, it is necessary to derive the small signal transfer function from the input node vg to the output node vs.
By small signal analysis, the small signal transfer function of the second-order low-pass filter from the input node vg to the output node vs can be expressed as:
in the formula (23), X is a current mirror scaling factor, gm1 is a small signal transconductance of the first transistor M1, gm2 is a small signal transconductance of the second transistor M2, gm3 is a small signal transconductance of the third transistor M3, C1 is a capacitance value of the first capacitor, C3 is a capacitance value of the second capacitor, C2 is a capacitance value of the third capacitor, and S is a transfer function.
From equation (23), the circuit shown in fig. 9 forms a novel third order low pass filter. If c2=0 (i.e. no third capacitor C2 is provided), the transfer function of equation (23) is simplified to equation (12), i.e. the second order low pass filter transfer function shown in fig. 6. The second order low pass filter structure shown in fig. 6 is only a specific example of the third order low pass filter structure shown in fig. 9.
Alternatively, in the third-order low-pass filter shown in fig. 9, the input node vg is used as an input terminal, and the gate node vg3 (instead of the output node vs) of the third-order low-pass filter is used as an output terminal, so as to form a third-order band-pass filter. The transfer function has the same denominator as equation (23), and is not described here.
In the third-order low-pass filter, the input node is used as an input end, and the grid node of the third-order low-pass filter is used as an output end, so that a third-order band-pass filter is formed; and carrying out summation operation on the third-order band-pass filter and the third-order low-pass filter to form the high-frequency gain controllable third-order low-pass filter.
Similar to the analysis of the high-frequency gain controllable second-order low-pass filter of fig. 8, in the third-order low-pass filter shown in fig. 9, the same third-order low-pass filter can be shared: the third-order low-pass filter with parasitic zero is obtained from the output node vs of the third-order low-pass filter, the third-order band-pass filter is obtained from the output of the grid node vg3 of the third-order low-pass filter, and the third-order band-pass filter and the third-order low-pass filter are summed, so that the structure can realize a very compact high-frequency gain controllable third-order low-pass filter capable of accurately controlling the high-frequency gain. Preferably, after the third-order low-pass filter with parasitic zero is obtained from the output node vs of the third-order low-pass filter, a first-order low-pass filter or the existing source follower or the source follower with negative feedback can be connected in series, and the third-order low-pass filter without parasitic zero is obtained after the zero is eliminated.
The foregoing is merely a preferred embodiment of the present invention and it should be noted that modifications and adaptations to those skilled in the art may be made without departing from the principles of the present invention, which are intended to be comprehended within the scope of the present invention.
Claims (7)
1. A filter structure, characterized in that the filter structure comprises a source follower and a first capacitor module;
the source follower comprises a first transistor and a current mirror;
the grid electrode of the first transistor is used as an input node of the source electrode follower, the drain electrode of the first transistor is connected with a first current source, and the source electrode of the first transistor is used as an output node of the source electrode follower;
the current mirror comprises a third transistor and a fourth transistor which are in common gate and common source, wherein a common end, which is connected with a grid electrode of the third transistor, is used as a grid electrode node of the current mirror, a drain electrode of the third transistor is short-circuited with the grid electrode node and then is used as an input end of the current mirror to be coupled to the first current source, a source electrode of the third transistor is grounded, a drain electrode of the fourth transistor is used as an output end of the current mirror to be connected to the output node, a source electrode of the fourth transistor is grounded, and the fourth transistor is used as a current load of the first transistor;
the first capacitor module is connected to an output node of the source follower to form a first-order low-pass filter, and the bandwidth of the first-order low-pass filter is adjusted through a current mirror image proportion factor of the current mirror;
the filter structure further comprises a second capacitor module, wherein the second capacitor module is connected with the grid node to form a second-order low-pass filter, and the natural frequency and the quality factor of the second-order low-pass filter are adjusted through the current mirror image proportion factor of the current mirror;
in the second-order low-pass filter, the input node is taken as an input end, and the grid node of the second-order low-pass filter is taken as an output end, so that a second-order band-pass filter is formed;
the second-order low-pass filter and the first-order low-pass filter or the source follower are connected in series to form a second-order low-pass filter without parasitic zero; and carrying out summation operation on the second-order band-pass filter and the second-order low-pass filter without parasitic zero points to form the high-frequency gain controllable second-order low-pass filter.
2. The filter structure of claim 1, wherein the source follower further comprises a second transistor, the second transistor is a folded common gate transistor of the first transistor, a gate of the second transistor is configured to receive a bias voltage, a source of the second transistor is shorted to a drain of the first transistor and connected to the first current source, a drain of the second transistor is connected to an input terminal of the current mirror, and the third transistor is configured to act as a current load of the second transistor.
3. The filter structure of claim 2 wherein one or more transistors in series with and co-gate with the second transistor are provided between the first current source and the drain of the third transistor.
4. The filter structure of claim 2, wherein in the second-order low-pass filter, the filter further comprises a third capacitor module, and the third capacitor module is connected to the source of the folded common-gate transistor of the second-order low-pass filter to form a third-order low-pass filter.
5. The filter structure of claim 4, wherein in the third-order low-pass filter, a third-order band-pass filter is formed with the input node as an input terminal and the gate node as an output node.
6. The filter structure of claim 4, wherein in the third-order low-pass filter, a third-order band-pass filter is formed with the input node as an input terminal and the gate node of the third-order low-pass filter as an output terminal; and carrying out summation operation on the third-order band-pass filter and the third-order low-pass filter to form the high-frequency gain controllable third-order low-pass filter.
7. The filter structure of claim 1, wherein the capacitive module is a capacitor or an array of capacitors.
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| CN110212876A (en) * | 2019-06-24 | 2019-09-06 | 南京芯耐特半导体有限公司 | Follower circuit structure with built-in negative-feedback |
| CN111124032B (en) * | 2019-12-20 | 2021-11-05 | 睿兴科技(南京)有限公司 | Filter circuit for suppressing noise interference and micro control system |
| CN115051702B (en) * | 2022-08-16 | 2022-11-22 | 英彼森半导体(珠海)有限公司 | Wide-swing low-power-consumption source follower |
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