CN109546913A - A kind of capacitor miniaturization motor driver - Google Patents
A kind of capacitor miniaturization motor driver Download PDFInfo
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- CN109546913A CN109546913A CN201811581927.3A CN201811581927A CN109546913A CN 109546913 A CN109546913 A CN 109546913A CN 201811581927 A CN201811581927 A CN 201811581927A CN 109546913 A CN109546913 A CN 109546913A
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- 239000003990 capacitor Substances 0.000 title claims abstract description 25
- 238000006243 chemical reaction Methods 0.000 claims abstract description 43
- 230000001360 synchronised effect Effects 0.000 claims abstract description 5
- 230000005347 demagnetization Effects 0.000 claims description 4
- 238000011897 real-time detection Methods 0.000 claims description 3
- 238000010586 diagram Methods 0.000 description 10
- 238000000034 method Methods 0.000 description 4
- 230000001105 regulatory effect Effects 0.000 description 4
- 230000009466 transformation Effects 0.000 description 4
- 239000003985 ceramic capacitor Substances 0.000 description 3
- 238000001514 detection method Methods 0.000 description 3
- 230000004907 flux Effects 0.000 description 3
- 239000010409 thin film Substances 0.000 description 3
- 230000003313 weakening effect Effects 0.000 description 3
- 230000033228 biological regulation Effects 0.000 description 2
- 238000005457 optimization Methods 0.000 description 2
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Classifications
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/05—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for damping motor oscillations, e.g. for reducing hunting
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
- H02P21/141—Flux estimation
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
- H02P21/18—Estimation of position or speed
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/22—Current control, e.g. using a current control loop
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Abstract
The invention proposes a kind of capacitors to minimize motor driver, which includes control unit, inductor, AC-DC converting circuit, DC link section and DC conversion circuit.Supply voltage v of the ac-dc converter circuit to AC power sourceinFull-wave rectification is carried out, which has the capacitor in parallel with the outlet side of the ac-dc converter circuit, and the DC voltage v of output rippledc, after which is converted into exchange for the output of the DC link section using switch, connected permanent magnet synchronous motor is supplied, which controls the switch.The present invention detects motor speed, and it is fluctuated according to compressor rotary speed and calculates motor torque compensation, inhibit load torque fluctuation, control compressor even running, further, the anti-saturation control for being characterized in that introducing of this driving device being capable of volitional check voltage output when voltage instruction exceeds the output area of driver.
Description
Technical Field
The invention belongs to the technical field of motor driving, and particularly relates to a capacitor miniaturized motor driving device.
Background
Along with the improvement of energy conservation requirements of consumers on electromechanical products, the variable frequency motor driver with higher efficiency is more and more widely applied. The direct current bus voltage of the conventional variable frequency driver is in a stable state, and the inversion part is relatively independent from the input alternating current voltage, so that the control of the inversion part does not need to consider the instantaneous change of the input voltage, and the control method is convenient to realize. However, this design method requires an electrolytic capacitor with a large capacitance, which increases the size and cost of the driver. In addition, electrolytic capacitors have a limited lifetime and their effective operating time tends to be a bottleneck for the lifetime of the drive.
In order to solve the problems, a strategy of replacing an electrolytic capacitor with a thin film capacitor or a ceramic capacitor with a small capacitance value is provided in the related scheme, compared with a conventional AC/DC/AC driving circuit, a PFC part is omitted, and the miniaturized capacitor can not only realize cost reduction, but also eliminate service life bottleneck caused by the electrolytic capacitor. However, the capacitance value of the thin film capacitor or the ceramic capacitor on the direct-current bus voltage is very small, and is usually only 1% -2% of that of the conventional high-voltage electrolytic capacitor; the voltage of the direct current bus fluctuates greatly along with the input voltage of the power supply, the lowest voltage is only dozens of volts, and the lowest voltage of the direct current bus needs to be controlled to ensure the stability of a control system; further, when the inverter works, the capacitance of the bus and the inductance L on the ac power supply side generate LC resonance, which causes large control instability of system harmonic, and a special control strategy needs to be added to solve the problem, eliminate LC resonance, and realize stable operation of the compressor.
Disclosure of Invention
The invention provides a capacitor miniaturization motor driving device for overcoming the defects in the prior art. The invention can automatically track the fluctuating load and carry out the anti-saturation control of the system, thereby ensuring the stability of the speed regulating system.
The purpose of the invention is realized by the following technical scheme: a capacitive miniaturized motor drive apparatus comprising: a control unit 2, an inductor 3, an ac/dc conversion circuit 4, a dc link unit 5, and an ac/dc conversion circuit 6; the AC-DC conversion circuit 4 is used for supplying power voltage v to the AC power supply 1inFull-wave rectification is performed, one end of the inductor 3 is connected to an alternating current power supply 1, the other end is connected to an alternating current/direct current conversion circuit 4, the direct current link part 5 has a capacitor 5a connected in parallel to the output side of the alternating current/direct current conversion circuit 4, and outputs a pulsating direct current voltage vdcThe dc/ac conversion circuit 6 converts the output of the dc link unit 5 into ac by a switch and supplies the ac to the permanent magnet synchronous motor 7 connected thereto, and the control unit 2 receives a speed commandDetecting voltage v of input powerinPhase thetageCurrent iinDC bus voltage vdcAnd motor current iu、iv、iwControl finger of output DC/AC conversion circuit 6Let Tu、Tv、TwThe motor control is realized;
the control part 2 comprises a torque compensation module for detecting the speed fluctuation of the motor rotor and calculating a torque compensation value according to the speed and torque fluctuation:
ωerip＝ωe-ωeref
ωmrip＝ωerip/p
wherein, ω iseFor estimating electromagnetic speed, omega, of electric machineserefRepresenting the motor command electromagnetic speed, p representing the motor pole pair number, K being the torque compensation gain coefficient, omegamripIn order for the mechanical rotational speed to fluctuate,to the mechanical rotational speed, AωcAnd AωsAre all fundamental coefficients, HPF denotes a high pass filter, ThfRepresenting the high pass filter delay time, ωmIndicating the motor speed.
Further, the control section 2 further includes a waveform generator module according to vin、θgeCalculating the waveform of the Q-axis current waveform generator according to the motor load; the Q-axis current waveform generator waveform has two shapes, including:
waveform generator shape 1:
waveform generator shape 2:
wherein, Wf(θge) As output variable, vinFor real-time detection of the value of the supply voltage, VθdTo this end the phase of the mains voltage within a half period of the mains voltage is thetadVoltage of time, VPeakIs the magnitude of the supply voltage, θgeFor the phase estimate of the input voltage, thetadThe phase corresponding to the current dead zone;
the shape of the waveform generator is determined according to the strategy of using the waveform generator.
Further, the waveform generator usage strategy includes:
when motor frequency omegae>ωhighThe waveform generator shape 2 is selected when the motor frequency omegae<ωlowThe waveform generator shape 1 is selected when ω islow≦ωe≦ωhighKeeping the current waveform generator unchanged; or when the DC-AC conversion circuit 6 outputs power Pinv>PhighThe waveform generator shape 2 is selected when the DC-AC conversion circuit 6 outputs the power Pinv<PlowThe waveform generator shape 1 is selected when P islow≦Pinv≦PhighKeeping the current waveform generator unchanged;
the power of the direct-alternating current conversion circuit 6 is calculated according to the following formula:
Pinv＝Vuiu+Vviv+Vwiw
wherein, Vu，Vv，VwAre respectively provided withIs a three-phase voltage command i of the DC-AC conversion circuit 6u, v and wu、iv、iwThe three-phase actual current of the motor is respectively.
Further, the Q-axis current initial command value is calculated by the following formula:
in the formula TpIndicating a torque command, iq_ref0Indicates the initial command value of the Q-axis current,representing the rotor speed estimate, keIs the motor back electromotive force coefficient, Ld、LqDivided into DQ-axis inductances id_refIs a D-axis current command value, KPAs a proportional coefficient of the controller, KiIs the integral coefficient of the controller.
Further, the control part 2 further comprises a capacitance current compensation module for calculating capacitance power Pc：
Compensated current command iqccComprises the following steps:
wherein, thetageC is the capacitance value of a capacitor connected in parallel between the input ends of the DC-AC conversion circuit 6, V is the phase estimation value of the input voltagePeakIs the amplitude of the voltage, omega, of the AC power supplyinIs the voltage frequency of the AC power supply, p is the number of pole pairs of the motor, omegaeIs the motor rotor speed.
Further, the total current command value of the Q axis is:
iq_ref1＝iq_ref0+iqcc。
further, the control part 2 further includes a weak magnetic control module, the weak magnetic control module includes a weak magnetic controller and a limiting unit, the weak magnetic controller is configured to calculate a D-axis current instruction initial value:
wherein, Id0Is the initial value of the D-axis current command, KiIn order to integrate the control coefficients of the motor,V1is the output voltage amplitude, v, of the DC-AC conversion circuit 6dIs D-axis voltage, vqIs the Q-axis voltage, VmaxIs the maximum output voltage, V, of the DC-AC converter circuit 6dcIs the dc bus voltage of the motor.
Further, the clipping unit performs clipping processing on the D-axis current instruction initial value to obtain a D-axis current instruction:
wherein, IdemagFor the demagnetization current limit value of the motor, ImtpaControl of corresponding D-axis Current value, i, for MTPAq_ref1Is the total current command value of the Q axis, keIs the motor back electromotive force coefficient.
Further, the control part 2 further comprises a current amplitude limiting control module for realizing the limitation of the DQ output current; the final DQ axis current command is calculated according to the following formula:
wherein imaxThe maximum current value allowed to be output by the dc/ac conversion circuit 6.
Further, the control section 2 obtains a final DQ-axis current command id_refAnd iq_refAnd detecting and calculating the actual current i of the DQdAnd iqRespectively carrying out PI control on the D-axis current and the Q-axis current, and then adding decoupling and calculating to obtain a DQ axis voltage command VdAnd VqAnd αβ axis voltage command V is obtained by coordinate conversionαAnd VβThen converted into a u, V and w three-phase voltage command Vu、Vv、VwFinally, the sum V is calculatedu、Vv、VwEquivalent pulse Tu、Tv、TwAnd output to the motor through a dc-ac conversion circuit 6.
Drawings
FIG. 1 is a block diagram of a capacitive miniaturized motor driving apparatus according to the present invention;
FIG. 2 is a block diagram of a control structure of the capacitor miniaturized motor driving apparatus according to the present invention;
FIG. 3 is a phase locked loop block diagram;
FIG. 4 is a torque compensation module control block diagram;
FIG. 5 is a block diagram of a field weakening current control module;
FIG. 6 is a block diagram of a DQ axis current clipping control module;
FIG. 7 is a block diagram of a DQ axis voltage generation and anti-saturation control module.
Detailed Description
The technical solutions in the embodiments of the present invention will be described clearly and completely with reference to the accompanying drawings in the embodiments of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention.
Fig. 1 is a schematic structural diagram of a capacitor-miniaturized motor driving apparatus according to an embodiment of the present invention.
It should be noted that the capacitor miniaturized motor driving apparatus according to the embodiment of the present invention may be applied to an inverter motor, and referring to fig. 1, in a circuit of the inverter motor, an AC power supply AC is connected to the motor through a rectifier circuit and an inverter circuit, and in the embodiment of the present invention, a thin film capacitor or a ceramic capacitor 5a having a small capacitance value may be connected in parallel between input terminals of the inverter circuit.
As shown in fig. 1, the capacitor-miniaturized motor driving apparatus according to the embodiment of the present invention includes: a control unit 2, an inductor 3, an ac-dc converter circuit 4, a dc link unit 5, and a dc-ac converter circuit 6.
In fig. 1, a module 1 is a system power supply, and a module 7 is an equivalent circuit diagram of a permanent magnet synchronous motor. The AC-DC conversion circuit 4 is used for supplying power voltage v to the AC power supply 1inFull-wave rectification is performed, and the dc link unit 5 has a capacitor 5a connected in parallel with the output side of the ac/dc conversion circuit 4 and outputs a pulsating dc voltage vdcThe dc/ac conversion circuit 6 converts the output of the dc link unit 5 into ac by a switch, supplies the ac to the connected permanent magnet synchronous motor 7, and controls the switch by the control unit 2. The control part2 for receiving speed instructionDetecting voltage v of input powerinPhase thetageCurrent iinDC bus voltage vdcAnd motor current iu、iv、iwAnd outputs a control instruction T of the DC/AC conversion circuit 6u、Tv、TwAnd motor control is realized. The direct-alternating current conversion circuit 6 is an inverter circuit.
FIG. 3 illustrates an input voltage phase detection PLL module for obtaining an instantaneous voltage value V of an input AC power sourcegeAnd according to the instantaneous value V of the voltage of the AC power supplygeCalculating an input voltage phase estimate θge。
Specifically, as shown in fig. 3, the input voltage phase detection phase-locked loop module may include a cosine calculator, a first multiplier, a low-pass filter, a first PI regulator, and an integrator. Wherein the cosine calculator is used for estimating the phase of the input voltage in the last calculation periodgePerforming cosine calculation to obtain a first calculated value, and a first multiplier for multiplying the instantaneous voltage value V of the AC power supplygeThe first calculated value is multiplied by the second calculated value to obtain a second calculated value. The low-pass filter is used for low-pass filtering the second calculated value to obtain a third calculated value, wherein the bandwidth of the low-pass filter is lower than the voltage frequency of the alternating current power supply, and in one embodiment of the invention, the bandwidth of the low-pass filter is lower than the voltage frequency omega of the alternating current power supplyg1/5 of (1). The first PI regulator is used for performing PI regulation on the third calculated value to output a fourth calculated value, and the integrator is used for performing PI regulation on the fourth calculated value and the voltage frequency omega of the alternating current power supplygThe sum is subjected to integral calculation to obtain an input voltage phase estimation value theta of the current calculation periodge。
A position/velocity estimator estimates a rotor position of the electric machine to obtain a rotor angle estimateAnd rotor speed estimate
First, the estimated value of the effective magnetic flux of the motor in the axial direction of the fixed coordinate systems α and β can be calculated according to the current and the voltage on the fixed coordinate systems, and the specific calculation formula is as follows:
wherein,andestimated values of the effective flux in the α and β axial directions, v, respectively, for the motorαAnd vβVoltage in the direction of the α and β axes, iαAnd iβCurrent in the direction of the α and β axes, respectively.
The rotor angle estimate is then further calculatedAnd rotor speed estimateThe specific calculation formula is as follows:
wherein, Kp_pllAnd Ki_pllProportional and integral parameters, ω, of the phase-locked loop PI controller, respectivelyfIs the bandwidth of the velocity low pass filter, θerrIs an estimate of the deviation angle.
As shown in fig. 2, the Q-axis current instruction calculation module includes a second PI regulator, a waveform generator, an initial current calculation unit, a capacitance current compensation unit, and a superposition unit.
The Q-axis current command calculation module includes:
a second PI regulator for PI regulating a difference between the motor speed command and the rotor speed estimate to output a torque amplitude command;
a waveform generator for generating an output variable from the input voltage phase estimate;
an initial current calculation unit, wherein the initial current calculation unit is used for multiplying the output variable by the torque amplitude instruction and then dividing the multiplied output variable by a motor torque coefficient to obtain a Q-axis current instruction initial value;
the capacitance current compensation unit is used for generating compensation current according to the input voltage phase estimation value;
a superimposing unit configured to superimpose the compensation current on the Q-axis current instruction initial value to obtain the Q-axis current instruction.
Wherein the second PI regulator is used for regulating the command speedAnd estimating the velocityPerforming PI control after difference is made, and outputting a torque command Tp。
Wherein KPAs a proportional coefficient of the controller, KiIs the integral coefficient of the controller.
The waveform generator is used for generating an output variable W according to the shape and phase of an input voltagef。
An initial current calculating unit for calculating an output variable WfAnd torque amplitude command TpAfter multiplication, the value is further converted into an initial value i of a Q-axis current commandq_ref0。
In the formula TpIndicating a torque command, iq_ref0Indicates the initial command value of the Q-axis current,representing the rotor speed estimate, keIs the motor back electromotive force coefficient, Ld、LqDivided into DQ-axis inductances id_refIs a D-axis current command value,
in an embodiment of the present invention, two shape waveform generators are included:
waveform generator shape 1 calculates the output variable according to the following formula:
waveform generator shape 2 calculates the output variable according to the following equation:
wherein, Wf(θge) Is the output variable, vinFor real-time detection of the value of the supply voltage, VθdTo this end the phase of the mains voltage within a half period of the mains voltage is thetadVoltage of time, VPeakIs the magnitude of the supply voltage, θgeFor said input voltage phase estimate, θdThe phase corresponding to the current dead zone.
The waveform generator 1 has the advantages that the input current harmonic wave is small; the disadvantage is that the peak value of the motor phase current is large.
Compared with the waveform generator 1, the waveform generator 2 has the disadvantages of large input current harmonic and small motor phase current peak value.
When in specific use, the frequency omega is determined according to the running frequency of the motoreIt is decided which waveform generator to use. In particular, when the motor frequency ωe>ωhighThe waveform generator 2 is selected when the motor frequency omegae<ωlowThe waveform generator 1 is selected when ω islow≦ωe≦ωhighWhile keeping the current waveform generator unchanged. Wherein ω ishigh、ωlowThe value relation is omegahigh>ωlowIn the present embodiment, ωhighIs 150Hz, omegalowIs 130 Hz.
The specific selection method of the waveform generator can also be realized by the following ways:
when in specific use, the power P is output according to the DC-AC conversion circuit 6invIt is decided which waveform generator to use. In particular, when the motor frequency Pinv>PhighThe waveform generator 2 is selected when the motor frequency P isinv<PlowThe waveform generator 1 is selected when P islow≦Pinv≦PhighWhile keeping the current waveform generator unchanged. Wherein P ishigh、PlowThe value relationship is as follows, Phigh>PlowIn this embodiment, PhighIs 1100W, PlowIs 900W.
The power of the direct-alternating current conversion circuit 6 is calculated according to the following formula:
Pinv＝Vuiu+Vviv+Vwiw
wherein, Vu，Vv，VwAre three-phase voltage commands i of the DC-AC conversion circuit 6u, v and w respectivelyu、iv、iwThe three-phase actual current of the motor is respectively.
In an embodiment of the present invention, the capacitance current compensation unit may calculate the compensation current according to the following formula:
wherein, thetageC is capacitance value of capacitor connected in parallel between input ends of the inverter circuit, V is estimated value of phase of the input voltagePeakIs the voltage amplitude, omega, of the AC sourceinIs the voltage frequency of the AC power supply, p is the number of pole pairs of the motor, keIs the motor back electromotive force coefficient, Ld、LqDivided into DQ-axis inductances id_refIs a D-axis current command value, ωeIs the motor rotor speed.
In one embodiment of the invention, the phase parameter θ is setdThe phase corresponding to the current dead zone can be selected as 0.1-0.2 rad by default.
Referring to fig. 4, in the embodiment of the present invention, the control part 2 includes a torque compensation module for detecting the speed fluctuation of the motor rotor and calculating the torque compensation current value according to the speed and torque fluctuation to realize the ripple load suppression, specifically, according to the estimation of the motor electromagnetismSpeed of rotation omegaeAnd motor command electromagnetic speed omegaerefCalculating the electromagnetic rotation speed fluctuation of the motor and converting the electromagnetic rotation speed fluctuation into mechanical rotation speed fluctuation omegamrip。
ωerip＝ωe-ωeref
ωmrip＝ωerip/p
Wherein, ω iseFor estimating electromagnetic speed, omega, of electric machineserefRepresenting the commanded electromagnetic speed of the motor and p representing the pole pair number.
Will omegamripViewed as the mechanical speed of rotationAnd performing a fourier transform:
neglecting the high frequency waveform component more than two times, when only retaining the fundamental wave, can simplify as:
thereby obtaining a fundamental coefficient AωcAnd Aωs。
In the process of addingAngle lag compensation atan (T)hfωm) Calculating to obtain the torque compensation Tc。
Wherein K is a torque compensation gain factor.
Initial current calculation unit for TpCumulative TcPost-multiplication waveform generator WfGenerating a new torque command Te，TeDivided by the motor back emf constant keThen obtaining an initial instruction value i of the Q-axis currentq_ref0：
Q-axis current initial command value iq_ref0Adding a current instruction value i output by a capacitance current compensation moduleqccObtaining a current command value i of the Q axisq_ref1。
iq_ref1＝iq_ref0+iqcc。
With reference to fig. 5, the control unit 2 further includes a weak magnetic control module for calculating a D-axis weak magnetic current command id_ref1：
Specifically, the field weakening control module comprises: the weak magnetic controller is used for controlling the difference between the maximum output voltage of the inverter circuit and the output voltage amplitude of the inverter circuit to obtain a D-axis current instruction initial value; and the amplitude limiting unit is used for carrying out amplitude limiting processing on the D-axis current instruction initial value to obtain the D-axis current instruction.
Further, the flux weakening controller calculates the D-axis current instruction initial value according to the following formula:
wherein id0Is the initial value of the D-axis current command, KiIn order to integrate the control coefficients of the motor,V1for the output voltage amplitude of the inverter circuitValue vdIs D-axis voltage, vqIs the Q-axis voltage, VmaxIs the maximum output voltage, V, of the inverter circuitdcIs the dc bus voltage of the motor.
Further, to prevent the motor from demagnetizing, i is limitedd_ref1Can not be lower than demagnetization current Idemag. In addition, to improve the driving efficiency, the D-axis current command needs to be less than or equal to the D-axis current I corresponding to the MTPA controlmtpa. Therefore, the clipping unit obtains the D-axis current command according to the following formula:
wherein, IdemagAnd the current limit value is the demagnetization current limit value of the motor.
In conjunction with FIG. 6, a further embodiment is based on obtaining a DQ-axis current command id_ref1And iq_ref1Performing amplitude limiting control to satisfySpecifically, the final DQ axis current command is calculated according to the following formula:
wherein imaxThe maximum current value allowed to be output by the dc/ac conversion circuit 6.
Further, the present embodiment obtains a DQ-axis current command id_refAnd iq_refAnd detecting and calculating the actual current i of the DQdAnd iqRespectively carrying out PI control on the D-axis current and the Q-axis current, and then adding decoupling and calculating to obtain a DQ axis voltage command VdAnd VqAnd the control part is characterized in that anti-saturation control is introduced, when the voltage instruction exceeds the output range of the driver, the voltage output can be automatically limited, and a specific control block diagram is shown in fig. 7. FIG. 7 is a DQ axis voltage generation and anti-saturation control module, the anti-saturation control working principle is: d-axis current command id_refAnd D-axis actual current idPerforming PI control after difference making, and accumulating decoupling control-omegaeLqiqObtaining the initial voltage instruction value of the D axisSimilarly, Q-axis current command iq_refAnd D-axis actual current iqPerforming PI control after difference making, and accumulating decoupling control omegaeLdid+ωeψfObtaining the initial voltage instruction value of the Q axisComputingAndmodulo v ofmAnd calculating a final voltage command value according to the following formula:
wherein v ismaxIs the maximum voltage value that the controller can output.
Further, willFeedback to D-axis current integral, willFed back to the Q-axis current integral for compensating the voltage in the next calculation cycle.
The control unit 2 further obtains an αβ -axis voltage command V by coordinate conversionαAnd VβThen converted into a u, V and w three-phase voltage command Vu、Vv、VwFinally, the sum V is calculatedu、Vv、VwEquivalent pulse Tu、Tv、TwAnd output to the motor through the inverter.
Specifically, the Q-axis voltage command and the D-axis voltage command may be calculated according to the following formulas:
wherein, VqFor Q-axis voltage command, VdFor D-axis voltage command, KdpAnd KdiProportional gain and integral gain, K, respectively, for D-axis current controlqpAnd KqiProportional gain and integral gain, omega, respectively, for Q-axis current controleIs the motor speed, keIs the motor back electromotive force coefficient, LdAnd LqAre respectively a D-axis inductor and a Q-axis inductor,denotes the integral of x (τ) over time, VmaxIs the maximum output voltage, v, of the inverter circuitmA vector sum of the d-axis voltage command and the q-axis voltage command of the motor is shown.
Obtaining the Q-axis voltage command VqAnd D-axis voltage command VdThen, V can be adjusted according to the rotor angle theta of the motorqAnd VdCarrying out Park inverse transformation to obtain a voltage command V on a fixed coordinate systemαAnd VβThe concrete transformation formula is as follows:
Vα＝Vdcosθ-Vqsinθ
Vβ＝Vdsinθ+Vqcosθ
where θ is the motor rotor angle, where the rotor angle estimate θ can be takenest。
Further, the voltage command V on the fixed coordinate system can be usedαAnd VβPerforming Clark inverse transformation to obtain three-phase voltage command Vu、VvAnd VwThe concrete transformation formula is as follows:
Vu＝Vα
then the duty ratio calculation unit can calculate the duty ratio according to the DC bus voltage and the three-phase voltage instruction to obtain a duty ratio control signal, namely a three-phase duty ratio Tu、TvAnd TwThe specific calculation formula is as follows:
Tu＝(Vu+0.5Vdc)/Vdc
Tv＝(Vv+0.5Vdc)/Vdc
Tw＝(Vw+0.5Vdc)/Vdc
wherein, VdcIs the dc bus voltage.
The duty ratio control signal controls the switch of the inverter circuit in real time, and the control of the motor is realized.
According to the capacitor miniaturized motor driving device provided by the embodiment of the invention, relevant parameters are obtained through an input voltage phase detection phase-locked loop module, a position/speed estimator and the like, two waveform generators are designed, a load torque compensation module is designed, a Q-axis current instruction and a D-axis current instruction are calculated, then the Q-axis voltage instruction and the D-axis voltage instruction are further obtained, a duty ratio control signal is generated, and therefore an inverter circuit is controlled through the duty ratio control signal to control a motor. Therefore, the waveform generator can be automatically switched according to the system running state, the harmonic optimization and the phase current peak value optimization of the pressing machine are considered, the input current waveform of the motor meets the harmonic requirement, the load torque compensation value is calculated according to the speed fluctuation, the torque command is accumulated, and the system torque compensation and the stable running of the speed regulating system are realized.
The capacitor miniaturization motor driving device provided by the invention is described in detail, a specific example is applied in the text to explain the principle and the implementation mode of the invention, and the description of the embodiment is only used for helping to understand the method and the core idea of the invention; meanwhile, for a person skilled in the art, according to the idea of the present invention, there may be variations in the specific embodiments and the application scope, and in summary, the content of the present specification should not be construed as a limitation to the present invention.
Claims (10)
1. A capacitor-miniaturized motor drive device, comprising: a control unit (2), an inductor (3), an AC/DC conversion circuit (4), a DC link unit (5), and a DC/AC conversion circuit (6); the AC-DC conversion circuit (4) is used for supplying power voltage v to the AC power supply (1)inFull-wave rectification is performed, one end of the inductor (3) is connected to an alternating current power supply (1), the other end is connected to an alternating current/direct current conversion circuit (4), the direct current chain part (5) has a capacitor (5a) connected in parallel to the output side of the alternating current/direct current conversion circuit (4), and outputs a pulsating direct current voltage vdcAnd the DC-AC conversion powerThe output of the DC link part (5) is converted into AC by a switch on the circuit (6) and then supplied to a permanent magnet synchronous motor (7) connected with the circuit, and the control part (2) is used for receiving a speed commandDetecting voltage v of input powerinPhase thetageCurrent iinDC bus voltage vdcAnd motor current iu、iv、iwAnd outputs a control instruction T of the direct current-alternating current conversion circuit (6)u、Tv、TwThe motor control is realized;
the control part (2) comprises a torque compensation module, and is used for detecting the speed fluctuation of the motor rotor and calculating a torque compensation value according to the speed and torque fluctuation:
ωerip＝ωe-ωeref
ωmrip＝ωerip/p
wherein, ω iseFor estimating electromagnetic speed, omega, of electric machineserefRepresenting the motor command electromagnetic speed, p representing the motor pole pair number, K being the torque compensation gain coefficient, omegamripIn order for the mechanical rotational speed to fluctuate,to the mechanical rotational speed, AωcAnd AωsAre all fundamental coefficients, HPF denotes a high pass filter, ThfRepresenting the high pass filter delay time, ωmIndicating the motor speed.
2. The capacitive miniaturized motor driving device according to claim 1, wherein the control portion (2) further comprises a waveShape generator module according to vin、θgeCalculating the waveform of the Q-axis current waveform generator according to the motor load; the Q-axis current waveform generator waveform has two shapes, including:
waveform generator shape 1:
waveform generator shape 2:
wherein, Wf(θge) As output variable, vinFor real-time detection of the value of the supply voltage, VθdTo this end the phase of the mains voltage within a half period of the mains voltage is thetadVoltage of time, VPeakIs the magnitude of the supply voltage, θgeFor the phase estimate of the input voltage, thetadThe phase corresponding to the current dead zone;
the shape of the waveform generator is determined according to the strategy of using the waveform generator.
3. The capacitive miniaturized motor driver according to claim 2, wherein said waveform generator usage strategy comprises:
when motor frequency omegae>ωhighThe waveform generator shape 2 is selected when the motor frequency omegae<ωlowThe waveform generator shape 1 is selected when ω islow≦ωe≦ωhighKeeping the current waveform generator unchanged; or when the DC-AC conversion circuit (6) outputs power Pinv>PhighThe waveform generator shape 2 is selected when the output power P of the DC-AC conversion circuit (6)inv<PlowTime-selective waveform generator shapeForm 1, when Plow≦Pinv≦PhighKeeping the current waveform generator unchanged;
the power of the direct current-alternating current conversion circuit (6) is calculated according to the following formula:
Pinv＝Vuiu+Vviv+Vwiw
wherein, Vu，Vv，VwU, v and w three-phase voltage commands i of the DC-AC conversion circuit (6)u、iv、iwThe three-phase actual current of the motor is respectively.
4. The capacitor-miniaturized motor-driving device of claim 3, wherein the initial command value of the Q-axis current is calculated by the following formula:
in the formula TpIndicating a torque command, iq_ref0Indicates the initial command value of the Q-axis current,representing the rotor speed estimate, keIs the motor back electromotive force coefficient, Ld、LqDivided into DQ-axis inductances id_refIs a D-axis current command value, KPAs a proportional coefficient of the controller, KiIs the integral coefficient of the controller.
5. Capacitive miniaturised motor drive according to claim 4 characterised in that the control part (2) further comprises a capacitive current compensation module for calculating the capacitive power Pc：
Compensated current command iqccComprises the following steps:
wherein, thetageC is the capacitance value of a capacitor connected in parallel between the input ends of the DC-AC conversion circuit (6) for the phase estimation value of the input voltage, VPeakIs the amplitude of the voltage, omega, of the AC power supplyinIs the voltage frequency of the AC power supply, p is the number of pole pairs of the motor, omegaeIs the motor rotor speed.
6. The capacitor-miniaturized motor-driving device of claim 5, wherein the total Q-axis current command value is:
iq_ref1＝iq_ref0+iqcc。
7. the capacitance-type miniaturized motor driving device according to claim 6, wherein the control section (2) further includes a weak magnetic control module including a weak magnetic controller and a clipping unit, the weak magnetic controller being configured to calculate a D-axis current instruction initial value:
wherein id0Is the initial value of the D-axis current command, KiIs the integral coefficient of the controller and is, V1is the output voltage amplitude, v, of the DC-AC conversion circuit (6)dIs D-axis voltage, vqIs the Q-axis voltage, VmaxIs the maximum output voltage, V, of the DC-AC converter circuit (6)dcIs the dc bus voltage of the motor.
8. The capacitance-miniaturized motor driving device according to claim 7, wherein the clipping unit performs clipping processing on the D-axis current instruction initial value to obtain a D-axis current instruction:
wherein, IdemagFor the demagnetization current limit value of the motor, ImtpaControl of corresponding D-axis Current value, i, for MTPAq_ref1Is the total current command value of the Q axis, keIs the motor back electromotive force coefficient.
9. The capacitive miniaturized motor driving device according to claim 8, wherein said control section (2) further comprises a current clipping control module for achieving DQ output current limiting; the final DQ axis current command is calculated according to the following formula:
wherein imaxThe maximum current value allowed to be output by the direct current-alternating current conversion circuit (6).
10. The capacitance-type miniaturized motor driving device according to claim 9, wherein the control section (2) obtains a final DQ-axis current command id_refAnd iq_refAnd detecting and calculating the actual current i of the DQdAnd iqRespectively carrying out PI control on the D-axis current and the Q-axis current, and then adding decoupling and calculating to obtain a DQ axis voltage command VdAnd VqAnd αβ axis voltage command V is obtained by coordinate conversionαAnd VβThen converted into a u, V and w three-phase voltage command Vu、Vv、VwFinally, the sum V is calculatedu、Vv、VwEquivalent pulse Tu、Tv、TwAnd output to the motor through a direct current-to-alternating current conversion circuit (6).
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