CN111800037B - Electrolytic-capacitor-free control system, control method, compressor and refrigeration equipment - Google Patents

Electrolytic-capacitor-free control system, control method, compressor and refrigeration equipment Download PDF

Info

Publication number
CN111800037B
CN111800037B CN202010543088.7A CN202010543088A CN111800037B CN 111800037 B CN111800037 B CN 111800037B CN 202010543088 A CN202010543088 A CN 202010543088A CN 111800037 B CN111800037 B CN 111800037B
Authority
CN
China
Prior art keywords
power
voltage
current
axis
given
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Active
Application number
CN202010543088.7A
Other languages
Chinese (zh)
Other versions
CN111800037A (en
Inventor
韩一博
魏会军
徐常升
张东盛
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Gree Green Refrigeration Technology Center Co Ltd of Zhuhai
Original Assignee
Gree Green Refrigeration Technology Center Co Ltd of Zhuhai
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Gree Green Refrigeration Technology Center Co Ltd of Zhuhai filed Critical Gree Green Refrigeration Technology Center Co Ltd of Zhuhai
Priority to CN202010543088.7A priority Critical patent/CN111800037B/en
Publication of CN111800037A publication Critical patent/CN111800037A/en
Application granted granted Critical
Publication of CN111800037B publication Critical patent/CN111800037B/en
Active legal-status Critical Current
Anticipated expiration legal-status Critical

Links

Images

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/10—Arrangements for controlling torque ripple, e.g. providing reduced torque ripple
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/05—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for damping motor oscillations, e.g. for reducing hunting
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H02P25/024—Synchronous motors controlled by supply frequency
    • H02P25/026—Synchronous motors controlled by supply frequency thereby detecting the rotor position
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2205/00—Indexing scheme relating to controlling arrangements characterised by the control loops
    • H02P2205/01—Current loop, i.e. comparison of the motor current with a current reference
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2205/00—Indexing scheme relating to controlling arrangements characterised by the control loops
    • H02P2205/07—Speed loop, i.e. comparison of the motor speed with a speed reference

Abstract

The invention provides an electrolytic capacitor-free frequency conversion drive control system, a control method, a compressor and refrigeration equipment, relates to the technical field of frequency conversion drive, and adopts a three-loop closed-loop feedback control structure to realize the consistency of the power input power of a power supply, the power of an inverter and the power of a load motor of the electrolytic capacitor-free frequency conversion drive control system. By applying the invention, the power factor of a power grid of a control system can be improved by about 35.9%, the voltage of the direct-current side bus is improved from a periodic pulsating voltage signal to a relatively stable direct-current voltage signal, and the torque fluctuation phenomenon caused by the voltage pulsation of the direct-current side bus is inhibited to a certain extent.

Description

Electrolytic-capacitor-free control system, control method, compressor and refrigeration equipment
Technical Field
The invention relates to the technical field of variable frequency driving, in particular to an electrolytic-capacitor-free control system, a control method, a compressor and refrigeration equipment.
Background
The permanent magnet synchronous motor has the advantages of simple structure, reliable performance, high power density and the like, and is often applied to compressors of refrigeration equipment such as air conditioners, refrigerators and the like. A bus capacitor with high capacitance value and high withstand voltage value is connected in parallel at a direct current side bus end of a vector control system of the permanent magnet synchronous motor, and the vector control system has the functions of stabilizing the direct current side bus voltage and reducing voltage ripples and can provide the stable direct current side bus voltage for a load motor. Generally, the bus capacitor is made of an electrolytic capacitor, but the electrolytic capacitor has a large volume and a high price and is easily damaged in a working environment with a large temperature difference. Therefore, in order to improve the performance of the control system and reduce the production cost of the controller, the electrolytic capacitor with large capacitance value can be replaced by the thin film capacitor with smaller capacitance value, the service life of the replaced bus capacitor is prolonged to some extent, and the overall cost of the control system is saved. However, after the bus capacitor is replaced by the thin film capacitor from the electrolytic capacitor, the capacitance value of the bus capacitor is reduced, the capacity of storing electric energy is reduced, and a large-amplitude pulsation phenomenon is generated on the voltage of the bus on the direct current side, and the pulsation effect of the voltage of the direct current bus can cause the torque of the motor to generate the fluctuation of twice the frequency of a power grid, so that the rotating speed of the motor cannot be kept stable; and the harmonic content of the input current at the power grid side is increased, the distortion degree is increased, and the power grid power factor of the control system is reduced.
In the prior art, a power factor correction circuit (PFC circuit) is often used to solve the problem of unstable dc-side bus voltage of a control system. The PFC circuit can generate a corresponding sinusoidal current signal according to a required current parameter so as to stabilize the voltage of a direct-current side bus, but an energy storage element is additionally added in the PFC circuit, so that the current in the circuit generates a phase delay phenomenon, and the power factor of the circuit is reduced; meanwhile, the power loss of the circuit is additionally increased by a fully-controlled switch element in the PFC circuit, and for a refrigerator control system with lower operating power, the active power of the system promoted by the PFC circuit is not obviously higher than the additionally increased system power, so that the improvement effect of the PFC circuit applied to the refrigerator control system is not obvious.
Based on the above reasons, it is necessary to design a control method for optimizing and improving a thin film bus capacitor motor control system, which can suppress torque and rotation speed ripple phenomena caused by dc side bus voltage ripple, improve the power factor of the power grid of the control system, and drive a motor to operate stably, on the premise of ensuring correct operation of the control system and not additionally increasing circuit elements and circuit power loss.
Disclosure of Invention
In view of the above, the present invention provides a control system for an electrolytic capacitor-less variable frequency drive unit and a method thereof, which can achieve the technical effects of improving the power factor of the control system and/or improving the stability of the operation of the drive motor through the design of the control system and the control method.
In order to achieve the above object, the invention adopts the following technical scheme:
a control system for an electrolytic capacitor-less variable frequency drive unit, wherein,
the electrolytic capacitor-free variable frequency driving unit comprises the following components in sequential connection: the system comprises a network side input power circuit, a rectifying circuit, a thin film capacitor circuit, an inverter and a permanent magnet synchronous motor driven by the inverter; wherein, the film capacitor in the film capacitor circuit is used as a bus capacitor, and the voltage U on the film capacitordcThe capacitance value is the voltage of a direct-current side bus, and C is the capacitance value of a bus capacitor;
the control system comprises the following components which are connected in sequence: the system comprises a rotating speed control unit, a power control unit, a current control unit, a Park inverse transformation module and a space vector pulse width modulation module, wherein the space vector pulse width modulation module is used for controlling the inverter; wherein the content of the first and second substances,
the rotating speed control unit is used for: outputting an inverter given output power P to the power control unit based on feedback controlinv *；
The power control unit is configured to: based on feedback control, a d-axis given current I of the motor in a rotating rectangular coordinate system is output to the current control unitd *And q-axis set current Iq *；
The current control unit is used for: based on feedback control, outputting d-axis given voltage U of the motor under the rotating rectangular coordinate system to the Park inverse transformation moduled *And q-axis given voltage Uq *；
The Park inverse transformation module is used for outputting alpha axis given voltage U of the motor under the static rectangular coordinate system to the space vector pulse width modulation moduleα *And beta axis given voltage Uβ *；
The space vector pulse width modulation module is used for modulating the space vector pulse width according to the DC side bus voltage UdcAlpha axis given voltage Uα *And beta axis given voltage Uβ *Controlling the inverter to drive the permanent magnet synchronous motor;
and the number of the first and second electrodes,
the rotating speed control unit comprises a rotating speed error module, a rotating speed PI module and a given input power calculation module, wherein,
the rotation speed error module is used for setting the angular frequency omega according to the rotor of the permanent magnet synchronous motor*Rotor real-time angular frequency omega of permanent magnet synchronous motorrObtaining a rotation speed error;
the rotating speed PI module is used for carrying out PI adjustment on the rotating speed error to obtain a given input power instantaneous value P of the motor*；
The given input power calculation module calculates given output power P of the inverter by using at least the following formulainv *And output to the power control unit to realize power feedback control:
Figure GDA0003196324800000031
Figure GDA0003196324800000032
Figure GDA0003196324800000033
wherein, Pin *Giving an input power signal, P, to the network-side power supplyCIs the power of the bus capacitor, omega is the angular frequency of the input voltage of the network side power supply, thetagIs the phase angle of the mains-side supply input voltage.
Preferably, the first and second liquid crystal materials are,
the rotating speed control unit also comprises a rotating speed pulsation power calculation module, and,
the rotating speed pulsation power calculation module is used for generating a rotating speed pulsation compensation signal through the proportional resonance controller so as to inhibit the rotating speed fluctuation phenomenon generated by the motor rotor, wherein the resonance frequency is the DC side bus voltage UdcThe fluctuation frequency of (a);
the given input power calculation module calculates given output power P of the inverter by using at least the following formulainv *And output to the power control unit:
Figure GDA0003196324800000034
Figure GDA0003196324800000035
wherein, PLCompensating power, omega, for fluctuations in rotational speedr *Speed ripple compensation signal, T, for proportional resonant controller outputLAnd loading the torque for the motor in real time.
Preferably, the first and second liquid crystal materials are,
when phase compensation is not required, Pin *＝P*。
Preferably, the first and second liquid crystal materials are,
the given input power calculation module includes: a phase tracking-signal generator module for generating a phase tracking signal according to a network-side voltage U in the network-side power supply circuitinNet side current IinObtaining the frequency and the phase of a power signal at the input end of the network side power supply;
and is used for generating a sine wave signal with the same frequency and phase as the power signal of the power input end of the network side according to the frequency and phase of the power signal of the power input end, thereby calculating P through the following formulain *：
Figure GDA0003196324800000041
Preferably, the first and second liquid crystal materials are,
the phase tracking-signal generator module comprises: the device comprises a current and voltage detection module, a phase compensator based on a phase-locked loop function and a signal generator; wherein the content of the first and second substances,
the current and voltage detection module is used for detecting the voltage and the amplitude, the frequency and the phase of the current at the power input end and inputting the voltage and the phase into the signal generator;
the signal generator is used for multiplying the input end voltage signal and the input end current signal according to the multiplier included in the signal generator to obtain the amplitude, the frequency and the phase of the power signal at the power input end of the power supply;
and the phase compensator is used for generating a sine wave signal with the same frequency and phase as the power signal of the power input end according to the obtained frequency and phase of the power signal of the power input end.
Preferably, the first and second liquid crystal materials are,
the rotation speed pulsation power calculation module comprises: a proportional resonance control module and a compensation power signal calculation module, wherein,
a proportional resonance control module for setting a resonance frequency, and for,
outputting a rotational speed pulsation compensation signal omega according to the resonance frequency and the cut-off frequencyr；
A compensation power signal calculation module for calculating the compensation power signal according to the rotational speed ripple compensation signal omegarAnd motor real-time load torque TLCalculating the power P for compensating the fluctuation of the rotation speedL。
Preferably, the first and second liquid crystal materials are,
feedback control of the power control unit is established at a given output power P of the inverterinv *Load power P running in real time with motorloadIs above the error of (a), wherein,
Figure GDA0003196324800000051
wherein id、iqD-axis and q-axis real-time currents respectively;
and the current control unit is used for controlling the current according to the power error delta PinvPI regulation is carried out to obtain a given q-axis current i of the motorqA first step of; at the same time, use idObtaining a given d-axis current i of the motor in a control mode of 0d*。
Preferably, the first and second liquid crystal materials are,
the feedback control of the current control unit is established as follows: given d-axis current i of the machinedReal-time current i of x and d axesdD-axis current error Δ I obtained by difference calculationdAbove, and a given q-axis current i of said machineqReal-time current i of x and q axesqQ-axis current error Δ I obtained by difference calculationqAbove;
the current control unit controls the current according to the d-axis current error Delta IdQ-axis current error Δ IqPI regulation is respectively carried out to obtain d-axis given voltage U of the motord *And q-axis given voltage Uq *。
In addition, the invention also discloses a control method for the variable-frequency drive unit without the electrolytic capacitor, which comprises the following steps:
s100, collecting the voltage U of a direct current side bus at two ends of a thin film capacitor in a variable frequency drive unit without an electrolytic capacitordcAcquiring the real-time rotating speed of the rotor of the permanent magnet synchronous motor driven by the electrolytic-capacitor-free variable-frequency driving unit to obtain the real-time angular frequency omegar；
S200, obtaining given output power P of an inverter in the electrolytic-capacitor-free variable-frequency drive unit based on feedback controlinv *；
S300, obtaining d-axis given current I of the motor under a rotating rectangular coordinate system based on feedback controld *And q-axis set current Iq *；
S400, obtaining d-axis given voltage U of the motor under the rotating rectangular coordinate system based on feedback controld *And q-axis given voltage Uq *；
S500, setting a voltage U according to a d axis of the motord *And q-axis given voltage Uq *Further obtaining the alpha axis given voltage U of the motor under a static rectangular coordinate system through coordinate transformationα *And beta axis given voltage Uβ *；
S600, according to the direct-current side bus voltage UdcAlpha axis given voltage Uα *And beta axis given voltage Uβ *Driving the permanent magnet synchronous motor through an inverter in the electrolytic capacitor-free variable frequency driving unit;
wherein the content of the first and second substances,
step S200 includes the steps of:
s201, setting angular frequency omega according to rotor of permanent magnet synchronous motor*Rotor real-time angular frequency omega of permanent magnet synchronous motorrObtaining a rotation speed error;
s202, carrying out PI adjustment on the rotation speed error to obtain a given input power instantaneous value P of the motor*；
S203, calculating to obtain the given output power P of the inverter by using at least the following formulainv *To implement power feedback control:
Figure GDA0003196324800000061
Figure GDA0003196324800000062
Figure GDA0003196324800000063
wherein, Pin *Giving an input power signal, P, to the network-side power supplyCIs the power of the bus capacitor, omega is the angular frequency of the input voltage of the network side power supply, thetagThe phase angle of the input voltage of the network side power supply is shown, and C is the capacitance value of the bus capacitor.
Preferably, the first and second liquid crystal materials are,
the following steps are also included between steps S202 and S203:
s2021, generating a rotation speed pulsation compensation signal through a proportional resonance controller to suppress rotation speed fluctuation generated by a motor rotor, wherein the resonance frequency is a direct current side bus voltage UdcThe fluctuation frequency of (a);
s2023, calculating to obtain given output power P of the inverter by using at least the following formulainv *To replace the inverter given output power P in step S203inv *The calculation method of (2):
Figure GDA0003196324800000064
Figure GDA0003196324800000065
wherein, PLCompensating power, omega, for fluctuations in rotational speedr *Speed ripple compensation signal, T, for proportional resonant controller outputLAnd loading the torque for the motor in real time.
The invention also provides a compressor which uses the control method or the control system.
The invention also provides refrigeration equipment which is provided with the compressor.
Has the advantages that:
the invention provides a control method of variable frequency drive without electrolytic capacitor, which adopts a three-loop closed-loop feedback control structure to realize the control method for keeping the power input power of a power supply, the power of an inverter and the power of a load motor of a variable frequency drive control system without electrolytic capacitor consistent, and the control method is applied to improve the power factor of a power grid of the control system by about 35.9 percent and solve the problem of current phase compensation at the input side of the power supply.
In addition, the invention can further improve the DC side bus voltage from a periodic pulsating voltage signal to a relatively stable DC voltage signal, and simultaneously, the torque fluctuation phenomenon caused by the DC side bus voltage pulsation is also inhibited.
In conclusion, the invention can comprehensively solve the problem that the motor torque pulsates at 2 times of the power supply voltage frequency due to the unstable direct-current side bus voltage of the electrolytic capacitor-free variable frequency drive control system; the problem that the power of the power grid side is not matched with the load power of the motor when the variable-frequency drive control system without the electrolytic capacitor operates is solved, and the power factor of the power grid of the control system is improved; the problem of the motor rotational speed that no electrolytic capacitor variable frequency drive control system drive motor produced undulant by a wide margin under well low frequency operating mode is solved.
Drawings
The above and other objects, features and advantages of the present disclosure will become more apparent by describing in detail exemplary embodiments thereof with reference to the attached drawings. The drawings described below are merely exemplary embodiments of the present disclosure, and other drawings may be derived by those skilled in the art without inventive effort.
FIG. 1 is a topological diagram of a variable frequency drive unit without electrolytic capacitors;
FIG. 2 is a schematic structural diagram of a control system of a variable frequency drive unit without an electrolytic capacitor according to an embodiment of the present invention;
FIG. 3 is a schematic structural diagram of a rotational speed control unit and a power control unit of a control system without an electrolytic capacitor frequency conversion driving unit according to an embodiment of the present invention;
FIG. 4 is a flow chart illustrating a method for controlling a variable frequency drive unit without an electrolytic capacitor according to an embodiment of the present invention;
FIG. 5 is a graph of input voltage and input current waveforms for a control system employing a conventional control strategy;
FIG. 6 is a graph of a DC side bus voltage waveform for a control system employing a conventional control strategy;
FIG. 7 is a waveform of motor speed for a control system employing a conventional control strategy;
FIG. 8 is an enlarged view of a motor speed waveform for a control system employing a conventional control strategy;
FIG. 9 is a waveform diagram of input terminal voltage and input terminal current of the variable frequency drive control system without electrolytic capacitor according to the present invention;
FIG. 10 is a voltage waveform diagram of a DC-side bus of the electrolytic capacitor-less variable frequency drive control system of the present invention;
FIG. 11 is a waveform of the motor rotation speed of the variable frequency drive control system without electrolytic capacitor according to the present invention;
FIG. 12 is an enlarged view of the motor rotation speed waveform of the electrolytic capacitor-free variable frequency drive control system of the present invention.
Detailed Description
In order to make the technical solutions of the present invention better understood, the technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the drawings in the embodiments of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention.
In the prior art, a vector control system of an electrolytic capacitor-free variable frequency drive control system is generally composed of a power supply, a rectifier circuit, an inverter, a filter and the like. The rectifying circuit converts an input ac voltage into a dc voltage and transmits the dc voltage to the inverter, and the converted dc voltage is also referred to as a bus voltage. The inverter generates three-phase alternating current driving voltage by utilizing the bus voltage by means of pulse width modulation signals generated by the control system to drive the motor to run.
Firstly, a circuit topological diagram of the variable-frequency drive control system without electrolytic capacitors is briefly described. The variable-frequency drive control system without the electrolytic capacitor has a topological structure as shown in figure 1, and comprises a rectifying circuit, a power circuit, a thin-film capacitor, an inverter and a permanent magnet motor. Wherein the content of the first and second substances,
the power circuit is expressed as an alternating current power supply in the whole system and is used for supplying power to the whole system so as to ensure the normal operation of each device, and L is an inductor which is connected in series between the anode of the alternating current power supply and the input end in the rectifying circuit and is mainly used for inhibiting the impact of current at the moment of electrifying and harmonic interference in the operation process.
The rectification circuit is a unidirectional uncontrollable rectifier, and the specific interior of the rectification circuit mainly comprises four diodes, and as the unidirectional uncontrollable rectifier is mature, the use mode is not described herein, and a person skilled in the art can select a unidirectional uncontrollable rectifier with proper performance according to needs, and the use mode is not limited herein.
And the thin film capacitor positioned on the bus side is connected to two ends of the output end of the rectifying circuit and is used for absorbing higher harmonic waves of voltage and providing energy for maintaining normal operation of the motor under the condition of high-power operation. The capacitance of the bus capacitor-film capacitor is 20(uF), which is greatly reduced compared with the conventional DC bus electrolytic capacitance 220 (uF).
The inverter in the implementation of the present invention may be a three-phase voltage source inverter, configured to receive the voltage pulse of the pulse width modulation module, and control the motor according to the voltage pulse, because the three-phase voltage source inverter is mature, details are not described here regarding the usage mode thereof, and a person skilled in the art may select a single three-phase voltage source inverter with appropriate performance as needed, which is not limited here.
In one embodiment, the present invention discloses a control system for an electrolytic capacitor-less variable frequency drive unit, wherein,
the electrolytic capacitor-free variable frequency driving unit comprises the following components in sequential connection: the system comprises a network side input power circuit, a rectifying circuit, a thin film capacitor circuit, an inverter and a permanent magnet synchronous motor driven by the inverter; wherein, the film capacitor in the film capacitor circuit is used as a bus capacitor, and the voltage U on the film capacitordcThe capacitance value is the voltage of a direct-current side bus, and C is the capacitance value of a bus capacitor;
the control system comprises the following components which are connected in sequence: the system comprises a rotating speed control unit, a power control unit, a current control unit, a Park inverse transformation module and a space vector pulse width modulation module, wherein the space vector pulse width modulation module is used for controlling an inverter; wherein the content of the first and second substances,
a rotational speed control unit for: outputting the inverter given output power P to the power control unit based on the feedback controlinv *；
A power control unit to: based on feedback control, a d-axis given current I of the motor under a rotating rectangular coordinate system is output to the current control unitd *And q-axis set current Iq *；
A current control unit for: based on feedback control, d-axis given voltage U of the motor under the rotating rectangular coordinate system is output to the Park inverse transformation moduled *And q-axis given voltage Uq *；
A Park inverse transformation module for outputting the given voltage U of the alpha axis of the motor under the static rectangular coordinate system to the space vector pulse width modulation moduleα *And beta axis given voltage Uβ *；
A space vector pulse width modulation module for modulating the pulse width according to the DC side bus voltage UdcAlpha axis given voltage Uα *And beta axis given voltage Uβ *Controlling the inverter to drive the permanent magnet synchronous motor;
and the number of the first and second electrodes,
the rotating speed control unit comprises a rotating speed error module, a rotating speed PI module and a given input power calculation module, wherein,
a rotation speed error module for setting angular frequency omega according to the rotor of the PMSM*Rotor real-time angular frequency omega of permanent magnet synchronous motorrObtaining a rotation speed error;
a rotation speed PI module for PI regulating the rotation speed error to obtain the given input power instantaneous value P of the motor*；
The given input power calculation module calculates given output power P of the inverter by using at least the following formulainv *And output to the power control unit to realize power feedback control:
Figure GDA0003196324800000101
Figure GDA0003196324800000102
Figure GDA0003196324800000103
wherein, Pin *Giving an input power signal, P, to the network-side power supplyCIs the power of the bus capacitor, omega is the angular frequency of the input voltage of the network side power supply, thetagIs the phase angle of the mains-side supply input voltage.
With this embodiment, a three-loop, feedback-controlled system is proposed, and it can be found that by the above Pin *The correlation coefficient of the power phase compensation introduced by the calculation formula of (2) enables the embodiment to solve the problem of the power phase compensation through a three-loop feedback control mode. The power phase compensation solves the problem that the power factor of a power grid system is low in the energy storage and energy release process under the condition that the phases of the power grid voltage and the power grid current are asynchronous. How to solve the problem of the rotation speed pulsation and the like will be further described later by other embodiments.
The present embodiment also specifically describes a method of the PI regulator, and the output value of the PI regulator can be changed to convert the current into power by the following method:
the time domain function expression of the PI regulator is as follows:
Figure GDA0003196324800000111
where u (t) is the regulator output variable, e (t) is the regulator input variable, KpIs a proportionality coefficient, KiIs an integral coefficient. The transfer function of a PI regulator can also be expressed in the form of an s-domain transfer function:
G(s)＝Ks(τss+1)τss
where G(s) is the regulator output variable, s is the regulator input variable, KsAs a gain factor, τsIs the system time constant.
As is well known, the output of the PI regulator is an equivalent control quantity, not true current or power, and the two functional expressions are expressions of the PI regulator in different dimensions. If it is desired to set the output variable q-axis of the rotating speed PI regulator in the traditional control system to a given current iqConverting the system parameter into given power P, and calculating the system parameter K of the PI regulator to be changed according to the mathematical relation between the rotating speed of the permanent magnet synchronous motor and the q-axis current and the mathematical relation between the rotating speed and the powerpAnd KiOr system parameter K to be changed in the transfer function of the s-domain PI regulatorsAnd τsModifying PI regulation by the calculated parametersThe system parameters (both time domain and s domain) of the node can realize the conversion of the output variable of the rotating speed PI regulator from q-axis given current to given power.
In another embodiment of the present invention, the substrate is,
the rotation speed control unit further comprises a rotation speed pulsation power calculation module, and,
a rotation speed pulsation power calculation module for generating a rotation speed pulsation compensation signal through a proportional resonance controller to suppress the rotation speed fluctuation phenomenon generated by the motor rotor, wherein the resonance frequency is the DC side bus voltage UdcThe fluctuation frequency of (a);
the DC side bus voltage UdcThe fluctuation frequency of (a) is determined to be constant in a certain control system, and is generally obtained by direct calculation. The inventor finds that: given a frequency f of the AC supply voltagedc(usually 50Hz), in the electrolytic capacitor-free control system, after it is processed into a dc voltage signal by the rectifying circuit, the dc voltage is a "wave-shaped" sine or sine-like signal because the dc-side bus capacitor cannot stabilize the rectified dc voltage. The 'wave-shaped' DC voltage signal is obtained by inverting the negative half-axle voltage part of the AC voltage signal to the positive half-axle by the rectifying circuit, so that the frequency of the 'wave-shaped' DC voltage signal is f of the AC power supply voltage frequencydc2 times of, i.e. UdcThe fluctuation frequency of (2). This 2 times fdcThe voltage signal fluctuation phenomenon of the frequency is also a main cause of the rotation speed fluctuation of the electrolytic capacitor-free system.
The given input power calculation module calculates given output power P of the inverter by using at least the following formulainv *And output to the power control unit:
Figure GDA0003196324800000121
Figure GDA0003196324800000122
wherein, PLCompensating power, omega, for fluctuations in rotational speedr *Speed ripple compensation signal, T, for proportional resonant controller outputLAnd loading the torque for the motor in real time.
In another embodiment of the present invention, the substrate is,
when phase compensation is not required, Pin *＝P*。
In another embodiment of the present invention, the substrate is,
the given input power calculation module includes: a phase tracking-signal generator module for generating a phase tracking signal according to a network-side voltage U in the network-side power supply circuitinNet side current IinObtaining the frequency and the phase of a power signal at the input end of the network side power supply;
and for generating a sine wave signal having the same frequency and phase as the power signal at the power input terminal of the network side according to the frequency and phase of the power signal at the power input terminal of the power supply, thereby calculating P by the following formulain *：
Figure GDA0003196324800000123
In another embodiment of the present invention, the substrate is,
the phase tracking-signal generator module comprises: the device comprises a current and voltage detection module, a phase compensator based on a phase-locked loop function and a signal generator; wherein the content of the first and second substances,
the current and voltage detection module is used for detecting the amplitude, frequency and phase of the voltage and current at the input end of the power supply and inputting the amplitude, frequency and phase into the signal generator;
the signal generator is used for multiplying the input end voltage signal and the input end current signal according to the multiplier included in the signal generator to obtain the amplitude, the frequency and the phase of the power signal at the power input end of the power supply;
and the phase compensator is used for generating a sine wave signal with the same frequency and phase as the power signal of the power input end according to the obtained frequency and phase of the power signal of the power input end.
In another embodiment of the present invention, the substrate is,
the rotating speed pulsation power calculation module comprises: a proportional resonance control module and a compensation power signal calculation module, wherein,
a proportional resonance control module for setting a resonance frequency, and for,
outputting a rotational speed pulsation compensation signal omega according to the resonance frequency and the cut-off frequencyr；
A compensation power signal calculation module for calculating a compensation power signal omega according to the rotational speed ripplerAnd motor real-time load torque TLCalculating the power P for compensating the fluctuation of the rotation speedL。
In another embodiment of the present invention, the substrate is,
feedback control of the power control unit, based on a given output power P of the inverterinv *Load power P running in real time with motorloadIs above the error of (a), wherein,
Figure GDA0003196324800000131
wherein id、iqD-axis and q-axis real-time currents respectively;
the current control unit controls the current according to the power error Δ PinvPI regulation is carried out to obtain a given q-axis current i of the motorqA first step of; at the same time, use idObtaining a given d-axis current i of the motor in a control mode of 0d*。
In another embodiment of the present invention, the substrate is,
the feedback control of the current control unit is established by: given d-axis current i of the motordReal-time current i of x and d axesdD-axis current error Δ I obtained by difference calculationdAbove, and a given q-axis current i of the motorqReal-time current i of x and q axesqQ-axis current error Δ I obtained by difference calculationqAbove;
the current control unit controls the current according to the d-axis current error Delta IdQ-axis current error Δ IqPI regulation is respectively carried out to obtain d-axis given voltage U of the motord *And q-axis given voltage Uq *。
With respect to the above-described related embodiments, it can be seen that fig. 3 illustrates the feedback control principle of the related respective control units as a whole.
In one embodiment, as shown in fig. 4, the present embodiment proposes a control method of variable frequency driving without electrolytic capacitor to solve the problem of power phase compensation through three-loop negative feedback. And, the problem that the rotating speed of the motor cannot be kept stable and the power factor of the power grid of the control system will be reduced due to the reduction of the bus capacitance value is further solved through other embodiments. See below for details.
In another embodiment, as shown in fig. 4, the present invention discloses a control method for an electrolytic capacitor-less variable frequency drive unit, comprising the steps of:
s100, collecting the voltage U of a direct current side bus at two ends of a thin film capacitor in a variable frequency drive unit without an electrolytic capacitordcAcquiring the real-time rotating speed of the rotor of the permanent magnet synchronous motor driven by the electrolytic-capacitor-free variable-frequency driving unit to obtain the real-time angular frequency omegar；
S200, obtaining given output power P of an inverter in the electrolytic-capacitor-free variable-frequency drive unit based on feedback controlinv *；
S300, obtaining d-axis given current I of the motor under a rotating rectangular coordinate system based on feedback controld *And q-axis set current Iq *；
S400, obtaining d-axis given voltage U of the motor under the rotating rectangular coordinate system based on feedback controld *And q-axis given voltage Uq *；
S500, setting a voltage U according to a d axis of the motord *And q-axis given voltage Uq *Further obtaining the alpha axis given voltage U of the motor under a static rectangular coordinate system through coordinate transformationα *And beta axis given voltage Uβ *；
S600, according to the DC side bus voltage UdcAlpha axis given voltage Uα *And beta axis given voltage Uβ *By means of electroless platingAn inverter in the capacitance variable frequency driving unit drives a permanent magnet synchronous motor;
wherein the content of the first and second substances,
step S200 includes the steps of:
s201, setting angular frequency omega according to rotor of permanent magnet synchronous motor*Rotor real-time angular frequency omega of permanent magnet synchronous motorrObtaining a rotation speed error;
s202, carrying out PI adjustment on the rotation speed error to obtain a given input power instantaneous value P of the motor*；
S203, calculating to obtain the given output power P of the inverter by using at least the following formulainv *To implement power feedback control:
Figure GDA0003196324800000141
Figure GDA0003196324800000142
Figure GDA0003196324800000151
wherein, Pin *Giving an input power signal, P, to the network-side power supplyCIs the power of the bus capacitor, omega is the angular frequency of the input voltage of the network side power supply, thetagThe phase angle of the input voltage of the network side power supply is shown, and C is the capacitance value of the bus capacitor.
Preferably, the first and second liquid crystal materials are,
the following steps are also included between steps S202 and S203:
s2021, generating a rotation speed pulsation compensation signal through a proportional resonance controller to suppress rotation speed fluctuation generated by a motor rotor, wherein the resonance frequency is a direct current side bus voltage UdcThe fluctuation frequency of (a);
s2023, calculating to obtain given output power P of the inverter by using at least the following formulainv *To replace the inverter given output power in step S203Pinv *The calculation method of (2):
Figure GDA0003196324800000152
Figure GDA0003196324800000153
wherein, PLCompensating power, omega, for fluctuations in rotational speedr *Speed ripple compensation signal, T, for proportional resonant controller outputLAnd loading the torque for the motor in real time.
In another embodiment, the present invention further provides a control method for an electrolytic-capacitor-free variable frequency driving unit, which is implemented as follows:
s1: circuit parameters required by the control system are collected, and the method specifically comprises the following steps: a. to AC side input voltage UinAnd an AC side input current IinSampling is carried out; b. to DC side bus voltage UdcSampling is carried out; c. for any two-phase current i in three-phase current of motoraAnd ibSampling is carried out, and the third phase current can be calculated through the collected two-phase currents; d. real-time phase angle theta and real-time rotation speed omega for motor rotorrSampling is carried out, and particularly preferably, the real-time phase angle theta and the real-time rotating speed omega of the motor rotor are obtained by calculating the variation of the motor rotor angle in one electrical cycler；
S2: to the real-time phase current i obtained by samplingaAnd ibPerforming Clarke transformation to respectively obtain real-time alpha-axis current i under an alpha-beta-axis static rectangular coordinate systemαAnd real time beta axis current iβTo i, pairɑAnd iβCarrying out Park transformation to respectively obtain d-axis real-time current i under a rectangular rotating coordinate systemdAnd q-axis real-time current iq；
S3: calculating d-axis given current and q-axis given current;
for ease of understanding, the d-axis given current and the q-axis given current will be calculated in this implementationThe flow steps are roughly divided into four steps, first, the real-time rotation speed ω of the motor rotor is determinedrAnd comparing the error rotation speed with the given rotation speed omega to obtain an error rotation speed delta w, and performing PI regulation on the error rotation speed to obtain a given input power instantaneous value P of the motor.
It should be noted that the PI regulation mentioned in this embodiment can be specifically realized by the existing Proportional integral controller (PI controller, also called Proportional regulating and integral regulator, PI regulator), and its main functions are: and forming a control deviation according to the given value and the actual output value, and linearly combining the proportion and the integral of the deviation to form a control quantity to control the controlled object. That is, the error rotation speed is PI-regulated, so that the instantaneous value of the given input power of the motor can be obtained, unlike the conventional vector control system in which the output variable of the PI regulator is iq, in this embodiment, the rotation speed PI regulator is regulated, and the output variable is P, which calculates the input and output variables of the system and the motor through a variable-power that is uniform for each part. The first step in the power calculation is to obtain the power values of the various parts, here the instantaneous value P of the given input power.
Secondly, combining the instantaneous value P of the given input power of the motor with the frequency and phase parameters of the power input power obtained by sampling to obtain the given power input power PinA first step of; will give the power input power PinAs the subtracted number and the bus capacitance power PCSubtracting to obtain the given output power P of the inverterinv*。
And then comparing the real-time power of the motor load end with the given output power of the inverter to obtain the error power of the inverter.
And finally, carrying out PI regulation on the error power of the inverter to obtain d-axis given current and q-axis given current.
S4, comparing d-axis real-time current idAnd d-axis set current idObtaining d-axis error current, and comparing q-axis real-time current iqAnd q-axis set current iqAnd obtaining a q-axis error current.
S5 error of d axisPI regulation is carried out on the difference current to obtain d-axis given voltage udPerforming PI regulation on the q-axis error current to obtain a q-axis given voltage uq*。
S6, setting voltage u to d axis according to real-time rotating speeddGiven voltage u on x and q axesqCarrying out Park inverse transformation to obtain alpha axis given voltage uαGiven voltage u on x and beta axesβ*；
S7, setting voltage u according to alpha axisαGiven voltage u on the x and beta axesβSum bus voltage UdcAnd carrying out pulse width modulation on the inverter, and controlling the motor through the inverter.
In the control method provided by this embodiment, the input voltage at the ac side, the input current at the ac side, the bus voltage at the dc side, the phase current of the motor, and the rotor position are obtained by sampling the circuit of the control system, and then the given output power P of the inverter is obtained by calculationinvReal-time motor power PloadControlling an inverter through a power balance principle and three sets of PI regulators to generate a driving signal to drive a motor to operate; the control method adopts a three-loop closed-loop feedback method, realizes the control method for keeping the power input power, the inverter power and the load motor power of the electrolytic capacitor-free variable frequency drive control system consistent, improves the power factor of a power grid of the control system by about 35.9 percent by applying the control method, improves the voltage of a direct-current side bus from a periodic pulsating voltage signal to a relatively stable direct-current voltage signal, and simultaneously inhibits the torque fluctuation phenomenon caused by the voltage pulsation of the direct-current side bus.
In the prior art, a power factor correction circuit (PFC circuit) is often used to solve the problem of unstable dc-side bus voltage of a control system, wherein the PFC circuit has the following principle: the full-control type switching element is used for outputting pulse width modulation waveforms, the charging and discharging time of energy storage elements (capacitors, inductors and the like) in the circuit is controlled, and then a specific sine-like current signal is generated to stabilize the voltage of a direct-current side bus, and the power factor is improved. If the input voltage of the alternating current power supply is usinft and the input current is isinft, the input power of the alternating current power supply is uisin2 wt. Neglecting circuit losses, the rectified bus side power should be the same as the ac side mains input power. If the dc-side bus voltage is stabilized to the voltage value U, the bus current is kept at the sine signal of uisin2 wt/U. The PFC circuit may generate a corresponding sinusoidal current signal to stabilize the dc side bus voltage according to the desired current parameters. A
Taking the problem of bus voltage instability as an example, the invention obtains the given output power P of the inverterinvThe process of the steps provides the rotation speed fluctuation compensation power, and is also divided into two steps for facilitating understanding, and the real-time rotation speed omega of the motor rotor is firstly measuredrAdjusting to obtain the compensation power P of the fluctuation of the rotating speedL(ii) a Secondly, a given power input power PinSubtracting the bus capacitance power Pc as a subtracted number, and then adding the subtracted number to the rotation speed fluctuation compensation power PLAdding to obtain given output power P of the inverterinvThat is to say that the first and second electrodes,
Figure GDA0003196324800000171
it can be understood that the invention utilizes the proportional resonance adjustment to generate the rotation speed pulsation compensation signal for inhibiting the rotation speed fluctuation phenomenon generated by the motor rotor. The following is only used to educate the relevant formula of the invention in the front for understanding:
the method for calculating the compensation power of the rotation speed fluctuation comprises the following steps: firstly, a proportional resonance controller is utilized to obtain a rotating speed signal for inhibiting rotating speed pulsation; wherein, the resonance frequency of the proportional resonance controller is set as the DC side bus voltage UdcThe fluctuation frequency of (a);
the resonance controller transfer function of the proportional resonance controller is shown as formula (1),
Figure GDA0003196324800000181
wherein G isR(s) is a rotational speed signal for suppressing rotational speed pulsation, KRFor the resonance coefficient, for controlling the output gain, w, of the resonance controllercIs the proportional resonant controller cutoff frequency. Taking the frequency of the household alternating current as 50Hz, the direct current side bus voltage UdcIs 2 times the ac frequency, this voltage fluctuation frequency is namely 100Hz, so the resonance frequency of the proportional resonance controller can now be set to 2 times the network side input voltage frequency, typically 100 Hz. Theoretically, the proportional resonant controller has infinite gain at a resonant frequency point, and can generate a compensation signal for inhibiting the pulsation of a specific resonant frequency and reduce the rotation speed pulsation degree of a motor rotor.
Secondly, the output variable of the proportional resonance adjustment is a rotating speed signal for inhibiting the rotating speed pulsation, in order to keep the control variable in the control link consistent, the rotating speed signal needs to be converted into a power signal, according to a power calculation formula of the motor, the rotating speed and the mechanical torque can be multiplied, and the obtained result is the rotating speed pulsation compensation power of the motor. The calculation method is shown as formula (2)
Figure GDA0003196324800000182
Wherein, PLCompensating power for speed fluctuations, wr *For the pulsating speed suppression signal, TLAnd loading the torque for the motor in real time.
The proportional resonance adjustment is utilized to collect and control the pulsation component existing in the rotating speed of the motor, and the control signal is introduced into a power calculation link to inhibit the pulsation component in the rotating speed of the motor and inhibit the fluctuation phenomenon of the rotating speed of the motor.
The bus capacitance power is obtained by calculating parameters of a bus capacitance value and a direct-current side bus voltage, and the specific calculation method comprises the following steps:
Figure GDA0003196324800000183
wherein, PCFor bus capacitive power, UdcIs the DC side bus voltage, C is the bus capacitance, w is the power input voltage frequency, θgIs the phase angle of the power supply input voltage.
The given power input power is obtained by calculating parameters of input voltage and input current at an alternating current side, and the specific calculation method comprises the following steps:
assuming that the input voltage of the power supply is an ideal sinusoidal signal, the input current is an ideal sinusoidal signal and has the same phase and frequency as the input voltage, and the input power factor of the power supply is 1, the input power of the power supply is as shown in equation (4):
Pin＝UinIin＝UI sin2(wt+θg)----------------------------(4)
wherein, PinU, I is the power input power, grid voltage amplitude and current amplitude, respectively, w is the power input voltage frequency, thetagIs the phase angle of the power supply input voltage;
the sine waveform with the same frequency and phase of the power supply input power signal is used as a frequency coefficient and multiplied by the given input power instantaneous value P of the motor to obtain the given input power signal Pin*，PinThe expression of the formula is shown as the formula (5):
Figure GDA0003196324800000191
the specific calculation method for the d-axis given current and the q-axis given current is as follows: under ideal conditions, the inverter delivers a given output power PinvThe real-time running power of the motor is equal to that of the motor, and the actual running power of the motor can be calculated by the formula (6)
Figure GDA0003196324800000192
Wherein, PloadFor the real-time operating power of the motor, Ud*、uqGiven voltages, i, for d-q axes respectivelyd、iqD-q axis real-time currents respectively; setting the inverter to a given output power PinvComparing the real-time running power of the motor to obtain the error power delta P of the inverterinv(ii) a Will be delta PinvPI regulation is carried out to obtain a given q-axis current i of the motorqA first step of; at the same time, use idObtaining a given d-axis current i of the motor in a control mode of 0d*。
In S5, the d-axis given voltage and the q-axis given voltage are calculated by using a voltage balance equation of the motor in a rotating rectangular coordinate system, and the specific calculation method is as follows:
Figure GDA0003196324800000193
wherein u isd、uqFor d-q axis voltage, i, of the motord、iqFor d-q axis currents of the machine, Rd、RqIs d-q axis resistance, Ld、LqD-q axis inductance, w is the real-time rotor speed, i.e. the electrical angular velocity, #fIs a permanent magnet flux linkage; the sampled real-time current i of the motord、iqWith a given current id*、iqRespectively comparing to obtain d-q axis error current, respectively carrying out PI regulation to obtain given d-q axis voltage u of motor operationd*、uq*。
In another embodiment, as shown in fig. 2, the present invention comprehensively provides a control system for an electrolytic capacitor-free variable frequency drive unit, which comprises a rotation speed control unit, a power control unit and a current control unit, wherein the rotation speed control unit comprises a rotation speed error module, a rotation speed PI module, a given input power calculation module and a rotation speed ripple power calculation module, wherein an output variable of the rotation speed PI module is adjusted to a given power input power; a phase tracking-signal generator in the given input power calculation module generates specific frequency and phase coefficient to be multiplied by the input power of a given power supply to generate given direct-current side bus power with a pulsating state, the given input power calculation module further processes the given direct-current side bus power, and the given direct-current side bus power is calculated with the rotating speed pulsating compensation power output by the rotating speed pulsating power calculation module and the direct-current side bus capacitance power obtained by sampling to finally obtain given output power of the inverter;
the power control unit comprises a power error module, a power PI module and a motor load power calculation module, wherein the power error module compares the given output power of the inverter with the real-time power of a motor load end output by the motor load power calculation module to obtain a power difference input value power PI module, and the given motor q-axis current is obtained;
the current control unit comprises a d-axis current error module, a d-axis current PI module, a q-axis current error module and a q-axis current PI module, wherein the d-axis current error module compares d-axis given current with d-axis real-time current to obtain d-axis error current, and inputs the d-axis error current into the d-axis current error module to obtain d-axis given voltage required by the operation of the motor; the q-axis current error module compares the q-axis given current with the q-axis real-time current to obtain a q-axis error current, and the q-axis error current is input into the q-axis current error module to obtain a q-axis given voltage required by the motor to operate.
Preferably, the rotational speed pulsation power calculation module comprises a proportional resonance control module and a compensation power signal calculation module, wherein the resonance frequency of the proportional resonance control module is set to be a motor rotational speed fluctuation signal, namely, the direct-current side bus voltage UdcThe real-time rotating speed of the motor rotor is input into the proportional resonance control module to output a rotating speed fluctuation compensation power signal; the compensation power signal calculation module converts the compensation rotating speed signal into a compensation power signal and inputs the compensation power signal into the given input power calculation module to complete the subsequent control process. Further preferably, the main part of the proportional resonance control module is a resonance controller, and the resonance controller generates a rotation speed compensation signal corresponding to a rotation speed pulsation signal acting on a specific resonance frequency. Preferably, the compensation power calculating module converts the rotation speed compensation signal into the power compensation signal by multiplying the fluctuating rotation speed compensation signal output by the resonance controller by a given load torque when the refrigerator is operated according to a formula P ═ w × T, that is, a calculation method in which the power is equal to the angular velocity multiplied by the torque, to obtain the fluctuating power compensation signal.
Preferably, the phase tracking-signal generator module includes a current-voltage detection module, a phase compensator based on a phase-locked loop function, and a signal generator. The current and voltage detection module can detect the amplitude, frequency and phase of the input end voltage and the input end current and input the amplitude, frequency and phase into the signal generator, and a multiplier in the signal generator multiplies the input end voltage signal by the input end current signal to obtain the amplitude, frequency and phase of the power signal at the input end of the power supply; the phase compensator generates a sine wave signal with the same frequency and phase as the power signal at the power input end according to the obtained frequency and phase parameters of the power signal at the power input end, and the sine wave signal is used as the frequency and phase coefficient of the power signal at the given inverter end.
Preferably, the current and voltage detecting module is configured at the input end power supply side of the circuit, and is configured to detect the real-time current and the real-time voltage of the input end power supply respectively, and calculate the real-time power of the input end power supply.
Preferably, the real-time power of the input power supply is obtained by multiplying the real-time current and the real-time voltage of the input power supply, and the input power supply is an alternating current power supply, so that the input power is a sinusoidal signal.
Preferably, the signal generator is capable of calculating the maximum value of the input power by using the maximum value of the input voltage and the maximum value of the input current obtained by sampling, extracting the frequency and phase parameters of the input power signal by using the maximum value of the input power, and generating a basic sine wave signal with the frequency and phase parameters consistent with the frequency and phase parameters of the input power signal.
Preferably, the phase compensator applies a phase-locked loop strategy that adjusts the frequency and phase of the varying input power signal input to the phase compensator using the basic sine wave signal output by the signal generator as a reference signal such that the frequency and phase of the adjusted input power signal coincide with the frequency and phase of the reference signal.
FIG. 3 is a schematic diagram of a rotation speed control unit and a power control unit of an electrolytic capacitor-free frequency conversion drive control system according to another embodiment, in which a main portion of a proportional resonance control module is a resonance controller, the resonance controller generates a rotation speed compensation signal corresponding to a specific resonance frequency by using a rotation speed ripple signal acting on the specific resonance frequency, a compensation power calculation module converts the rotation speed compensation signal into a rotation speed power compensation signal, and the power compensation signal is equal to the phase of an angular speed and a torqueThe calculating method of the multiplication multiplies the fluctuating rotating speed compensation signal output by the resonance controller by the given load torque when the refrigerator runs to obtain the rotating speed fluctuating power compensation signal. The composite power control module is used for controlling the output power of the given power supply and the power P of the given direct-current side businCapacitance power P of bus on direct current sideCActual operating power P of the motor loadloadCompensating power P for speed fluctuationLAnd processing to obtain the given d-q axis current of the motor in operation, and performing subsequent control steps by using the given d-q axis current to drive the permanent magnet synchronous motor to operate. The given power supply input power P is obtained by comparing a given rotating speed with the actual rotating speed of the motor to obtain a difference value through PI regulation; given DC side bus power Pin-derived by the co-action of a phase tracking-signal generator module with a given power supply input power P; the given d-q axis current composite power control module operated by the motor and the power PI regulator are obtained through the combined action; given inverter output power P in a composite power control moduleinvThe calculation method is as follows: p ═ Pin*-PC+PL(ii) a The given q-axis current of the motor operation is obtained by comparing the output power of the given inverter with the actual power of the motor load to obtain a difference value through PI regulation. Given d-axis current of motor operation is represented by idObtained under the control principle of 0.
According to the control strategy of the invention, the variable frequency drive control system without electrolytic capacitor is used for driving a certain refrigerator compressor and analyzing the control effect of the refrigerator compressor, and meanwhile, the traditional vector control system is used for driving the same refrigerator compressor and comparing the control effects of the two control systems. (hereinafter, the traditional vector control system is abbreviated as a traditional system, and the variable-frequency drive control system without electrolytic capacitor is abbreviated as an optimization system);
the capacitance value adopted by the direct-current side bus capacitor of the variable-frequency drive control system without the electrolytic capacitor is 20(uF), and is greatly reduced compared with the common direct-current bus electrolytic capacitance value 220 (uF). In the traditional vector control system, the bus capacitance is changed to 20(uF), the load torques of the two control systems are set to be completely the same, the given rotating speed is 2400 (revolutions per minute), the motor is started at the rated rotating speed and in the full-load state, and the control effects of the two control systems under the same load and the same rotating speed are respectively analyzed.
Fig. 5 is a diagram of input side current and voltage waveforms of a conventional system, and fig. 9 is a diagram of input side current and voltage waveforms of an optimized system. Comparing fig. 5 and fig. 9, it can be seen that the input side current distortion phenomenon in the conventional system is severe, and the input side current waveform of the optimized system has a high sine degree and is relatively synchronous with the input side voltage phase. Through further calculation, the power factor of the optimized system is improved by about 0.177 and the power factor improvement amplitude is about 35.9% compared with the traditional system under the condition of 2400 (revolutions per minute) rotating speed.
Fig. 6 is a voltage waveform diagram of a dc-side bus of a conventional system, and fig. 10 is a voltage waveform diagram of a dc-side bus of an optimized system. Comparing fig. 6 and 10, it can be found that the dc-side bus voltage of the optimized system is about 275V at the minimum value and about 330V at the maximum value; the DC bus voltage of the conventional system is about 83.3V at the minimum value and about 410V at the maximum value. Therefore, the voltage stability of the direct current side bus of the optimized system is obviously improved.
Fig. 7 and 8 are rotation speed waveform diagrams of a conventional system, and fig. 11 and 12 are rotation speed waveform diagrams of an optimized system. Comparing the amplified rotation speed waveform diagrams of fig. 7 and fig. 11, calculating and analyzing the parameters in the diagrams can obtain that the minimum value of the rotation speed is about 2361, the maximum value is about 2427, the fluctuation amount of the rotation speed is about 50 (revolutions per minute) and the average value of the rotation speed is about 2418 (revolutions per minute) when the motor is stably operated in the conventional system under the condition of the given rotation speed 2400 (revolutions per minute); under the same condition, the minimum value of the rotating speed of the motor in the stable running process is about 2375, the maximum value is about 2394, the fluctuation amount of the rotating speed is about 20 (revolutions per minute), and the average value of the rotating speed is about 2388 (revolutions per minute). The optimization system has a very obvious effect of inhibiting the motor rotating speed fluctuation phenomenon on the premise of applying a control strategy without electrolytic capacitor.
In conclusion, the control method for variable frequency driving without electrolytic capacitors provided by the invention can improve the power factor of a power grid of a control system and improve the running stability of a driving motor through the design of the control method. The control system using the control method provided by the implementation is additionally provided with a phase compensator, a resonance controller and other control units on the basis of a rotating speed PI regulator, and utilizes a rotating speed ripple compensation signal generated by the resonance controller to inhibit the rotating speed ripple phenomenon caused by the voltage fluctuation of a direct-current side bus of the electrolytic-capacitor-free variable-frequency drive control system; the PI controller, the resonance controller and the phase compensator are combined to form a composite control unit to improve the speed outer ring structure of the traditional vector control system, the control unit can inhibit the large-amplitude fluctuation phenomenon of the intermediate frequency and low frequency rotating speed of the motor, and the electrolytic capacitor-free variable frequency drive control system applying the control unit can reduce the low frequency rotating speed fluctuation in the motor from about +/-50 (revolutions per minute) to about +/-20 (revolutions per minute).
It should be noted that the terms "first," "second," and the like in the description and claims of the present invention and in the drawings described above are used for distinguishing between similar elements and not necessarily for describing a particular sequential or chronological order. It is to be understood that the data so used is interchangeable under appropriate circumstances such that the embodiments of the invention described herein are capable of operation in sequences other than those illustrated or described herein. Furthermore, the terms "comprises," "comprising," and "having," and any variations thereof, are intended to cover a non-exclusive inclusion, such that a process, method, system, article, or apparatus that comprises a list of steps or elements is not necessarily limited to those steps or elements expressly listed, but may include other steps or elements not expressly listed or inherent to such process, method, article, or apparatus.
The above-mentioned serial numbers of the embodiments of the present invention are merely for description and do not represent the merits of the embodiments.
In the above embodiments of the present invention, the descriptions of the respective embodiments have respective emphasis, and for parts that are not described in detail in a certain embodiment, reference may be made to related descriptions of other embodiments.
In the embodiments provided in the present application, it should be understood that the disclosed technology can be implemented in other ways. The above-described embodiments of the apparatus are merely illustrative, and for example, a division of a unit may be a division of a logic function, and an actual implementation may have another division, for example, a plurality of units or components may be combined or may be integrated into another system, or some features may be omitted, or may not be executed. In addition, the shown or discussed mutual coupling or direct coupling or communication connection may be an indirect coupling or communication connection through some interfaces, units or modules, and may be in an electrical or other form.
The units described as separate parts may or may not be physically separate, and parts displayed as units may or may not be physical units, may be located in one place, or may be distributed on a plurality of units. Some or all of the units can be selected according to actual needs to achieve the purpose of the solution of the embodiment.
In addition, functional units in the embodiments of the present invention may be integrated into one processing unit, or each unit may exist alone physically, or two or more units are integrated into one unit. The integrated unit can be realized in a form of hardware, and can also be realized in a form of a software functional unit.
The integrated unit, if implemented in the form of a software functional unit and sold or used as a stand-alone product, may be stored in a computer readable storage medium. Based on such understanding, the technical solution of the present invention may be embodied in the form of a software product, which is stored in a storage medium and includes instructions for causing a computer device (which may be a personal computer, a server, or a network device) to execute all or part of the steps of the method according to the embodiments of the present invention. And the aforementioned storage medium includes: a U-disk, a Read-Only Memory (ROM), a Random Access Memory (RAM), a removable hard disk, a magnetic or optical disk, and other various media capable of storing program codes.
The foregoing is only a preferred embodiment of the present invention, and it should be noted that it is obvious to those skilled in the art that various modifications and improvements can be made without departing from the principle of the present invention, and these modifications and improvements should also be considered as the protection scope of the present invention.

Claims (10)

1. A control system for an electrolytic capacitor-less variable frequency drive unit, characterized by: wherein the content of the first and second substances,
the electrolytic capacitor-free variable frequency driving unit comprises the following components in sequential connection: the system comprises a network side input power circuit, a rectifying circuit, a thin film capacitor circuit, an inverter and a permanent magnet synchronous motor driven by the inverter; wherein, the film capacitor in the film capacitor circuit is used as a bus capacitor, and the voltage U on the film capacitordcThe capacitance value is the voltage of a direct-current side bus, and C is the capacitance value of a bus capacitor;
the control system comprises the following components which are connected in sequence: the system comprises a rotating speed control unit, a power control unit, a current control unit, a Park inverse transformation module and a space vector pulse width modulation module, wherein the space vector pulse width modulation module is used for controlling the inverter; wherein the content of the first and second substances,
the rotating speed control unit is used for: outputting an inverter given output power P to the power control unit based on feedback controlinv *；
The power control unit is configured to: based on feedback control, a d-axis given current I of the motor in a rotating rectangular coordinate system is output to the current control unitd *And q-axis set current Iq *；
The current control unit is used for: based on feedback control, outputting d-axis given voltage U of the motor under the rotating rectangular coordinate system to the Park inverse transformation moduled *And q-axis given voltage Uq *；
The Park inverse transformation module is used for outputting alpha axis given voltage U of the motor under the static rectangular coordinate system to the space vector pulse width modulation moduleα *And beta axis given voltage Uβ *；
The space vector pulse width modulation module is used for modulating the space vector pulse width according to the DC side bus voltage UdcAlpha axis given voltage Uα *And beta axis given voltage Uβ *Controlling the inverter to drive the permanent magnet synchronous motor;
and the number of the first and second electrodes,
the rotating speed control unit comprises a rotating speed error module, a rotating speed PI module, a given input power calculation module and a rotating speed pulsation power calculation module, wherein,
the rotation speed error module is used for setting the angular frequency omega according to the rotor of the permanent magnet synchronous motor*Rotor real-time angular frequency omega of permanent magnet synchronous motorrObtaining a rotation speed error;
the rotating speed PI module is used for carrying out PI adjustment on the rotating speed error to obtain a given input power instantaneous value P of the motor*；
The rotating speed pulsation power calculation module is used for generating a rotating speed pulsation compensation signal through the proportional resonance controller so as to inhibit the rotating speed fluctuation phenomenon generated by the motor rotor, wherein the resonance frequency is the DC side bus voltage UdcThe fluctuation frequency of (a);
the given input power calculation module calculates given output power P of the inverter by using at least the following formulainv *And output to the power control unit to realize power feedback control:
Figure FDA0003206994470000021
Figure FDA0003206994470000022
Figure FDA0003206994470000023
Figure FDA0003206994470000024
wherein, Pin *Giving an input power signal, P, to the network-side power supplyCIs the power of the bus capacitor, omega is the angular frequency of the input voltage of the network side power supply, thetagThe phase angle of the input voltage of the network side power supply;
wherein, PLCompensating power, omega, for fluctuations in rotational speedr *Speed ripple compensation signal, T, for proportional resonant controller outputLAnd loading the torque for the motor in real time.
2. The control system of claim 1, wherein: wherein the content of the first and second substances,
when phase compensation is not required, Pin *＝P*。
3. The control system according to any one of claims 1 to 2,
the given input power calculation module includes: a phase tracking-signal generator module for generating a phase tracking signal according to a network-side voltage U in the network-side power supply circuitinNet side current IinObtaining the frequency and the phase of a power signal at the input end of the network side power supply;
and is used for generating a sine wave signal with the same frequency and phase as the power signal of the power input end of the network side according to the frequency and phase of the power signal of the power input end, thereby calculating P through the following formulain *：
Figure FDA0003206994470000025
4. The control system of claim 3, wherein: wherein the content of the first and second substances,
the phase tracking-signal generator module comprises: the device comprises a current and voltage detection module, a phase compensator based on a phase-locked loop function and a signal generator; wherein the content of the first and second substances,
the current and voltage detection module is used for detecting the voltage and the amplitude, the frequency and the phase of the current at the power input end and inputting the voltage and the phase into the signal generator;
the signal generator is used for multiplying the input end voltage signal and the input end current signal according to the multiplier included in the signal generator to obtain the amplitude, the frequency and the phase of the power signal at the power input end of the power supply;
and the phase compensator is used for generating a sine wave signal with the same frequency and phase as the power signal of the power input end according to the obtained frequency and phase of the power signal of the power input end.
5. The control system of claim 3, wherein: wherein the content of the first and second substances,
the rotation speed pulsation power calculation module comprises: a proportional resonance control module and a compensation power signal calculation module, wherein,
a proportional resonance control module for setting a resonance frequency, and for,
outputting a rotational speed pulsation compensation signal omega according to the resonance frequency and the cut-off frequency* r；
A compensation power signal calculation module for calculating the compensation power signal according to the rotational speed ripple compensation signal omega* rAnd motor real-time load torque TLCalculating the power P for compensating the fluctuation of the rotation speedL。
6. The control system according to any one of claims 1 to 2, wherein: wherein the content of the first and second substances,
feedback control of the power control unit is established at a given output power P of the inverterinv *Load power P running in real time with motorloadIs above the error of (a), wherein,
Figure FDA0003206994470000031
wherein id、iqD-axis and q-axis real-time currents respectively; u. ofd *Setting a voltage for the d-axis; u. ofq *Setting voltage for q axis;
and the current control unit is based on the power errorΔPinvPI regulation is carried out to obtain a given q-axis current i of the motorqA first step of; at the same time, use idObtaining a given d-axis current i of the motor in a control mode of 0d*。
7. The control system according to any one of claims 1 to 2, wherein: wherein the content of the first and second substances,
the feedback control of the current control unit is established as follows: given d-axis current i of the machinedReal-time current i of x and d axesdD-axis current error Δ I obtained by difference calculationdAbove, and a given q-axis current i of said machineqReal-time current i of x and q axesqQ-axis current error Δ I obtained by difference calculationqAbove;
the current control unit controls the current according to the d-axis current error Delta IdQ-axis current error Δ IqPI regulation is respectively carried out to obtain d-axis given voltage U of the motord *And q-axis given voltage Uq *。
8. A control method for an electrolytic capacitor-free variable frequency drive unit is characterized in that: the method comprises the following steps:
s100, collecting the voltage U of a direct current side bus at two ends of a thin film capacitor in a variable frequency drive unit without an electrolytic capacitordcAcquiring the real-time rotating speed of the rotor of the permanent magnet synchronous motor driven by the electrolytic-capacitor-free variable-frequency driving unit to obtain the real-time angular frequency omegar；
S200, obtaining given output power P of an inverter in the electrolytic-capacitor-free variable-frequency drive unit based on feedback controlinv *；
S300, obtaining d-axis given current I of the motor under a rotating rectangular coordinate system based on feedback controld *And q-axis set current Iq *；
S400, obtaining d-axis given voltage U of the motor under the rotating rectangular coordinate system based on feedback controld *And q-axis given voltage Uq *；
S500, according to d-axis supply of the motorConstant voltage Ud *And q-axis given voltage Uq *Further obtaining the alpha axis given voltage U of the motor under a static rectangular coordinate system through coordinate transformationα *And beta axis given voltage Uβ *；
S600, according to the direct-current side bus voltage UdcAlpha axis given voltage Uα *And beta axis given voltage Uβ *Driving the permanent magnet synchronous motor through an inverter in the electrolytic capacitor-free variable frequency driving unit;
wherein the content of the first and second substances,
step S200 includes the steps of:
s201, setting angular frequency omega according to rotor of permanent magnet synchronous motor*Rotor real-time angular frequency omega of permanent magnet synchronous motorrObtaining a rotation speed error;
s202, carrying out PI adjustment on the rotation speed error to obtain a given input power instantaneous value P of the motor*；
S203, calculating to obtain the given output power P of the inverter by using at least the following formulainv *To implement power feedback control:
Figure FDA0003206994470000041
Figure FDA0003206994470000042
Figure FDA0003206994470000043
Figure FDA0003206994470000051
wherein, Pin *Giving an input power signal, P, to the network-side power supplyCAs bus bar electricityCapacity, ω is the angular frequency of the input voltage of the network-side power supply, θgThe phase angle of the input voltage of the network side power supply is shown, and C is the capacitance value of the bus capacitor;
wherein, PLCompensating power, omega, for fluctuations in rotational speedr *Speed ripple compensation signal, T, for proportional resonant controller outputLAnd loading the torque for the motor in real time.
9. A compressor, characterized by: comprising a control system according to any one of claims 1-7 or using a control method according to claim 8.
10. A refrigeration apparatus, characterized by: comprising the compressor of claim 9.
CN202010543088.7A 2020-06-15 2020-06-15 Electrolytic-capacitor-free control system, control method, compressor and refrigeration equipment Active CN111800037B (en)

Priority Applications (1)

Application Number Priority Date Filing Date Title
CN202010543088.7A CN111800037B (en) 2020-06-15 2020-06-15 Electrolytic-capacitor-free control system, control method, compressor and refrigeration equipment

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
CN202010543088.7A CN111800037B (en) 2020-06-15 2020-06-15 Electrolytic-capacitor-free control system, control method, compressor and refrigeration equipment

Publications (2)

Publication Number Publication Date
CN111800037A CN111800037A (en) 2020-10-20
CN111800037B true CN111800037B (en) 2021-12-03

Family

ID=72804727

Family Applications (1)

Application Number Title Priority Date Filing Date
CN202010543088.7A Active CN111800037B (en) 2020-06-15 2020-06-15 Electrolytic-capacitor-free control system, control method, compressor and refrigeration equipment

Country Status (1)

Country Link
CN (1) CN111800037B (en)

Families Citing this family (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN112928962B (en) * 2021-02-05 2022-06-24 哈尔滨工业大学（深圳） Electrolytic capacitor-free variable frequency driving system control method, controller and storage medium
CN113364372B (en) * 2021-06-04 2023-05-12 深圳市振邦智能科技股份有限公司 Electroless control method for compressor
CN114337417B (en) * 2021-12-30 2023-10-31 海信空调有限公司 Motor control method, air conditioner and computer readable storage medium
CN117254745B (en) * 2023-11-17 2024-03-22 深圳市精锐昌科技有限公司 Operation control method, system and storage medium of motor

Family Cites Families (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP1480326A4 (en) * 2002-02-25 2016-10-12 Daikin Ind Ltd Motor control method and its apparatus
CN104617850A (en) * 2014-12-31 2015-05-13 广西科技大学 Double-closed-loop controller and double-closed-loop control method of permanent magnet synchronous motor
JP6735622B2 (en) * 2016-07-22 2020-08-05 日立ジョンソンコントロールズ空調株式会社 Motor drive device and refrigeration and air conditioning equipment using the same
CN106788115B (en) * 2017-01-24 2019-07-09 南京航空航天大学 Variable frequency drive control system and control method based on no electrolytic capacitor inverter
CN108809177B (en) * 2018-06-26 2021-08-03 Tcl空调器(中山)有限公司 Electrolytic capacitor-free motor driving method, electrolytic capacitor-free motor driving device, electronic apparatus, and storage medium

Also Published As

Publication number Publication date
CN111800037A (en) 2020-10-20

Similar Documents

Publication Publication Date Title
CN111800037B (en) Electrolytic-capacitor-free control system, control method, compressor and refrigeration equipment
KR101995864B1 (en) Inverter control apparatus and method thereof
CN105515354B (en) Method for controlling a compressor, device and system
CN108923721B (en) Motor variable frequency driving system and multi-split central air conditioner
CN104685778B (en) Method for controlling power conversion apparatus
WO2017198042A1 (en) Direct-current bus voltage fluctuation suppression method and control method for dual-pwm current converter
CN109546913B (en) Capacitor miniaturization motor driving device
CN106961232B (en) Control method of high-power-factor diode rectifier permanent magnet motor adopting small-capacity thin-film capacitor
CN113098365B (en) Method and system for suppressing network side current harmonic of motor driving system without electrolytic capacitor
CN110034690B (en) Vienna rectifier model prediction virtual flux linkage control method
JP6521131B1 (en) Power converter
CN110601623B (en) Method for reducing DC bus voltage ripple amplitude of permanent magnet synchronous motor frequency converter
CN107689761A (en) Damping control method in no electrolytic capacitor permanent magnet synchronous electric air conditioner drive system
CN112653342B (en) Complex vector current loop decoupling control device and method under static coordinate system
CN106208868A (en) No electrolytic capacitor motor driven systems and control method, device
CN113067483B (en) Bus capacitor miniaturization compressor driving control method
CN110545046A (en) Parallel PWM rectifier circulating current restraining method based on virtual vector
Bašić et al. Energy efficient control of a stand-alone wind energy conversion system with AC current harmonics compensation
CN112787495B (en) Variable frequency controller and control method thereof, variable frequency electric appliance and electronic equipment
CN110513846B (en) Control method for air conditioner compressor without electrolytic capacitor
CN108809177B (en) Electrolytic capacitor-free motor driving method, electrolytic capacitor-free motor driving device, electronic apparatus, and storage medium
CN109039180B (en) Fractional order control method for grid connection process of doubly-fed induction generator
CN109039191B (en) Motor control method for optimized operation of Quasi-Z source indirect matrix converter
CN114928076B (en) Double closed-loop control method of virtual synchronous machine without alternating-current voltage sensor
CN114301361B (en) Control method of electrolytic capacitor-free permanent magnet synchronous motor driving system based on bus current control

Legal Events

Date Code Title Description
PB01 Publication
PB01 Publication
SE01 Entry into force of request for substantive examination
SE01 Entry into force of request for substantive examination
GR01 Patent grant
GR01 Patent grant