CN109660183B - Capacitor miniaturization motor driving device - Google Patents

Capacitor miniaturization motor driving device Download PDF

Info

Publication number
CN109660183B
CN109660183B CN201811581940.9A CN201811581940A CN109660183B CN 109660183 B CN109660183 B CN 109660183B CN 201811581940 A CN201811581940 A CN 201811581940A CN 109660183 B CN109660183 B CN 109660183B
Authority
CN
China
Prior art keywords
current
voltage
conversion circuit
motor
waveform generator
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Expired - Fee Related
Application number
CN201811581940.9A
Other languages
Chinese (zh)
Other versions
CN109660183A (en
Inventor
霍军亚
王高林
朱良红
张国柱
徐殿国
赵楠楠
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Harbin Institute of Technology
Original Assignee
Harbin Institute of Technology
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Harbin Institute of Technology filed Critical Harbin Institute of Technology
Priority to CN201811581940.9A priority Critical patent/CN109660183B/en
Publication of CN109660183A publication Critical patent/CN109660183A/en
Application granted granted Critical
Publication of CN109660183B publication Critical patent/CN109660183B/en
Expired - Fee Related legal-status Critical Current
Anticipated expiration legal-status Critical

Links

Images

Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P29/00—Arrangements for regulating or controlling electric motors, appropriate for both AC and DC motors
    • H02P29/50—Reduction of harmonics
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation

Landscapes

  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Control Of Ac Motors In General (AREA)
  • Control Of Motors That Do Not Use Commutators (AREA)

Abstract

The invention provides a capacitor miniaturization motor driving device which comprises a control part, an inductor, an alternating current-direct current conversion circuit, a direct current link part and a direct current-alternating current conversion circuit. The AC-DC conversion circuit is used for supplying power voltage v to the AC power supplyinFull-wave rectification is performed, the DC link part is provided with a capacitor connected in parallel with the output side of the AC-DC conversion circuit and outputs pulsating DC voltage vdcThe dc-ac conversion circuit converts the output of the dc link unit into ac by a switch, supplies the ac to the connected permanent magnet synchronous motor, and controls the switch by the control unit. The driving device comprises two waveform generators, the waveform generators are automatically switched according to the running state of the system, and harmonic optimization and phase current peak value optimization of a pressing machine are considered; the system calculates LC resonance suppression current according to the input current and the voltage of the direct current bus, and adds the current to the Q-axis current instruction to realize system LC resonance suppression.

Description

Capacitor miniaturization motor driving device
Technical Field
The invention belongs to the technical field of motor driving, and particularly relates to a capacitor miniaturized motor driving device.
Background
Along with the improvement of energy conservation requirements of consumers on electromechanical products, the variable frequency motor driver with higher efficiency is more and more widely applied. The direct current bus voltage of the conventional variable frequency driver is in a stable state, and the inversion part is relatively independent from the input alternating current voltage, so that the control of the inversion part does not need to consider the instantaneous change of the input voltage, and the control method is convenient to realize. However, this design method requires an electrolytic capacitor with a large capacitance, which increases the size and cost of the driver. In addition, electrolytic capacitors have a limited lifetime and their effective operating time tends to be a bottleneck for the lifetime of the drive.
In order to solve the problems, a strategy of replacing an electrolytic capacitor with a thin film capacitor or a ceramic capacitor with a small capacitance value is provided in the related scheme, compared with a conventional AC/DC/AC driving circuit, a PFC part is omitted, and the miniaturized capacitor can not only realize cost reduction, but also eliminate service life bottleneck caused by the electrolytic capacitor. However, the capacitance value of the thin film capacitor or the ceramic capacitor on the direct-current bus voltage is very small, and is usually only 1% -2% of that of the conventional high-voltage electrolytic capacitor; the voltage of the direct current bus fluctuates greatly along with the input voltage of the power supply, the lowest voltage is only dozens of volts, and the lowest voltage of the direct current bus needs to be controlled to ensure the stability of a control system; further, when the inverter works, the capacitance of the bus and the inductance L on the ac power supply side generate LC resonance, which causes large control instability of system harmonic, and a special control strategy needs to be added to solve the problem, eliminate LC resonance, and realize stable operation of the compressor.
Disclosure of Invention
The invention provides a capacitor miniaturization motor driving device for overcoming the defects in the prior art. The invention can enable the input current waveform of the motor to meet the harmonic wave requirement, can ensure the stability of the speed regulating system, calculates the LC resonance suppression current according to the input current and the DC bus voltage, and adds the current to the Q-axis current instruction to realize the LC resonance suppression of the system.
The purpose of the invention is realized by the following technical scheme: a capacitive miniaturized motor drive apparatus comprising: a control unit 2, an inductor 3, an ac/dc conversion circuit 4, a dc link unit 5, and an ac/dc conversion circuit 6; the AC-DC conversion circuit 4 is used for supplying power voltage v to the AC power supply 1inFull-wave rectification is performed, one end of the inductor 3 is connected to an alternating current power supply 1, the other end is connected to an alternating current/direct current conversion circuit 4, the direct current link part 5 has a capacitor 5a connected in parallel to the output side of the alternating current/direct current conversion circuit 4, and outputs a pulsating direct current voltage vdcThe dc/ac conversion circuit 6 converts the output of the dc link unit 5 into ac by a switch and supplies the ac to the permanent magnet synchronous motor 7 connected thereto, and the control unit 2 receives a speed command
Figure BDA0001918097490000021
Detecting voltage v of input powerinPhase thetageCurrent iinDC bus voltage vdcAnd motor current iu、iv、iwAnd outputs a control instruction T of the DC/AC conversion circuit 6u、Tv、TwAnd motor control is realized.
Further, the control section 2 includes a waveform generator module according to vin、θgeCalculating the waveform of the Q-axis current waveform generator according to the motor load; the Q-axis current waveform generator waveform has two shapes, including:
waveform generator shape 1:
Figure BDA0001918097490000022
waveform generator shape 2:
Figure BDA0001918097490000023
Figure BDA0001918097490000024
wherein, Wf(θge) As output variable, vinFor real-time detection of the value of the supply voltage, VθdTo this end the phase of the mains voltage within a half period of the mains voltage is thetadVoltage of time, VPeakIs the magnitude of the supply voltage, θgeFor the phase estimate of the input voltage, thetadThe phase corresponding to the current dead zone;
the shape of the waveform generator is determined according to the strategy of using the waveform generator.
Further, the waveform generator usage strategy includes:
when motor frequency omegae>ωhighThe waveform generator shape 2 is selected when the motor frequency omegae<ωlowTime-selective waveform generator shape1 when ω islow≦ωe≦ωhighKeeping the current waveform generator unchanged; or when the DC-AC conversion circuit 6 outputs power Pinv>PhighThe waveform generator shape 2 is selected when the DC-AC conversion circuit 6 outputs the power Pinv<PlowThe waveform generator shape 1 is selected when P islow≦Pinv≦PhighKeeping the current waveform generator unchanged;
the power of the direct-alternating current conversion circuit 6 is calculated according to the following formula:
Pinv＝Vuiu+Vviv+Vwiw
wherein, Vu，Vv，VwAre three-phase voltage commands i of the DC-AC conversion circuit 6u, v and w respectivelyu、iv、iwThe three-phase actual current of the motor is respectively.
Further, the Q-axis current initial command value is calculated by the following formula:
Figure BDA0001918097490000031
Figure BDA0001918097490000032
in the formula TpIndicating a torque command, iq_ref0Indicates the initial command value of the Q-axis current,
Figure BDA0001918097490000033
representing the rotor speed estimate, KeIs the motor back electromotive force coefficient, Ld、LqAre respectively DQ axis inductance, id_refIs a D-axis current command value, KPAs a proportional coefficient of the controller, KiIs the integral coefficient of the controller.
Further, the control part 2 further comprises a capacitance current compensation module for calculating capacitance power Pc：
Figure BDA0001918097490000034
Compensated current command iqccComprises the following steps:
Figure BDA0001918097490000035
wherein, thetageC is the capacitance value of a capacitor connected in parallel between the input ends of the DC-AC conversion circuit 6, V is the phase estimation value of the input voltagePeakIs the amplitude of the voltage, omega, of the AC power supplyinIs the voltage frequency of the AC power supply, p is the number of pole pairs of the motor, omegaeIs the motor rotor speed.
Further, the control unit 2 further includes an LC resonance suppression module for calculating an LC resonance suppression compensation current value:
instantaneous power compensation quantity Pcom＝vdc×K×LPF(iin)
Compensating current
Figure BDA0001918097490000036
Where K is the compensation coefficient, LPF (i)in) Represents the pair iinAnd (4) low-pass filtering.
Further, the total current command value of the Q axis is:
iq_ref1＝iq_ref0+iqcom+iqcc。
further, the control part 2 further comprises a weak magnetic control module for calculating a weak magnetic current:
Figure BDA0001918097490000037
wherein iq_ref1Is the total current command value of the Q axis, KeIs the motor back electromotive force coefficient.
Further, the control part 2 further comprises a current amplitude limiting control module for realizing the limitation of the DQ output current; the final DQ axis current command is calculated according to the following formula:
Figure BDA0001918097490000041
Figure BDA0001918097490000042
wherein imaxThe maximum current value allowed to be output by the dc/ac conversion circuit 6.
Further, the control section 2 obtains a final DQ-axis current command id_refAnd iq_refAnd detecting and calculating the actual current i of the DQdAnd iqRespectively carrying out PI control on the D-axis current and the Q-axis current, and then adding decoupling and calculating to obtain a DQ axis voltage command VdAnd VqAnd then the alpha and beta axis voltage instruction V is obtained through coordinate conversionαAnd VβThen converted into a u, V and w three-phase voltage command Vu、Vv、VwFinally, the sum V is calculatedu、Vv、VwEquivalent pulse Tu、Tv、TwAnd output to the motor through a dc-ac conversion circuit 6.
Drawings
FIG. 1 is a block diagram of a capacitive miniaturized motor driving apparatus according to the present invention;
FIG. 2 is a block diagram of a control structure of the capacitor miniaturized motor driving apparatus according to the present invention;
FIG. 3 is a phase locked loop block diagram;
FIG. 4 is a block diagram of an LC resonance suppression module;
fig. 5 is a block diagram of a DQ axis current clipping control module.
Detailed Description
The technical solutions in the embodiments of the present invention will be described clearly and completely with reference to the accompanying drawings in the embodiments of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention.
Fig. 1 is a schematic structural diagram of a capacitor-miniaturized motor driving apparatus according to an embodiment of the present invention.
It should be noted that the capacitor miniaturized motor driving apparatus according to the embodiment of the present invention may be applied to an inverter motor, and referring to fig. 1, in a circuit of the inverter motor, an AC power supply AC is connected to the motor through a rectifier circuit and an inverter circuit, and in the embodiment of the present invention, a thin film capacitor or a ceramic capacitor 5a having a small capacitance value may be connected in parallel between input terminals of the inverter circuit.
As shown in fig. 1, the capacitor-miniaturized motor driving apparatus according to the embodiment of the present invention includes: a control unit 2, an inductor 3, an ac-dc converter circuit 4, a dc link unit 5, and a dc-ac converter circuit 6.
In fig. 1, a module 1 is a system power supply, and a module 7 is an equivalent circuit diagram of a permanent magnet synchronous motor. The AC-DC conversion circuit 4 is used for supplying power voltage v to the AC power supply 1inFull-wave rectification is performed, and the dc link unit 5 has a capacitor 5a connected in parallel with the output side of the ac/dc conversion circuit 4 and outputs a pulsating dc voltage vdcThe dc/ac conversion circuit 6 converts the output of the dc link unit 5 into ac by a switch, supplies the ac to the connected permanent magnet synchronous motor 7, and controls the switch by the control unit 2. The control part 2 is used for receiving a speed command
Figure BDA0001918097490000051
Detecting voltage v of input powerinPhase thetageCurrent iinDC bus voltage vdcAnd motor current iu、iv、iwAnd outputs a control instruction T of the DC/AC conversion circuit 6u、Tv、TwAnd motor control is realized. The direct-alternating current conversion circuit 6 is an inverter circuit.
FIG. 3 illustrates an input voltage phase detection PLL module for obtaining an instantaneous voltage value V of an input AC power sourcegeAnd according to the instantaneous value V of the voltage of the AC power supplygeCalculating an input voltage phase estimate θge。
Specifically, as shown in fig. 3, the input voltage phase detection phase-locked loop module may include a cosine calculator, a first multiplier, a low-pass filter, a first PI regulator, and an integrator. Wherein the cosine calculator is used for estimating the phase of the input voltage in the last calculation periodgePerforming cosine calculation to obtain a first calculated value, and a first multiplier for multiplying the instantaneous voltage value V of the AC power supplygeThe first calculated value is multiplied by the second calculated value to obtain a second calculated value. The low-pass filter is used for low-pass filtering the second calculated value to obtain a third calculated value, wherein the bandwidth of the low-pass filter is lower than the voltage frequency of the alternating current power supply, and in one embodiment of the invention, the bandwidth of the low-pass filter is lower than the voltage frequency omega of the alternating current power supply g1/5 of (1). The first PI regulator is used for performing PI regulation on the third calculated value to output a fourth calculated value, and the integrator is used for performing PI regulation on the fourth calculated value and the voltage frequency omega of the alternating current power supplygThe sum is subjected to integral calculation to obtain an input voltage phase estimation value theta of the current calculation periodge。
A position/velocity estimator estimates a rotor position of the electric machine to obtain a rotor angle estimate
Figure BDA0001918097490000052
And rotor speed estimate
Figure BDA0001918097490000053
The motor of the embodiment of the present invention may be a motor without a position sensor, and in one embodiment of the present invention, the above-described function of the position/velocity estimator may be implemented by a flux linkage observation method. Firstly, the estimated values of the effective magnetic flux of the motor in the directions of the alpha and beta axes of the fixed coordinate system can be calculated according to the current and the voltage of the fixed coordinate system, and the specific calculation formula is as follows:
Figure BDA0001918097490000061
wherein,
Figure BDA0001918097490000062
and
Figure BDA0001918097490000063
estimated values of the effective flux of the machine in the alpha and beta directions, v, respectivelyαAnd vβVoltages in the alpha and beta axis directions, iαAnd iβThe current in the alpha and beta axis directions, respectively.
The rotor angle estimate is then further calculated
Figure BDA0001918097490000064
And rotor speed estimate
Figure BDA0001918097490000065
The specific calculation formula is as follows:
Figure BDA0001918097490000066
Figure BDA0001918097490000067
wherein, Kp_pllAnd Ki_pllProportional and integral parameters, ω, of the phase-locked loop PI controller, respectivelyfIs the bandwidth of the velocity low pass filter, θerrIs an estimate of the deviation angle.
As shown in fig. 2, the Q-axis current instruction calculation module includes a second PI regulator, a waveform generator, an initial current calculation unit, a capacitance current compensation unit, and a superposition unit.
The Q-axis current command calculation module includes:
a second PI regulator for PI regulating a difference between the motor speed command and the rotor speed estimate to output a torque amplitude command;
a waveform generator for generating an output variable from the input voltage phase estimate;
an initial current calculation unit, wherein the initial current calculation unit is used for multiplying the output variable by the torque amplitude instruction and then dividing the multiplied output variable by a motor torque coefficient to obtain a Q-axis current instruction initial value;
the capacitance current compensation unit is used for generating compensation current according to the input voltage phase estimation value;
a superimposing unit configured to superimpose the compensation current on the Q-axis current instruction initial value to obtain the Q-axis current instruction.
Wherein the second PI regulator is used for regulating the command speed
Figure BDA0001918097490000068
And estimating the velocity
Figure BDA0001918097490000069
Performing PI control after difference is made, and outputting a torque command Tp。
Figure BDA00019180974900000610
Wherein KPAs a proportional coefficient of the controller, KiIs the integral coefficient of the controller.
The waveform generator is used for generating an output variable W according to the shape and phase of an input voltagef。
An initial current calculating unit for calculating an output variable WfAnd torque amplitude command TpAfter multiplication, the value is further converted into an initial value i of a Q-axis current commandq_ref0。
Figure BDA0001918097490000071
In an embodiment of the present invention, two shape waveform generators are included:
waveform generator shape 1 calculates the output variable according to the following formula:
Figure BDA0001918097490000072
waveform generator shape 2 calculates the output variable according to the following equation:
Figure BDA0001918097490000073
Figure BDA0001918097490000074
wherein, Wf(θge) Is the output variable, vinFor real-time detection of the value of the supply voltage, VθdTo this end the phase of the mains voltage within a half period of the mains voltage is thetadVoltage of time, VPeakIs the magnitude of the supply voltage, θgeFor said input voltage phase estimate, θdThe phase corresponding to the current dead zone.
The waveform generator 1 has the advantages that the input current harmonic wave is small; the disadvantage is that the peak value of the motor phase current is large.
Compared with the waveform generator 1, the waveform generator 2 has the disadvantages of large input current harmonic and small motor phase current peak value.
When in specific use, the frequency omega is determined according to the running frequency of the motoreIt is decided which waveform generator to use. In particular, when the motor frequency ωe>ωhighThe waveform generator 2 is selected when the motor frequency omegae<ωlowThe waveform generator 1 is selected when ω islow≦ωe≦ωhighWhile keeping the current waveform generator unchanged. Wherein ω ishigh、ωlowThe value relation is omegahigh>ωlowIn the present embodiment, ωhighIs 150Hz, omegalowIs 130 Hz.
The specific selection method of the waveform generator can also be realized by the following ways:
when in specific use, the power P is output according to the DC-AC conversion circuit 6invIt is decided which waveform generator to use. In particular, when the motor frequency Pinv>PhighThe waveform generator 2 is selected when the motor frequency P isinv<PlowThe waveform generator 1 is selected when P islow≦Pinv≦PhighWhile keeping the current waveform generator unchanged. Wherein P ishigh、PlowThe value relationship is as follows, Phigh>PlowIn this embodiment, PhighIs 1100W, PlowIs 900W.
The power of the direct-alternating current conversion circuit 6 is calculated according to the following formula:
Pinv＝Vuiu+Vviv+Vwiw
wherein, Vu，Vv，VwAre three-phase voltage commands i of the DC-AC conversion circuit 6u, v and w respectivelyu、iv、iwThe three-phase actual current of the motor is respectively.
In an embodiment of the present invention, the capacitance current compensation unit may calculate the compensation current according to the following formula:
Figure BDA0001918097490000081
wherein, thetageC is capacitance value of capacitor connected in parallel between input ends of the inverter circuit, V is estimated value of phase of the input voltagePeakIs the voltage amplitude, omega, of the AC sourceinIs the voltage frequency of the AC power supply, p is the number of pole pairs of the motor, KeIs the motor back electromotive force coefficient, Ld、LqAre respectively DQ axis inductance, id_refIs a D-axis current command value, ωeIs the motor rotor speed.
In one embodiment of the invention, the phase parameter θ is setdThe phase corresponding to the current dead zone can be selected as 0.1-0.2 rad by default.
Referring to fig. 4, in the embodiment of the invention, the LC resonance suppressing unit may subtract the low-pass filtered value from the input current, amplify the subtracted value in proportion, and multiply the amplified value by the dc bus voltage vdcSo as to obtain the instantaneous power compensation quantity PcomAnd then further calculating to obtain a compensation current iqcom. Specifically, the compensation current is calculated according to the following formula:
Pcom＝vdc×K×LPF(iin)
Figure BDA0001918097490000082
where K is the compensation coefficient, LPF (i)in) Represents the pair iinAnd (4) low-pass filtering.
Q-axis current initial command value iq_ref0Adding a current instruction value i output by an LC resonance suppression moduleqcomAnd a current instruction value i output by the capacitance current compensation moduleqccObtaining a current command value i of the Q axisq_ref1。
iq_ref1＝iq_ref0+iqcom+iqcc
The weak magnetic control module calculates a weak magnetic current instruction i according to the following formulad_ref1：
Figure BDA0001918097490000091
In conjunction with FIG. 5, a further embodiment is based on obtaining a DQ-axis current command id_ref1And iq_ref1Performing amplitude limiting control to satisfy
Figure BDA0001918097490000092
Specifically, the final DQ axis current command is calculated according to the following formula:
Figure BDA0001918097490000093
Figure BDA0001918097490000094
wherein imaxThe maximum current value allowed to be output by the dc/ac conversion circuit 6.
Further, the present embodiment obtains a DQ-axis current command id_refAnd iq_refAnd detecting and calculating the actual current i of the DQdAnd iqRespectively carrying out PI control on the D-axis current and the Q-axis current, and then adding decoupling and calculating to obtain a DQ axis voltage command VdAnd VqAnd then the alpha and beta axis voltage instruction V is obtained through coordinate conversionαAnd VβThen converted into a u, V and w three-phase voltage command Vu、Vv、VwFinally, the sum V is calculatedu、Vv、VwEquivalent pulse Tu、Tv、TwAnd output to the motor through the inverter circuit.
Specifically, the Q-axis voltage command and the D-axis voltage command may be calculated according to the following formulas:
Figure BDA0001918097490000095
Figure BDA0001918097490000096
Vd＝Vd0-ωeLqiq
Vq＝Vq0+ωeLdid+ωeKe
wherein, VqFor Q-axis voltage command, VdFor D-axis voltage command, KdpAnd KdiProportional gain and integral gain, K, respectively, for D-axis current controlqpAnd KqiProportional gain and integral gain, omega, respectively, for Q-axis current controleIs the motor speed, KeIs the motor back electromotive force coefficient, LdAnd LqAre respectively provided withAre D-axis and Q-axis inductors,
Figure BDA0001918097490000097
denotes the integral of x (τ) over time.
Obtaining the Q-axis voltage command VqAnd D-axis voltage command VdThen, V can be adjusted according to the rotor angle theta of the motorqAnd VdCarrying out Park inverse transformation to obtain a voltage command V on a fixed coordinate systemαAnd VβThe concrete transformation formula is as follows:
Vα＝Vd cosθ-Vq sinθ
Vβ＝Vd sinθ+Vq cosθ
where θ is the motor rotor angle, where the rotor angle estimate θ can be takenest。
Further, the voltage command V on the fixed coordinate system can be usedαAnd VβPerforming Clark inverse transformation to obtain three-phase voltage command Vu、VvAnd VwThe concrete transformation formula is as follows:
Vu＝Vα
Figure BDA0001918097490000101
Figure BDA0001918097490000102
then the duty ratio calculation unit can calculate the duty ratio according to the DC bus voltage and the three-phase voltage instruction to obtain a duty ratio control signal, namely a three-phase duty ratio Tu、TvAnd TwThe specific calculation formula is as follows:
Tu＝(Vu+0.5Vdc)/Vdc
Tv＝(Vv+0.5Vdc)/Vdc
Tw＝(Vw+0.5Vdc)/Vdc
wherein, VdcIs the dc bus voltage.
The duty ratio control signal controls the switch of the inverter circuit in real time, and the control of the motor is realized.
According to the capacitor miniaturized motor driving device provided by the embodiment of the invention, relevant parameters are obtained through an input voltage phase detection phase-locked loop module, a position/speed estimator and the like, two waveform generators are designed, an LC resonance suppression compensation module is designed, a Q-axis current instruction and a D-axis current instruction are calculated, then the Q-axis voltage instruction and the D-axis voltage instruction are further obtained, a duty ratio control signal is generated, and therefore an inverter circuit is controlled through the duty ratio control signal to control a motor. Therefore, the waveform generator can be automatically switched according to the system running state, the harmonic optimization and the phase current peak value optimization of the pressing machine are considered, the input current waveform of the motor meets the harmonic requirement, the LC resonance compensation current is calculated according to the input current and the direct current bus voltage, the current is added to the Q-axis current instruction, and the LC resonance suppression and the stable running of the speed regulating system are realized.
The capacitor miniaturization motor driving device provided by the invention is described in detail, a specific example is applied in the text to explain the principle and the implementation mode of the invention, and the description of the embodiment is only used for helping to understand the method and the core idea of the invention; meanwhile, for a person skilled in the art, according to the idea of the present invention, there may be variations in the specific embodiments and the application scope, and in summary, the content of the present specification should not be construed as a limitation to the present invention.

Claims (9)

1. A capacitor-miniaturized motor drive device, comprising: a control unit (2), an inductor (3), an AC/DC conversion circuit (4), a DC link unit (5), and a DC/AC conversion circuit (6); the AC-DC conversion circuit (4) is used for supplying power voltage v to the AC power supply (1)inFull-wave rectification is performed, one end of the inductor (3) is connected with an alternating current power supply (1), the other end is connected with an alternating current-direct current conversion circuit (4), and the direct current chain part (5) is provided with a capacitor connected with the alternating current-direct current conversion circuitA capacitor (5a) connected in parallel with the output side of the AC/DC conversion circuit (4) and outputting a pulsating DC voltage vdcThe DC/AC conversion circuit (6) converts the output of the DC link unit (5) into AC by means of a switch and supplies the AC to a permanent magnet synchronous motor (7) connected thereto, and the control unit (2) receives a speed command
Figure FDA0002882160340000011
Detecting voltage v of AC power supplyinCurrent iinDC bus voltage vdcThree-phase actual current i of motoru、iv、iwAnd an input voltage phase estimate θgeAnd outputs a pulse control instruction T of the direct current-alternating current conversion circuit (6)u、Tv、TwThe motor control is realized;
the control unit (2) includes a waveform generator module according to vin、θgeCalculating the waveform of the Q-axis current waveform generator according to the motor load; the Q-axis current waveform generator waveform has two shapes, including:
waveform generator shape 1:
Figure FDA0002882160340000012
waveform generator shape 2:
Figure FDA0002882160340000013
Figure FDA0002882160340000014
wherein, Wf(θge) As output variable, vinFor detecting the voltage of the AC power supply, VθdTo this end the phase of the mains voltage within a half period of the mains voltage is thetadVoltage of time, VPeakIs the magnitude of the supply voltage, θdIs an electric currentThe phase corresponding to the dead zone;
the shape of the waveform generator is determined according to the strategy of using the waveform generator.
2. The capacitive miniaturized motor driver of claim 1 wherein the waveform generator usage strategy comprises:
when motor frequency omegae>ωhighThe waveform generator shape 2 is selected when the motor frequency omegae<ωlowThe waveform generator shape 1 is selected when ω islow≦ωe≦ωhighKeeping the current waveform generator unchanged; or when the DC-AC conversion circuit (6) outputs power Pinv>PhighThe waveform generator shape 2 is selected when the output power P of the DC-AC conversion circuit (6)inv<PlowThe waveform generator shape 1 is selected when P islow≦Pinv≦PhighKeeping the current waveform generator unchanged;
the power of the direct current-alternating current conversion circuit (6) is calculated according to the following formula:
Pinv＝Vuiu+Vviv+Vwiw
wherein, Vu，Vv，VwU, v and w three-phase voltage commands i of the DC-AC conversion circuit (6)u、iv、iwThe three-phase actual current of the motor is respectively.
3. The capacitive miniaturized motor driving device according to claim 2, wherein the Q-axis current initial command value is calculated by the following formula:
Figure FDA0002882160340000021
Figure FDA0002882160340000022
in the formula TpIndicating a torque command, iq_ref0Indicates the initial command value of the Q-axis current,
Figure FDA0002882160340000023
representing the rotor speed estimate, KeIs the motor back electromotive force coefficient, Ld、LqAre respectively DQ axis inductance, id_refIs a D-axis current command value, KPAs a proportional coefficient of the controller, KiIs the integral coefficient of the controller.
4. A capacitive miniaturised motor drive according to claim 3 characterised in that the control part (2) further comprises a capacitive current compensation module for calculating the capacitive power Pc：
Figure FDA0002882160340000024
Compensated current command iqccComprises the following steps:
Figure FDA0002882160340000025
wherein, thetageC is the capacitance value of a capacitor connected in parallel between the input ends of the DC-AC conversion circuit (6) for the phase estimation value of the input voltage, VPeakIs the amplitude of the supply voltage, omegainIs the voltage frequency of the AC power supply, p is the number of pole pairs of the motor, omegaeIs the motor frequency.
5. The capacitance-type miniaturized motor driving device according to claim 4, wherein the control section (2) further includes an LC resonance suppression module for calculating an LC resonance suppression compensation current value:
instantaneous power compensation quantity Pcom＝vdc×K×LPF(iin)
Compensating current
Figure FDA0002882160340000031
Where K is the compensation coefficient, LPF (i)in) Represents the pair iinAnd (4) low-pass filtering.
6. The capacitor-miniaturized motor-driving device of claim 5, wherein the total Q-axis current command value is:
iq_ref1＝iq_ref0+iqcom+iqcc。
7. the capacitive miniaturized motor driving device according to claim 6, wherein said control section (2) further comprises a field weakening control module for calculating a field weakening current:
Figure FDA0002882160340000032
wherein iq_ref1Is the total current command value of the Q axis, KeIs the motor back electromotive force coefficient.
8. The capacitive miniaturized motor driving device according to claim 7, wherein said control section (2) further comprises a current clipping control module for achieving a DQ output current limitation; the final DQ axis current command value is calculated according to the following formula:
Figure FDA0002882160340000033
Figure FDA0002882160340000034
wherein imaxThe maximum current value allowed to be output by the direct current-alternating current conversion circuit (6).
9. The capacitor-miniaturized motor driving device according to claim 8, wherein the control unit (2) obtains a final DQ-axis current command value i'd_refAnd i'q_refAnd detecting and calculating the actual current i of the DQ axisdAnd iqRespectively carrying out PI control on the D-axis current and the Q-axis current, and then adding decoupling and calculating to obtain a DQ axis voltage command VdAnd VqAnd then the alpha and beta axis voltage instruction V is obtained through coordinate conversionαAnd VβThen converted into a u, V and w three-phase voltage command Vu、Vv、VwFinally, the sum V is calculatedu、Vv、VwEquivalent pulse control command Tu、Tv、TwAnd output to the motor through a direct current-to-alternating current conversion circuit (6).
CN201811581940.9A 2018-12-24 2018-12-24 Capacitor miniaturization motor driving device Expired - Fee Related CN109660183B (en)

Priority Applications (1)

Application Number Priority Date Filing Date Title
CN201811581940.9A CN109660183B (en) 2018-12-24 2018-12-24 Capacitor miniaturization motor driving device

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
CN201811581940.9A CN109660183B (en) 2018-12-24 2018-12-24 Capacitor miniaturization motor driving device

Publications (2)

Publication Number Publication Date
CN109660183A CN109660183A (en) 2019-04-19
CN109660183B true CN109660183B (en) 2021-03-12

Family

ID=66116522

Family Applications (1)

Application Number Title Priority Date Filing Date
CN201811581940.9A Expired - Fee Related CN109660183B (en) 2018-12-24 2018-12-24 Capacitor miniaturization motor driving device

Country Status (1)

Country Link
CN (1) CN109660183B (en)

Families Citing this family (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN114785210B (en) * 2022-06-04 2022-10-11 中国矿业大学 Permanent magnet synchronous motor current controller parameter setting method, device and system

Family Cites Families (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2012222842A (en) * 2011-04-04 2012-11-12 Panasonic Corp Motor drive device and electric appliance using the same
JP5807156B2 (en) * 2011-07-21 2015-11-10 パナソニックＩｐマネジメント株式会社 Motor drive inverter control circuit and vacuum cleaner
BR112016006158B1 (en) * 2013-09-30 2022-04-05 Daikin Industries Ltd. power converter device
CN105978433B (en) * 2016-05-31 2018-12-18 广东美的制冷设备有限公司 The motor driver and transducer air conditioning of capacitor miniaturization
CN106505527A (en) * 2016-12-19 2017-03-15 广东美的制冷设备有限公司 Motor drive protection device, over-voltage protection method and transducer air conditioning

Also Published As

Publication number Publication date
CN109660183A (en) 2019-04-19

Similar Documents

Publication Publication Date Title
CN109546913B (en) Capacitor miniaturization motor driving device
CN106559026B (en) A kind of control method of motor driven systems, control device and transducer air conditioning
KR101995864B1 (en) Inverter control apparatus and method thereof
US9257931B2 (en) Power conversion apparatus
US7053569B2 (en) Inverter control method and its device
CN106026072B (en) double-PWM converter direct-current bus voltage fluctuation suppression method and control method
WO2018113388A1 (en) Motor driving protection device, over-voltage protection method and inverter air conditioner
US9214881B2 (en) Power conversion apparatus
CN105515354B (en) Method, device and system for controlling compressor
CN106208868B (en) No electrolytic capacitor motor driven systems and its control method, device
CN105978433B (en) The motor driver and transducer air conditioning of capacitor miniaturization
KR101621994B1 (en) Control apparatus for regenerative medium voltage inverter
CN106208887B (en) No electrolytic capacitor motor driven systems and its control method, device
JP2008113514A (en) Power supply circuit and control circuit therewith
CN111800037B (en) Electrolytic-capacitor-free control system, control method, compressor and refrigeration equipment
JP5813934B2 (en) Power converter
JP4909857B2 (en) Converter device
CN109660183B (en) Capacitor miniaturization motor driving device
RU2486658C1 (en) Electric motor control device
JP5741000B2 (en) Power converter
JP4439846B2 (en) Multiphase current supply circuit
JP5477435B2 (en) Stability determination method, power conversion device
JP5838554B2 (en) Power converter
JP6003169B2 (en) Control method and control device
CN118473288A (en) Frequency converter motor system

Legal Events

Date Code Title Description
PB01 Publication
PB01 Publication
SE01 Entry into force of request for substantive examination
SE01 Entry into force of request for substantive examination
GR01 Patent grant
GR01 Patent grant
CF01 Termination of patent right due to non-payment of annual fee
CF01 Termination of patent right due to non-payment of annual fee

Granted publication date: 20210312