CN109347387B - Motor control method and control device based on model prediction - Google Patents

Motor control method and control device based on model prediction Download PDF

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CN109347387B
CN109347387B CN201811320909.XA CN201811320909A CN109347387B CN 109347387 B CN109347387 B CN 109347387B CN 201811320909 A CN201811320909 A CN 201811320909A CN 109347387 B CN109347387 B CN 109347387B
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axis
carrier
quadrature
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CN109347387A (en
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胡余生
郭伟林
张良浩
于安波
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Gree Electric Appliances Inc of Zhuhai
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • H02P27/12—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation pulsing by guiding the flux vector, current vector or voltage vector on a circle or a closed curve, e.g. for direct torque control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation

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Abstract

The application relates to the technical field of motor control, and provides a motor control method and a motor control device based on model prediction.

Description

Motor control method and control device based on model prediction
Technical Field
The present disclosure relates to the field of motor control technologies, and in particular, to a motor control method and a motor control device based on model prediction.
Background
The control method of the permanent magnet synchronous motor based on model prediction can be mainly divided into direct prediction current control and PWM prediction current control according to different voltage vector action modes. Where PWM predictive current control is also referred to as deadbeat control. The method utilizes a current instruction and motor current and position information obtained by sampling in the period, accurately calculates a voltage vector to be acted in the next control period according to a motor discrete model, and modulates the voltage vector to act on the motor through SVPWM (space vector pulse width modulation), so that the motor current can accurately follow the current instruction value after the voltage vector is acted for one period. The SVPWM modulation strategy is the key to the realization of dead-beat control.
The SVPWM modulation technology has clear physical concept and higher direct-current voltage utilization rate, and adopts a seven-segment symmetric modulation mode of zero vector and effective vector. However, the symmetric vector distribution method is not a distribution method with minimum current ripple, and the constant switching frequency and the symmetric duty ratio cause problems such as current harmonics and noise pollution.
Disclosure of Invention
In order to solve the above technical problem or at least partially solve the above technical problem, the present application provides a motor control method and a control device based on model prediction.
The embodiment of the application provides a motor control method based on model prediction, which comprises the following steps:
acquiring a control command and motor parameters, and calculating a predicted voltage value according to model prediction;
performing coordinate conversion on the predicted voltage value to obtain an α axis component and a β axis component;
obtaining adjacent vector action time of different sectors according to the α axis component and the β axis component;
calculating zero vector action time with symmetric quadrature axis current errors according to the adjacent vector action time;
calculating the switch conduction time point according to the zero vector acting time;
and modulating and outputting a PWM pulse signal to act on the control motor.
Optionally, the step of obtaining the control command and the current cycle motor parameter includes: within one carrier period, the duty cycle is sampled and updated at the time when the carrier is zero and the carrier maximum, respectively.
Optionally, the calculating a zero vector action time with symmetric quadrature axis current error according to the adjacent vector action time includes: and calculating the zero vector action time with symmetrical quadrature axis current errors by reverse deduction of a prediction model according to the adjacent vector action time.
Optionally, the calculating a zero vector action time with symmetric quadrature axis current error according to the adjacent vector action time includes: obtaining quadrature axis voltage reference vectors with symmetric quadrature axis current errors in a model prediction mode according to the action time of adjacent vectors; and calculating the zero vector acting time through reverse estimation of a prediction model according to the quadrature axis voltage reference vector.
Optionally, according to the zero vector acting time, the sum of the maximum value of the quadrature axis current error affected by the voltage and the minimum value of the quadrature axis current error affected by the voltage is zero.
Optionally, the step of modulating the output PWM pulse includes: and modulating and outputting a PWM pulse signal by using the amplitude of the triangular carrier wave and the three-phase modulation wave to act on the inverter.
Optionally, according to the PWM pulse signal output by the control method, in one carrier period, the first four segments of modulation signals are located in four periods when the carrier rises, and the last four segments of modulation signals are located in four periods when the carrier falls.
Optionally, the PWM pulse signal output according to the control method is asymmetric with respect to the maximum time of the carrier within one carrier period.
Optionally, the quadrature current error is symmetric with respect to the quadrature current setpoint.
Optionally, the zero vector action time T is calculated01The formula of (1) is:
Figure BDA0001857475980000021
Figure BDA0001857475980000031
in the formula, R is a stator resistor; l is stator inductance, Ld＝Lq＝L；ωeIs the rotor electrical angular velocity; psifIs a permanent magnet flux linkage; i.e. id、iqDirect axis and quadrature axis currents; i.e. id ref，iq refSetting values for direct-axis and quadrature-axis currents; k denotes the Kth control period, TsIs a sampling period; t is1、T2For adjacent vectors to doThe time of use; Δ iqRepresenting the quadrature-axis current variation component, u, affected by voltageref q(k) For quadrature voltage reference vectors, from adjacent effective vectors V1、V2Synthesized according to the duty ratio.
Optionally, the calculating a switch conducting time point by the zero vector acting time includes:
the resulting formula when the carrier is in the first half cycle is:
Figure BDA0001857475980000032
the resulting formula when the carrier is in the second half of the period is:
Figure BDA0001857475980000033
at the same time to T01Clipping is performed so that 0 ≦ T01≦Ts-T1-T2；
Wherein, Ta、Tb、TcA switch on time point; t is1、T2Acting time for adjacent vectors; t is01Zero vector action time.
The device comprises an obtaining module, a processing module and a generating module, wherein the obtaining module is used for obtaining a control command and motor parameters, the processing module is used for predicting and calculating a predicted voltage value according to a model, carrying out coordinate conversion on the predicted voltage value to obtain α axis components and β axis components, obtaining adjacent vector action time of different sectors according to the α axis components and the β axis components, calculating zero vector action time with symmetrical quadrature axis current errors according to the adjacent vector action time, calculating a switch conduction time point according to the zero vector action time, and the generating module is used for generating a control signal of the motor according to the switch conduction time point.
Optionally, the apparatus further includes a determining module, configured to monitor a zero time of the carrier in the carrier period and a maximum time of the carrier, and trigger the control device to sample and update the duty ratio at the two times.
Optionally, the control device is a PI controller, a model predictive controller, or an SVPWM controller.
Compared with the prior art, the technical scheme provided by the embodiment of the application has the following advantages:
according to the method provided by the embodiment of the application, the action time of the zero vector and the effective vector which enable the current ripple to be minimum is accurately calculated through the prediction model, and the asymmetric duty ratio which enables the current ripple to be minimum is obtained. Under the condition of not increasing the switching frequency, the control performance of the motor control current is obviously improved, and the current harmonic content is reduced.
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The accompanying drawings, which are incorporated in and constitute a part of this specification, illustrate embodiments consistent with the present application and together with the description, serve to explain the principles of the application.
In order to more clearly illustrate the embodiments of the present application or the technical solutions in the prior art, the drawings needed to be used in the description of the embodiments or the prior art will be briefly described below, and it is obvious for those skilled in the art to obtain other drawings without inventive exercise.
Fig. 1 is a schematic structural diagram of a control system framework according to an embodiment of the present application.
Fig. 2 is a schematic diagram of a modulation principle of a motor control method based on model prediction according to an embodiment of the present application.
FIG. 3 is a diagram of base vector and sector distribution in an embodiment of the present application.
Fig. 4 is a modulation flow chart of a motor control method based on model prediction according to an embodiment of the present application.
Fig. 5 is a schematic diagram illustrating a current variation process in a sampling period according to an embodiment of the present application.
Detailed Description
In order to make the objects, technical solutions and advantages of the embodiments of the present application clearer, the technical solutions in the embodiments of the present application will be clearly and completely described below with reference to the drawings in the embodiments of the present application, and it is obvious that the described embodiments are some embodiments of the present application, but not all embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present application.
The embodiment of the application provides a motor control method based on model prediction, which is illustrated with reference to fig. 4 and mainly comprises the following steps:
s101, calculating a predicted voltage value by using the acquired control command and motor parameters through a motor discrete model; in a specific embodiment of the present application, in one carrier period, sampling is performed at the time when the carrier is zero and the carrier maximum value, and correspondingly, duty ratio updating of the PWM pulse signal is performed at the two times. The specific calculation method can be performed by selecting the operation mode commonly used in the field.
And S102, performing coordinate conversion on the predicted voltage value to obtain α axis components and β axis components, wherein the predicted voltage value can be obtained by selecting a park inverse conversion mode, and a specific obtaining method can be performed by selecting a common operation mode in the field.
And S103, calculating to obtain the action time of adjacent vectors of different sectors by using the α axis component, the β axis component and the reference voltage vector, wherein the action time of the adjacent vectors can be obtained by selecting a table look-up method, and the specific obtaining method can be carried out by selecting a common operation mode in the field.
S104, calculating zero vector action time with symmetrical quadrature axis current errors according to the adjacent vector action time; optionally, quadrature axis voltage reference vectors with symmetric quadrature axis current errors can be obtained in a model prediction mode according to action time of adjacent vectors; and calculating the zero vector acting time through reverse estimation of a prediction model according to the quadrature axis voltage reference vector. And according to the zero vector acting time, the sum of the maximum value of the quadrature axis current error influenced by the voltage and the minimum value of the quadrature axis current error influenced by the voltage can be made to be zero.
S105, calculating a switch conduction time point according to the zero vector acting time; in a carrier period, the first four segments of modulation signals are located in four segments of time when the carrier rises, and the last four segments of modulation signals are located in four segments of time when the carrier falls. The specific derivation method can be performed by selecting the operation mode commonly used in the field.
And S106, modulating and outputting the PWM pulse according to the switch conducting time point, and acting on the control motor. The PWM pulse signal can be selectively modulated and output by a triangular carrier wave and a three-phase modulation wave amplitude value and acts on the inverter. Optionally, the PWM pulse signal output according to the control method is asymmetric with respect to the maximum time of the carrier within one carrier period.
The duty ratio optimization modulation method provided by the application can obviously improve the control performance of the motor control current and reduce the current harmonic content under the condition of not increasing the switching frequency. Specifically, simulation analysis shows that when the duty ratio optimization modulation method provided by the application is used for execution, the q-axis current fluctuation can be reduced by 6.3%. The peak value of the q-axis current fluctuation can be reduced by 12.5%. The current harmonic content of the improved method is 3.24%, the current harmonic content of the traditional method is 3.63%, and the harmonic content is reduced by 10.7%.
According to the duty ratio optimization modulation method, on the basis of a PWM prediction control structure of a permanent magnet synchronous motor and aiming at an SVPWM modulation technology, firstly, a double-sampling double-updating (in a single carrier period) control strategy is introduced, seven-segment modulation is divided into a front four segment and a rear four segment, and asymmetric effective vector action time is obtained. Meanwhile, the zero vector action time for minimizing the current ripple is deduced through a prediction model. And finally, the optimal asymmetric switching duty ratio is obtained. The control method provided by the application can effectively reduce the current ripple and the current harmonic content on the premise of not increasing the switching frequency.
In the conventional control method, the sampling period calculation period and the carrier period are both T, and sampling and duty ratio are only performed when the carrier is 0. The sampling period of the double-sampling double-updating strategy is changed into Ts which is 0.5 times of T, namely, twice sampling and twice calculation are carried out in a carrier period, and the duty ratio is sampled and updated at the moment when the carrier is 0 and the maximum value of the carrier respectively. The first four sections are four sections of time when the carrier wave rises, and the last four sections are four sections of time when the carrier wave falls. The current variation process under the action of the first four segments of zero vectors and adjacent effective vectors is detailed in fig. 5.
The following is a specific example of a surface-mount permanent magnet synchronous motor:
fig. 1 is a schematic diagram of a frame structure of a control system according to an embodiment of the present invention, in which a current sensor 3 is connected to a driving circuit of a motor 1, a position sensor 2 is disposed on the motor 1, and the current sensor 3 and the position sensor 2 are both connected to a first control module 4 through signals, so as to transmit a state parameter of the motor 1 to the first control module 4. The proportional-integral controller 7 is in signal connection with the first control module 4, can generate a quadrature axis current given value, and sends the quadrature axis current given value to the first control module 4. The first control module 4 obtains a predicted voltage value according to the following formula (1) and inputs the predicted voltage value to the second control module 5. The second control module 5 executes the second to sixth steps described below, and finally modulates and outputs the PWM pulse signal to the inverter 6.
It is understood by those skilled in the art that the first control module 4 and the second control module 5 can be integrated into one DSP chip for execution.
The first step is as follows:
the motor state parameters can be obtained by sampling mainly according to sampling modes such as a current sensor, a position sensor and the like, and the predicted voltage value u is calculated according to a discrete model of the motord pre(k)，uq pre(k) As shown in formula 1.
Figure BDA0001857475980000071
Wherein R is stator resistance, LdIs stator d-axis inductance, LqAs stator q-axis inductance, stator inductance L in surface-mounted permanent magnet synchronous motord＝Lq＝L，ωeFor the electrical angular velocity, psi, of the rotorfIs a permanent magnet flux linkage id，iqIs dq-axis current, id ref，iq refThe given value of the current is K represents the Kth control period,Tsis the sampling period.
The second step is that:
for the predicted voltage value ud pre(k)，uq pre(k) Inverse park transform is performed to obtain its α -axis component uαAnd β Axis component uβ. If u isβ>0, then a equals 1, otherwise a equals 0; if it is not
Figure BDA0001857475980000072
B is 1, otherwise B is 0; if it is not
Figure BDA0001857475980000073
Then C is 1, otherwise C is 0. Thus sector number N ═ a +2B + 4C. The order of arrangement of sector numbers in space is shown in fig. 3.
The third step:
calculating the action time of the adjacent vectors, and enabling:
Figure BDA0001857475980000081
in the formula uαIs a reference voltage vector Urefα axis component of (u)βIs a reference voltage vector Urefβ axis component Uref＝uα+uβI.e. UrefThe resultant vector of the α -axis voltage component and the β -axis voltage component may be u in dq coordinate systemd ref，uq refIs also the adjacent vector V in the figure1，V2And (4) synthesizing. U shaperefFor ABC, dq, α, β, the composite form of the components in the three coordinate systems, or the neighboring valid vectors.
Solving motor ABC three-phase voltage uA、uBAnd uCThe d and q axis components, α and β axis components, are calculated by the following specific method:
Figure BDA0001857475980000082
Figure BDA0001857475980000083
in the formula udAnd uqD and q axis components of a motor reference voltage; u. ofαAnd uβα, β axis components of motor reference voltage MABC/αβIs a transformation matrix from ABC three-phase stationary coordinate system to αβ two-phase stationary coordinate system, Mαβ/dqIs a transformation matrix from αβ two-phase stationary coordinate system to dq two-phase rotating coordinate system, and the specific expression is as follows:
Figure BDA0001857475980000084
in the formula, theta is an included angle between the d axis and the α axis;
according to the above formula (2), the adjacent vector action time T of different sectors can be obtained by looking up the table1、T2As shown in table 1.
TABLE 1 Adjacent vector Duty ratios for different sectors
Figure BDA0001857475980000091
The fourth step:
calculating the optimal zero vector action time:
in the surface-mounted permanent magnet synchronous motor, the torque is only influenced by the q-axis current, so the embodiment only takes the minimum q-axis current ripple as an optimization target. The discretization expression of the q-axis current is shown as the following formula
iq(k+1)＝iqT+Δiq(3)
Figure BDA0001857475980000092
Figure BDA0001857475980000093
As can be seen from the formulas (3) to (5), the motor TsThe q-axis current at sampling period intervals is composed of two parts, whichInqTOnly with the current state of the machine and the time interval TsIt is related. Therefore iqTWhich can be considered as an inherent component of the motor current, is an uncontrollable quantity. Voltage-influenced q-axis current change component Δ iqIs a controllable component, Δ iqRepresenting voltage-dependent components of variation, with reference to a q-axis voltage reference vector uref q(k) And a sampling period TsRegarding, as known from the principle of vector synthesis, taking the III-th sector as an example, the q-axis voltage reference vector uref q(k) From adjacent effective vectors V1，V2Are synthesized according to the duty ratio, i.e.
Figure BDA0001857475980000101
In the formula: t is1，T2For adjacent effective vectors V1，V2Duration of action, V1qAs a valid vector V1Q-axis component of (V)2qAs a valid vector V2Q-axis component of (a).
By bringing formula (6) into formula (5)
Figure BDA0001857475980000102
As can be seen from the formula (7), Δ iqIs only related to the action time of the effective vector, and Δ iqOnly during the effective vector action time T of each control cycle1，T2Within the range.
In summary, the change process of the q-axis current in one control period can be regarded as the inherent component iqTAt time TsInternal variation, and controllable component Δ iqWith V1q、V2qAt time T1、T2The change of internal action, the superposition of the above two changes. i.e. iq(k) To iqThe change process of (k +1) in one control cycle is shown in fig. 5.
The first zero vector in FIG. 5 has an action time T01The current value i after the first zero vector actionq0Can be represented by the following formula
Figure BDA0001857475980000103
iq1，iq2For the current values under the action of the adjacent effective vectors, the change trend caused by the fact that the adjacent effective vectors only differ by 60 degrees is negligible, and i is divided intoq0To iq2Are combined into the same process. i.e. iq2This can be calculated from the following equation.
Figure BDA0001857475980000104
In the formula Tc＝T01+T1+T2。
The q-axis voltage reference vector u is obtained through model calculationref q(k) Already at the optimum constant value, i.e. Δ iqConstant, in steady state conditions, to obtain a minimum current ripple should make the sum of the maximum current error and the minimum current error caused by the current ripple zero, i.e.
Figure BDA0001857475980000111
The optimal T can be obtained by substituting the formula (8) and the formula (9) into the formula (10)01To obtain
Figure BDA0001857475980000112
Figure BDA0001857475980000113
The fifth step:
calculating the switch on time point Ta、Tb、Tc。
Taking sector III as an example, T of the improved methoda、Tb、TcThe following method can be used for calculation: when the triangular carrier is in the first half cycle,
Figure BDA0001857475980000114
when the triangular carrier is in the second half period,
Figure BDA0001857475980000115
at the same time to T01Clipping is performed so that 0 ≦ T01≦Ts-T1-T2. Other sectors may also be obtained in a similar manner. T isa、Tb、TcAlso, the modulation wave amplitude when the carrier amplitude is Ts, and the timing chart thereof is shown in fig. 2.
And a sixth step:
and modulating and outputting PWM pulses through a triangular carrier wave with the period T (wherein T is 2 multiplied by Ts) and the amplitude values Ta, Tb and Tc of the modulated wave, and acting on the inverter to control the motor. And completing the control of one control cycle.
The double-sampling double-updating control strategy and the method for solving the optimal zero vector action time of the minimum current ripple can also be applied to a PI controller or other control strategies adopting SVPWM modulation.
It is noted that, in this document, relational terms such as "first" and "second," and the like, may be used solely to distinguish one entity or action from another entity or action without necessarily requiring or implying any actual such relationship or order between such entities or actions. Also, the terms "comprises," "comprising," or any other variation thereof, are intended to cover a non-exclusive inclusion, such that a process, method, article, or apparatus that comprises a list of elements does not include only those elements but may include other elements not expressly listed or inherent to such process, method, article, or apparatus. Without further limitation, an element defined by the phrase "comprising an … …" does not exclude the presence of other identical elements in a process, method, article, or apparatus that comprises the element.
The above description is merely exemplary of the present application and is presented to enable those skilled in the art to understand and practice the present application. Various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without departing from the spirit or scope of the application. Thus, the present application is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.

Claims (10)

1. A method for model-based predictive motor control, comprising:
acquiring a control command and motor parameters, and calculating a predicted voltage value according to model prediction;
performing coordinate conversion on the predicted voltage value to obtain an α axis component and a β axis component;
obtaining adjacent vector action time of different sectors according to the α axis component and the β axis component;
calculating zero vector action time with symmetric quadrature axis current errors according to the adjacent vector action time; the sum of the maximum value of the quadrature axis current error influenced by the voltage and the minimum value of the quadrature axis current error influenced by the voltage is zero; which comprises the following steps: obtaining quadrature axis voltage reference vectors with symmetric quadrature axis current errors in a model prediction mode according to the action time of adjacent vectors; calculating zero vector action time through backward estimation of a prediction model according to the quadrature axis voltage reference vector; calculating to obtain the zero vector action time T01The formula of (1) is:
Figure FDA0002394302980000011
Figure FDA0002394302980000012
in the formula, R is a stator resistor; l is stator inductance, Ld＝Lq＝L；ωeIs the rotor electrical angular velocity; psifIs a permanent magnet flux linkage; i.e. id、iqDirect axis and quadrature axis currents; i.e. id ref，iq refSetting values for direct-axis and quadrature-axis currents; k represents the Kth control period; t issIs a sampling period; t is1、T2Acting time for adjacent vectors; u. ofref q(k) For quadrature voltage reference vectors, from adjacent effective vectors V1、V2Synthesized according to the duty ratio;
calculating the switch conduction time point according to the zero vector acting time;
and modulating and outputting a PWM pulse signal to act on the control motor.
2. The method of claim 1, wherein the step of obtaining the control command and the current cycle motor parameter comprises:
within one carrier period, the duty cycle is sampled and updated at the time when the carrier is zero and the carrier maximum, respectively.
3. The model prediction-based motor control method of claim 1, wherein the modulating the output PWM pulse step comprises:
and modulating and outputting a PWM pulse signal by using the amplitude of the triangular carrier wave and the three-phase modulation wave to act on the inverter.
4. The model-prediction-based motor control method of claim 1, wherein the first four modulated signals are located for four periods when the carrier rises and the second four modulated signals are located for four periods when the carrier falls in one carrier period according to the PWM pulse signal outputted from the control method.
5. The model-prediction-based motor control method of claim 1, wherein the PWM pulse signal output according to the control method is asymmetric with respect to a carrier maximum time within one carrier period.
6. The model-prediction-based motor control method of claim 1, wherein the zero vector action time with quadrature axis current error symmetry is calculated from the adjacent vector action times, wherein: the quadrature axis current error is symmetrical relative to the quadrature axis current given value.
7. The model-prediction-based motor control method of any one of claims 1 to 6, wherein the calculating a switch on time point with the zero vector action time comprises:
the resulting formula when the carrier is in the first half cycle is:
Figure FDA0002394302980000021
the resulting formula when the carrier is in the second half of the period is:
Figure FDA0002394302980000022
at the same time to T01Clipping is performed so that 0 ≦ T01≦Ts-T1-T2；
Wherein, Ta、Tb、TcA switch on time point; t is1、T2Acting time for adjacent vectors; t is01Zero vector action time.
8. A control device of a motor, characterized by comprising:
the acquisition module is used for acquiring a control command and motor parameters;
the processing module is used for predicting and calculating a predicted voltage value according to a model, carrying out coordinate conversion on the predicted voltage value to obtain α axis components and β axis components, obtaining adjacent vector action time of different sectors according to the α axis components and the β axis components, and calculating to obtain a zero vector with symmetrical quadrature axis current errors according to the adjacent vector action timeThe acting time; calculating the switch conduction time point according to the zero vector acting time; the sum of the maximum value of the quadrature axis current error influenced by the voltage and the minimum value of the quadrature axis current error influenced by the voltage is zero; which comprises the following steps: obtaining quadrature axis voltage reference vectors with symmetric quadrature axis current errors in a model prediction mode according to the action time of adjacent vectors; calculating zero vector action time through backward estimation of a prediction model according to the quadrature axis voltage reference vector; calculating to obtain the zero vector action time T01The formula of (1) is:
Figure FDA0002394302980000031
Figure FDA0002394302980000032
in the formula, R is a stator resistor; l is stator inductance, Ld＝Lq＝L；ωeIs the rotor electrical angular velocity; psifIs a permanent magnet flux linkage; i.e. id、iqDirect axis and quadrature axis currents; i.e. id ref，iq refSetting values for direct-axis and quadrature-axis currents; k represents the Kth control period; t issIs a sampling period; t is1、T2Acting time for adjacent vectors; u. ofref q(k) For quadrature voltage reference vectors, from adjacent effective vectors V1、V2Synthesized according to the duty ratio;
and the generating module is used for generating a control signal of the motor according to the switch conducting time point.
9. The control device of claim 8, further comprising a determining module for monitoring a carrier zero time and a carrier maximum time within a carrier period, and triggering the control device to sample and update the duty cycle at both times.
10. A control device according to claim 8 or 9, characterized in that the control device is a PI controller, a model predictive controller or a PWM controller.
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