CN110535389B - Oversampling prediction current control method for permanent magnet synchronous motor system - Google Patents

Oversampling prediction current control method for permanent magnet synchronous motor system Download PDF

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CN110535389B
CN110535389B CN201910845262.0A CN201910845262A CN110535389B CN 110535389 B CN110535389 B CN 110535389B CN 201910845262 A CN201910845262 A CN 201910845262A CN 110535389 B CN110535389 B CN 110535389B
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moment
motor
pwm duty
current
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CN110535389A (en
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王志强
杨明波
夏长亮
谢赛飞
金雪峰
张国政
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Tianjin Polytechnic University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/18—Estimation of position or speed
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop

Abstract

An oversampling prediction current control method for a permanent magnet synchronous motor system comprises the following steps: by control system at kTs、(k+0.25)Ts、(k+0.5)Ts、(k+0.75)TsSampling at all times; calculating a q-axis component of the motor reference current; at kTs、(k+0.5)TsSolving d and q axis components of the actual current of the motor at any moment; obtaining d and q axis components of the predicted voltage by using a motor discrete prediction model; adopts an asymmetric seven-segment two-level SVPWM modulation strategy at kTs、(k+0.5)TsThe duty ratio of six PWM pulses is calculated at the time point, and is at (k +0.25) Ts、(k+0.5)Ts、(k+0.75)Ts、(k+1)TsAnd outputting six paths of PWM pulses at any moment to act on the six-bridge arm inverter, and further actually outputting corresponding reference voltage to act on the motor. According to the invention, through twice voltage and current sampling, four times of motor rotor position angle sampling and four times of PWM duty ratio updating in one carrier cycle, the dynamic performance of the system under low switching frequency is effectively improved, and no static error and oscillation in a steady state exist.

Description

Oversampling prediction current control method for permanent magnet synchronous motor system
Technical Field
The invention relates to a permanent magnet synchronous motor. In particular to an oversampling prediction current control method for a permanent magnet synchronous motor system.
Background
Model Predictive Control (MPC) was originally developed for engineering applications, and has a strong background and wide applicability for industrial applications. The control method is successfully applied to a plurality of process control fields of petroleum, chemical engineering, aerospace, energy and the like. The permanent magnet synchronous motor control system mostly adopts a rotating speed and current double closed loop control structure, wherein the dynamic and steady state performance of a current inner loop is a key factor for improving the performance of the permanent magnet synchronous motor control system. The model predictive control predicts a voltage vector to be applied to the motor at the time k +1 by a predictive model using the motor state at the time k. After the voltage vector acts for one period, the motor current can accurately follow the command current value. The model predictive control enables good dynamic and steady state response of the motor current. However, for digital control, control delay is one of the main factors for restricting the loop-in-current and steady-state performance. Under the working condition of low switching frequency, the PWM duty ratio updating delay period is longer, at the moment, the actual values of the voltage, the current and the rotor position angle of the motor and the sampling value change greatly, so that a large error exists in the control quantity, the current control can generate oscillation and static error, the current oscillation can cause the mechanical vibration of the motor, and even can cause the driver to stop running due to overcurrent alarm; the current static difference can reduce the operation efficiency of the driving system, so that the driving system can not output rated torque and can not work in a torque control mode, and the like.
Disclosure of Invention
The technical problem to be solved by the invention is to provide a permanent magnet synchronous motor system oversampling and current predicting control method which can greatly improve the dynamic and steady state performance of a permanent magnet synchronous motor control system.
The technical scheme adopted by the invention is as follows: an oversampling prediction current control method for a permanent magnet synchronous motor system comprises the following steps:
1) at kTsTime sum (k +0.5) TsAt the moment, the control system samples the three-phase current of the motor ABC, the voltage of a direct-current bus and the electrical angular speed of a motor rotor; at kTsTime, (k +0.25) TsTime, (k +0.5) TsTime, (k +0.75) TsAt that time, the control system samples the rotor position angle, k 1, 2, 3 … …; t issIs IGA BT switching period;
2) under the control that the d shafting component of the motor reference current is zero, the q shafting component of the motor reference current is calculated through a rotating speed ring PI regulator
Figure GDA0002821675940000011
The method specifically comprises the following steps:
Figure GDA0002821675940000012
wherein the content of the first and second substances,
Figure GDA0002821675940000013
respectively are motor reference current d and q shafting components,
Figure GDA0002821675940000014
is the proportional coefficient of the rotating speed ring PI regulator,
Figure GDA0002821675940000015
for the integral coefficient, omega, of a speed loop PI regulatorrefIs a reference value of the rotating speed, and omega is the mechanical angular speed of the motor rotor;
3) solving for (k + x) T according to motor ABC three-phase currentsD and q shafting components i of actual current of time motord(k+x)、iq(k + x), specifically solved as:
Figure GDA0002821675940000021
where x is 0 in the first half and 0.5 in the second half of each carrier period, id(k + x) and iq(k + x) are respectively d and q shafting components of the actual current of the motor, iA(k)、iB(k) And iC(k) Is ABC three-phase current of the motor, MABC/αβIs a transformation matrix from ABC three-phase stationary shafting to alpha beta two-phase stationary shafting, Mαβ/dqThe specific expression is a transformation matrix from an alpha beta two-phase stationary shafting to a dq two-phase rotating shafting as follows:
Figure GDA0002821675940000022
Figure GDA0002821675940000023
in the formula, θ (k + x) is (k + x) TsThe included angle between the d axis system and the alpha axis system at the moment;
4) at kTsTime sum (k +0.5) TsAt the moment, the predicted values of the d and q shafting components of the actual current of the motor are obtained according to the current prediction model, including respectively predicting (k +0.25) TsTime sum (k +0.75) TsTime current d, q axis component
Figure GDA0002821675940000024
And
Figure GDA0002821675940000025
and
Figure GDA0002821675940000026
and
Figure GDA0002821675940000027
delay compensation as a voltage prediction model;
5) using a voltage prediction model, based on kT respectivelysTime sum (k +0.5) TsThe electric angular speed of the motor rotor at the moment, and components of a motor reference current d shafting and a q shafting
Figure GDA0002821675940000028
And the predicted values of the components of the d axis system and the q axis system of the actual current of the motor are obtained, so that the predicted current is (k +1) TsPredicted voltage d and q shafting components of time tracking reference current
Figure GDA0002821675940000029
And
Figure GDA00028216759400000210
at (k +1.5) TsPredicted voltage d and q shafting components of time tracking reference current
Figure GDA00028216759400000211
And
Figure GDA00028216759400000212
6) in each carrier period, an asymmetric seven-segment two-level SVPWM modulation method is adopted to calculate the PWM duty ratio T of the four-time three-phase invertera、TbAnd TcUpdating the calculation result of each time; at kTsTime judgment (k-0.25) TsPWM duty ratio calculated at the moment of kTsTime to (k +0.25) TsThere are several crossing points with the triangular carrier between the moments, if the number of crossing points is greater than 1, kTsThe three-phase PWM duty ratio calculated at the moment is equal to (k-0.25) TsThe three-phase PWM duty ratio at the moment, if the number of the intersection points is equal to 1, according to kTsReference voltage d and q shafting components calculated at moment
Figure GDA00028216759400000213
And rotor position angle θ (k) and at kTsTime to (k +0.25) TsOne-phase PWM duty ratio with intersection point between the moment and the triangular carrier wave, recalculating PWM duty ratios of other two phases, and if no intersection point exists, performing pulse width modulation (kT) according to the calculated PWM duty ratiosReference voltage d and q shafting components calculated at moment
Figure GDA00028216759400000214
And a rotor position angle θ (k) calculating a three-phase PWM duty cycle;
at (k +0.25) TsTime of day in terms of kTsReference voltage d and q shafting components calculated at moment
Figure GDA00028216759400000215
And (k +0.25) TsCalculating a three-phase PWM duty ratio according to a rotor position angle theta (k +0.25) at the moment;
at (k +0.5) TsTime judgment (k +0.25) TsThe three-phase PWM duty ratio calculated at the moment is (k +0.5) TsTime to (k +0.75) TsThere are several crossing points between the time and the triangular carrier, if the crossing point number is more than 1, (k +0.5) TsThe three-phase PWM duty ratio calculated at the moment is equal to (k +0.25) TsThe three-phase PWM duty ratio at the moment, if the number of the intersection points is equal to 1, according to (k +0.5) TsReference voltage d and q shafting components calculated at moment
Figure GDA0002821675940000031
And rotor position angle θ (k +0.5) and at (k +0.5) TsTime to (k +0.75) TsOne phase PWM duty ratio of the intersection point of the moment and the triangular carrier wave is recalculated, if the intersection point does not exist, the PWM duty ratio of the other two phases is recalculated according to (k +0.5) TsReference voltage d and q shafting components calculated at moment
Figure GDA0002821675940000032
And rotor position angle θ (k +0.5), calculating three-phase PWM duty cycle;
at (k +0.75) TsThe time is according to (k +0.5) TsReference voltage d and q shafting components calculated at moment
Figure GDA0002821675940000033
Figure GDA0002821675940000034
And (k +0.75) TsCalculating a three-phase PWM duty ratio according to a rotor position angle theta (k +0.75) at the moment; delaying the three-phase PWM duty ratio calculated at each moment by 0.25TsThen comparing the output voltage with the triangular carrier and outputting PWM pulse to act on the six-bridge arm inverter, further actually outputting corresponding reference voltage to act on the motor, and returning to the step 1) for circulation.
The oversampling prediction current control method for the permanent magnet synchronous motor system can greatly improve the dynamic and stable performance of the permanent magnet synchronous motor control system under the condition of keeping the switching frequency unchanged. The technical scheme of the invention has the following beneficial effects:
(1) according to the invention, the bus voltage and the three-phase current of the motor are sampled twice in one period to sample the position angle of the motor rotor four times, and the d shafting component and the q shafting component of the predicted voltage value are obtained through a model prediction algorithm, so that the steady-state performance of the system is guaranteed;
(2) according to the invention, the four PWM duty ratios are updated in one carrier cycle by an asymmetric SVPWM (space vector pulse width modulation) method, so that a reference basis is provided for improving the dynamic performance of a system;
(3) according to the invention, through twice voltage and current sampling, four times of motor rotor position angle sampling and four times of PWM duty ratio updating in one carrier cycle, the dynamic performance of the system under low switching frequency is effectively improved, and no static error and oscillation in a steady state exist.
Drawings
FIG. 1 is a diagram of a main circuit and control system of a three-phase two-level PWM rectifier;
FIG. 2 is a timing diagram of current sampling and PWM duty cycle update for a PMSM control system;
fig. 3 is a flowchart of an oversampling predictive current control method for a permanent magnet synchronous motor system according to the present invention.
Detailed Description
The oversampling prediction current control method of the permanent magnet synchronous motor system according to the present invention is described in detail below with reference to the embodiments and the drawings.
As shown in fig. 3, the oversampling prediction current control method for the permanent magnet synchronous motor system of the present invention includes the following steps:
1) at kTsTime, (k +0.5) TsAt any moment, the control system samples current, voltage and electric angular speed of the motor rotor, and the method specifically comprises the following steps: motor ABC three-phase current iA(k)、iA(k+0.5)、iB(k)、iB(k+0.5)、iC(k)、iC(k +0.5), DC bus voltage udc(k)、udc(k +1), electrical angular velocity ω of rotor of electric machinee(k)、ωe(k + 0.5); at kTsTime, (k +0.25) TsMoment (k +0.5)TsTime, (k +0.75) TsAt any moment, the control system samples the rotor position angle, and the method specifically comprises the following steps: theta (k), theta (k +0.25), theta (k +0.5) and theta (k + 0.75); k is 1, 2, 3 … …; t issIs the IGBT switching period;
2) under the control that the d shafting component of the motor reference current is zero, the q shafting component of the motor reference current is calculated through a rotating speed ring PI regulator
Figure GDA0002821675940000041
The method specifically comprises the following steps:
Figure GDA0002821675940000042
wherein the content of the first and second substances,
Figure GDA0002821675940000043
respectively are motor reference current d and q shafting components,
Figure GDA0002821675940000044
is the proportional coefficient of the rotating speed ring PI regulator,
Figure GDA0002821675940000045
for the integral coefficient, omega, of a speed loop PI regulatorrefIs a reference value of the rotating speed, and omega is the mechanical angular speed of the motor rotor;
3) according to motor ABC three-phase current iA(k+x)、iB(k+x)、iC(k + x), solving for (k + x) TsD and q shafting components i of actual current of time motord(k+x)、iq(k + x), specifically solved as:
Figure GDA0002821675940000046
where x is 0 in the first half and 0.5 in the second half of each carrier period, id(k + x) and iq(k + x) are d-axis and q-axis components of the actual current of the motor, iA(k)、iB(k) AndiC(k) is ABC three-phase current of the motor, MABC/αβIs a transformation matrix from ABC three-phase stationary shafting to alpha beta two-phase stationary shafting, Mαβ/dqThe specific expression is a transformation matrix from an alpha beta two-phase stationary shafting to a dq two-phase rotating shafting as follows:
Figure GDA0002821675940000047
Figure GDA0002821675940000048
in the formula, θ (k + x) is (k + x) TsThe included angle between the d axis system and the alpha axis system at the moment;
4) at kTsTime sum (k +0.5) TsAt the moment, the predicted values of the d and q shafting components of the actual current of the motor are obtained according to the current prediction model, including respectively predicting (k +0.25) TsTime sum (k +0.75) TsTime current d, q axis component
Figure GDA0002821675940000049
And
Figure GDA00028216759400000410
and
Figure GDA00028216759400000411
and
Figure GDA00028216759400000412
delay compensation as a voltage prediction model; the current prediction model is as follows:
Figure GDA0002821675940000051
in the formula (I), the compound is shown in the specification,
Figure GDA0002821675940000052
is (k-1) TsD and q shafting voltage predicted values obtained by time calculation,
Figure GDA0002821675940000053
Figure GDA0002821675940000054
is (k-0.5) TsD and q shafting voltage predicted values, T, obtained by time calculationsFor IGBT switching period, RsIs stator resistance, Ld、LqD, q axial components, psi, of stator inductance, respectivelyfFor rotor flux linkage, omegae(k) The electrical angular velocity of the motor rotor at time k.
5) Using a voltage prediction model, based on kT respectivelysTime sum (k +0.5) TsThe electric angular speed of the motor rotor at the moment, and components of a motor reference current d shafting and a q shafting
Figure GDA0002821675940000055
And the predicted values of the components of the d axis system and the q axis system of the actual current of the motor are obtained, so that the predicted current is (k +1) TsPredicted voltage d and q shafting components of time tracking reference current
Figure GDA0002821675940000056
And
Figure GDA0002821675940000057
at (k +1.5) TsPredicted voltage d and q shafting components of time tracking reference current
Figure GDA0002821675940000058
And
Figure GDA0002821675940000059
the voltage prediction model is as follows:
Figure GDA0002821675940000061
in the formula (I), the compound is shown in the specification,
Figure GDA0002821675940000062
respectively, the predicted voltage d and q are axial components, the upper mark PR represents the predicted value, TsFor IGBT switching period, RsIs stator resistance, Ld、LqD, q axial components, psi, of stator inductance, respectivelyfFor rotor flux linkage, omegaeIs the electrical angular velocity, omega, of the rotor of the motore(k) At time k, the electrical angular velocity, omega, of the motor rotore(k +0.5) the electrical angular velocity of the rotor of the motor at the moment k +0.5,
Figure GDA0002821675940000063
And the current q shafting component is used as the delay compensation of the voltage prediction model.
6) In each carrier period, an asymmetric seven-segment two-level SVPWM modulation method is adopted to calculate the PWM duty ratio T of the four-time three-phase invertera、TbAnd TcUpdating the calculation result of each time; at kTsTime judgment (k-0.25) TsPWM duty ratio calculated at the moment of kTsTime to (k +0.25) TsThere are several crossing points with the triangular carrier between the moments, if the number of crossing points is greater than 1, kTsThe three-phase PWM duty ratio calculated at the moment is equal to (k-0.25) TsThe three-phase PWM duty ratio at the moment, if the number of the intersection points is equal to 1, according to kTsReference voltage d and q shafting components calculated at moment
Figure GDA0002821675940000064
And rotor position angle θ (k) and at kTsTime to (k +0.25) TsOne-phase PWM duty ratio with intersection point between the moment and the triangular carrier wave, recalculating PWM duty ratios of other two phases, and if no intersection point exists, performing pulse width modulation (kT) according to the calculated PWM duty ratiosReference voltage d and q shafting components calculated at moment
Figure GDA0002821675940000065
And a rotor position angle θ (k) calculating a three-phase PWM duty cycle;
at (k +0.25) TsTime of day in terms of kTsTime of dayCalculated reference voltage d, q axis component
Figure GDA0002821675940000066
And (k +0.25) TsCalculating a three-phase PWM duty ratio according to a rotor position angle theta (k +0.25) at the moment;
at (k +0.5) TsTime judgment (k +0.25) TsThe three-phase PWM duty ratio calculated at the moment is (k +0.5) TsTime to (k +0.75) TsThere are several crossing points between the time and the triangular carrier, if the crossing point number is more than 1, (k +0.5) TsThe three-phase PWM duty ratio calculated at the moment is equal to (k +0.25) TsThe three-phase PWM duty ratio at the moment, if the number of the intersection points is equal to 1, according to (k +0.5) TsReference voltage d and q shafting components calculated at moment
Figure GDA0002821675940000067
And rotor position angle θ (k +0.5) and at (k +0.5) TsTime to (k +0.75) TsOne phase PWM duty ratio of the intersection point of the moment and the triangular carrier wave is recalculated, if the intersection point does not exist, the PWM duty ratio of the other two phases is recalculated according to (k +0.5) TsReference voltage d and q shafting components calculated at moment
Figure GDA0002821675940000068
And rotor position angle θ (k +0.5), calculating three-phase PWM duty cycle;
at (k +0.75) TsThe time is according to (k +0.5) TsReference voltage d and q shafting components calculated at moment
Figure GDA0002821675940000071
Figure GDA0002821675940000072
And (k +0.75) TsCalculating a three-phase PWM duty ratio according to a rotor position angle theta (k +0.75) at the moment; delaying the three-phase PWM duty ratio calculated at each moment by 0.25TsThen comparing the voltage with the triangular carrier and outputting PWM pulse to act on the six-bridge arm inverter so as to actually output corresponding reference voltage to act on the motor and returning to the stepStep 1) circulation is carried out. Wherein:
at kTsTime judgment (k-0.25) TsThe three-phase PWM duty ratio calculated at the moment is kTsThe calculation formula for recalculating the PWM duty ratios of the other two phases when the number of intersections between the time and the triangular carrier wave is equal to 1 is as follows:
Figure GDA0002821675940000073
in the formula, T1Is kTsTime to (k +0.25) TsTime, T, at which the updated one-phase PWM duty cycle intersects the triangular carrier between times2、T3The other two-phase PWM duty ratio for recalculation is at (k +0.25) TsTime to (k +0.5) TsThe time between the time instants at which the triangular carriers intersect,
Figure GDA0002821675940000074
theta (k +0.25) is (k +0.25) T for SVPWM modulation coefficientsThe electric angle of the motor rotor is calculated at any moment;
at kTsTime judgment (k +0.25) TsThe three-phase PWM duty ratio calculated at the moment is (k +0.5) TsTime to (k +0.75) TsThe calculation formula for recalculating the PWM duty ratios of the other two phases when the number of intersections between the time and the triangular carrier wave is equal to 1 is as follows:
Figure GDA0002821675940000075
in the formula, T4Is (k +0.5) TsTime to (k +0.75) TsTime, T, at which the updated one-phase PWM duty cycle intersects the triangular carrier between times5、T6The other two-phase PWM duty cycle for recalculation is at (k +0.75) TsTime to (k +1) TsThe time between the time instants at which the triangular carriers intersect,
Figure GDA0002821675940000076
for SVPWM modulationCoefficient, θ (k +0.75) is (k +0.75) TsTime of day electrical angle of rotor, TsFor IGBT switching cycles, UrefFor reference to the DC bus voltage value, UdcThe actual DC bus voltage value is obtained.

Claims (4)

1. An oversampling prediction current control method for a permanent magnet synchronous motor system is characterized by comprising the following steps:
1) at kTsTime sum (k +0.5) TsAt the moment, the control system samples the three-phase current of the motor ABC, the voltage of a direct-current bus and the electrical angular speed of a motor rotor; at kTsTime, (k +0.25) TsTime, (k +0.5) TsTime, (k +0.75) TsAt that time, the control system samples the rotor position angle, k 1, 2, 3 … …; t issIs the IGBT switching period;
2) under the control that the d shafting component of the motor reference current is zero, the q shafting component of the motor reference current is calculated through a rotating speed ring PI regulator
Figure FDA0002882445790000011
The method specifically comprises the following steps:
Figure FDA0002882445790000012
wherein the content of the first and second substances,
Figure FDA0002882445790000013
respectively are motor reference current d and q shafting components,
Figure FDA0002882445790000014
is the proportional coefficient of the rotating speed ring PI regulator,
Figure FDA0002882445790000015
for the integral coefficient, omega, of a speed loop PI regulatorrefIs a reference value of the rotating speed, and omega is the mechanical angular speed of the motor rotor;
3) solving for (k + x) T according to motor ABC three-phase currentsD and q shafting components i of actual current of time motord(k+x)、iq(k + x), specifically solved as:
Figure FDA0002882445790000016
where x is 0 in the first half and 0.5 in the second half of each carrier period, id(k + x) and iq(k + x) are respectively d and q shafting components of the actual current of the motor, iA(k)、iB(k) And iC(k) Is ABC three-phase current of the motor, MABC/αβIs a transformation matrix from ABC three-phase stationary shafting to alpha beta two-phase stationary shafting, Mαβ/dqThe specific expression is a transformation matrix from an alpha beta two-phase stationary shafting to a dq two-phase rotating shafting as follows:
Figure FDA0002882445790000017
Figure FDA0002882445790000018
in the formula, θ (k + x) is (k + x) TsThe included angle between the d axis system and the alpha axis system at the moment;
4) at kTsTime sum (k +0.5) TsAt the moment, the predicted values of the d and q shafting components of the actual current of the motor are obtained according to the current prediction model, including respectively predicting (k +0.25) TsTime sum (k +0.75) TsTime current d, q axis component
Figure FDA0002882445790000019
And
Figure FDA00028824457900000110
and
Figure FDA00028824457900000111
and
Figure FDA00028824457900000112
delay compensation as a voltage prediction model;
5) using a voltage prediction model, based on kT respectivelysTime sum (k +0.5) TsThe electric angular speed of the motor rotor at the moment, and components of a motor reference current d shafting and a q shafting
Figure FDA00028824457900000113
And the predicted values of the components of the d axis system and the q axis system of the actual current of the motor are obtained, so that the predicted current is (k +1) TsPredicted voltage d and q shafting components of time tracking reference current
Figure FDA0002882445790000021
And
Figure FDA0002882445790000022
at (k +1.5) TsPredicted voltage d and q shafting components of time tracking reference current
Figure FDA0002882445790000023
And
Figure FDA0002882445790000024
6) in each carrier period, an asymmetric seven-segment two-level SVPWM modulation method is adopted to calculate the PWM duty ratio T of the four-time three-phase invertera、TbAnd TcUpdating the calculation result of each time; at kTsTime judgment (k-0.25) TsPWM duty ratio calculated at the moment of kTsTime to (k +0.25) TsThere are several crossing points with the triangular carrier between the moments, if the number of crossing points is greater than 1, kTsThe three-phase PWM duty ratio calculated at the moment is equal to (k-0.25) TsThe three-phase PWM duty ratio at the moment, if the number of the intersection points is equal to 1According to kTsReference voltage d and q shafting components calculated at moment
Figure FDA0002882445790000025
And rotor position angle θ (k) and at kTsTime to (k +0.25) TsOne-phase PWM duty ratio with intersection point between the moment and the triangular carrier wave, recalculating PWM duty ratios of other two phases, and if no intersection point exists, performing pulse width modulation (kT) according to the calculated PWM duty ratiosReference voltage d and q shafting components calculated at moment
Figure FDA0002882445790000026
And a rotor position angle θ (k) calculating a three-phase PWM duty cycle;
at (k +0.25) TsTime of day in terms of kTsReference voltage d and q shafting components calculated at moment
Figure FDA0002882445790000027
And (k +0.25) TsCalculating a three-phase PWM duty ratio according to a rotor position angle theta (k +0.25) at the moment;
at (k +0.5) TsTime judgment (k +0.25) TsThe three-phase PWM duty ratio calculated at the moment is (k +0.5) TsTime to (k +0.75) TsThere are several crossing points between the time and the triangular carrier, if the crossing point number is more than 1, (k +0.5) TsThe three-phase PWM duty ratio calculated at the moment is equal to (k +0.25) TsThe three-phase PWM duty ratio at the moment, if the number of the intersection points is equal to 1, according to (k +0.5) TsReference voltage d and q shafting components calculated at moment
Figure FDA0002882445790000028
And rotor position angle θ (k +0.5) and at (k +0.5) TsTime to (k +0.75) TsOne phase PWM duty ratio of the intersection point of the moment and the triangular carrier wave is recalculated, if the intersection point does not exist, the PWM duty ratio of the other two phases is recalculated according to (k +0.5) TsReference voltage d and q shafting components calculated at moment
Figure FDA0002882445790000029
And rotor position angle θ (k +0.5), calculating three-phase PWM duty cycle;
at (k +0.75) TsThe time is according to (k +0.5) TsReference voltage d and q shafting components calculated at moment
Figure FDA00028824457900000210
Figure FDA00028824457900000211
And (k +0.75) TsCalculating a three-phase PWM duty ratio according to a rotor position angle theta (k +0.75) at the moment; delaying the three-phase PWM duty ratio calculated at each moment by 0.25TsThen comparing the output voltage with the triangular carrier and outputting PWM pulse to act on the six-bridge arm inverter, further actually outputting corresponding reference voltage to act on the motor, and returning to the step 1) for circulation.
2. The method for controlling the oversampling and predicting current of the permanent magnet synchronous motor system according to claim 1, wherein the current prediction model of the step 4) is as follows:
Figure FDA0002882445790000031
Figure FDA0002882445790000032
in the formula (I), the compound is shown in the specification,
Figure FDA0002882445790000033
is (k-1) TsD and q shafting voltage predicted values obtained by time calculation,
Figure FDA0002882445790000034
Figure FDA0002882445790000035
is (k-0.5) TsD and q shafting voltage predicted values, T, obtained by time calculationsFor IGBT switching period, RsIs stator resistance, Ld、LqD, q axial components, psi, of stator inductance, respectivelyfFor rotor flux linkage, omegae(k) The electrical angular velocity of the motor rotor at time k.
3. The method for controlling the oversampling predictive current of the permanent magnet synchronous motor system according to claim 1, wherein the voltage prediction model of step 5) is as follows:
Figure FDA0002882445790000036
Figure FDA0002882445790000037
in the formula (I), the compound is shown in the specification,
Figure FDA0002882445790000041
respectively, the predicted voltage d and q are axial components, the upper mark PR represents the predicted value, TsFor IGBT switching period, RsIs stator resistance, Ld、LqD, q axial components, psi, of stator inductance, respectivelyfFor rotor flux linkage, omegaeIs the electrical angular velocity, omega, of the rotor of the motore(k) At time k, the electrical angular velocity, omega, of the motor rotore(k +0.5) the electrical angular velocity of the rotor of the motor at the moment k +0.5,
Figure FDA0002882445790000042
And the current q shafting component is used as the delay compensation of the voltage prediction model.
4. The oversampling predictive current control method for a permanent magnet synchronous motor system according to claim 1, wherein in step 6):
at kTsTime judgment (k-0.25) TsThe three-phase PWM duty ratio calculated at the moment is kTsTime to (k +0.25) TsThe calculation formula for recalculating the PWM duty ratios of the other two phases when the number of intersections between the time and the triangular carrier wave is equal to 1 is as follows:
Figure FDA0002882445790000043
T3＝0.5mTs sinθ(k+0.25)+T2+T1
in the formula, T1Is kTsTime to (k +0.25) TsTime, T, at which the updated one-phase PWM duty cycle intersects the triangular carrier between times2、T3The other two-phase PWM duty ratio for recalculation is at (k +0.25) TsTime to (k +0.5) TsThe time between the time instants at which the triangular carriers intersect,
Figure FDA0002882445790000044
theta (k +0.25) is (k +0.25) T for SVPWM modulation coefficientsThe electric angle of the motor rotor is calculated at any moment;
at kTsTime judgment (k +0.25) TsThe three-phase PWM duty ratio calculated at the moment is (k +0.5) TsTime to (k +0.75) TsThe calculation formula for recalculating the PWM duty ratios of the other two phases when the number of intersections between the time and the triangular carrier wave is equal to 1 is as follows:
Figure FDA0002882445790000045
T6＝T4-0.5mTssinθ(k+0.75)-T5
in the formula, T4Is (k +0.5) TsTime to (k +0.75) TsTime, T, at which the updated one-phase PWM duty cycle intersects the triangular carrier between times5、T6The other two-phase PWM duty cycle for recalculation is at (k +0.75) TsTime to (k +1) TsThe time between the time instants at which the triangular carriers intersect,
Figure FDA0002882445790000046
theta (k +0.75) is (k +0.75) T, which is the SVPWM modulation coefficientsTime of day electrical angle of rotor, TsFor IGBT switching cycles, UrefFor reference to the DC bus voltage value, UdcThe actual DC bus voltage value is obtained.
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