CN108631600B - Double-active-bridge converter minimum reflux power double-direction internal phase-shifting control method - Google Patents
Double-active-bridge converter minimum reflux power double-direction internal phase-shifting control method Download PDFInfo
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of dc power input into dc power output
- H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
- H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
- H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
- H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M3/3353—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having at least two simultaneously operating switches on the input side, e.g. "double forward" or "double (switched) flyback" converter
Abstract
The invention discloses a double-directional internal phase-shifting control method for minimum reflux power of a double-active-bridge converter. Under the condition of constant power transmission, compared with dual phase-shift modulation, dual bidirectional internal phase-shift modulation has the advantage of inhibiting the influence of external phase-shift duty ratio disturbance on internal phase-shift duty ratio, so that a control system is more stable, and the transmission power is more stable. The method provides a minimum reflux power control method based on dual bidirectional inner phase shift modulation. The method can find the optimal operating point of the reflux power by automatic adjustment when different powers are output, thereby greatly reducing the reflux power and effectively improving the efficiency of the converter. Therefore, compared with the dual phase-shift modulation, the method is more suitable for being applied to a system with a dual-active bridge converter as an energy transmission unit, especially for the occasions with high requirements on control stability. Has wide prospect and easy popularization.
Description
Technical Field
The invention belongs to the energy transmission technology of a bidirectional direct current converter in the field of electrical engineering, and particularly relates to a double-bidirectional internal phase-shifting control method for minimum reflux power of a double-active-bridge converter.
Background
The Dual Active Bridge (DAB) converter has the characteristics of bidirectional power flow, high power density, low cost, high reliability and high efficiency. Are increasingly used in high power energy transmission. Common modulation methods include traditional phase shift, extended phase shift, dual phase shift, and triple phase shift. Aiming at different modulation methods, the problem of backflow power exists in the energy transmission process, namely when the AC output voltage U of the primary side H bridgeabAnd the inductor current iLWhen the direction is opposite, the energy stored in the inductor flows back to the input voltage U1On the side, the power of this portion is the return power Q.
The return power Q directly affects the transmission efficiency of the dual active bridge converter. Therefore, there is increasing interest in how to achieve minimum backflow power operation. The method not only has deep theoretical analysis on the method by academic papers, but also has an engineering method for practical application. Currently, the optimization for the DAB reflux power is mainly for different modulation methods. For example, the invention and patent application "a method for reducing backflow power of a DC-DC converter with a buffer" (CN 107911028A) and the invention and patent application "minimum backflow power phase-shift control method for an isolated bidirectional DC converter" (CN 106981992A) are filed.
The invention discloses a DC-DC converter with a buffer and a method for reducing reflux power in 2018, 4.13.A of Chinese patent application publication No. CN 107911028A, which is directed to a method for reducing reflux power under the condition of expanding phase shift modulation, and has the following disadvantages:
1. although the optimized control of the reflux power is realized, the expanded phase shift modulation method is adopted, the output waveform of the primary side H bridge is three levels, and the output waveform of the secondary side H bridge is two levels, namely the original secondary side H bridge has different voltage conversion states, and the control system is not easy to realize.
2. The expanded phase-shift modulation introduces an internal phase-shift angle only in the H bridge on one side of the transformer, and has the same degree of freedom as the control of the dual phase-shift modulation, but compared with the dual phase-shift modulation, the expanded phase-shift modulation has asymmetry and poorer dynamic performance.
The invention discloses a minimum reflux power phase-shift control method of an isolated bidirectional direct-current converter, which is disclosed by Chinese patent application publication CN 106981992A in 25.7.2017, is a method for reducing reflux power under dual phase-shift modulation, and has the following defects:
1. although the optimal control of the reflux power is realized, the modulation method shifts the phase duty ratio D outwards when the constant power is transmitted2Slight variations will produce an internally shifted duty cycle D1Large variations result in instability of the control system. Therefore, this modulation method is not suitable for the case where the stability is required to be high.
2. By adopting dual phase-shift modulation, the small change of the duty ratio of the external phase shift can generate large surge current under light load, and the IGBT of the switching tube can be damaged under severe conditions.
Entitled "Novel Dual Phase-Shift Control With Bidirectional Current and Stable Power Control", X.Liu, Senior Member, IEEE, Z.Q.Zhu, Fellow, IEEE, David A.Stone, Martin P.Foster, W.Q.Chu, Iain Urqhart, and James Greenough, & Power Electronics, IEEE Transactions on, vol.32, No.5pp.4095-5106, 2017 ("Dual Bidirectional Bridge Converter With Low Current and Stable Power Control"; IEEE Phase Shift telegraphy-Electronics, journal of the same year 32. No.5 pp.2015-5106 "), but the issue of Power output stabilization is not addressed in the aforementioned Dual Phase-Shift Control System — constant Power output System — No. 4096.
A typical application of the dual-active-bridge converter is an energy router, which is used as an energy transmission unit to perform bidirectional energy transmission and stabilize a dc bus voltage in a system. For such applications, the stability of the dual active bridge converter control system appears to be of paramount importance. The reduction of the reflux power plays an important role in improving the system efficiency. Therefore, designing the minimum reflux power control meeting the requirement of dual bi-directional internal phase shift modulation has important significance for improving the system efficiency. The method solves the problem of how to realize the minimum reflux power operation when outputting different powers based on dual bidirectional inner phase shift modulation.
Disclosure of Invention
The invention aims to further solve the problem of realizing the minimum reflux power operation when different powers are output on the basis of the existing dual-bi-directional internal phase-shifting modulation method, thereby greatly reducing the reflux power of the dual-active-bridge converter and realizing that the theoretical value of the reflux power is zero in a certain power range.
In order to solve the technical problem of the invention, the adopted technical scheme is as follows: a double-active-bridge converter minimum reflux power double-bidirectional internal phase-shifting control method relates to a topological structure of a double-active-bridge converter, which comprises a direct-current voltage source and an input capacitor CiA primary side H bridge, a phase-shifting inductor L, a high-frequency transformer, a secondary side H bridge and an output capacitor CoAnd a load resistor R; the primary side H-bridge consists of four switching tube IGBTs, four anti-parallel diodes and four parasitic capacitors, wherein the four switching tube IGBTSs are respectively recorded as switching tubes IGBTSs1IGBTS (insulated Gate Bipolar translator) switching tube2IGBTS (insulated Gate Bipolar translator) switching tube3IGBTS (insulated Gate Bipolar translator) switching tube4Four anti-parallel diodes are respectively marked as D1、D2、D3、D4And four parasitic capacitances are respectively denoted as C1、C2、C3、C4(ii) a The secondary side H bridge consists of four switching tube IGBTs, four anti-parallel diodes and four parasitic capacitors, wherein the four switching tube IGBTS are respectively recorded as switching tubes IGBTS5IGBTS (insulated Gate Bipolar translator) switching tube6IGBTS (insulated Gate Bipolar translator) switching tube7IGBTS (insulated Gate Bipolar translator) switching tube8Four anti-parallel diodes are respectively marked as D5、D6、D7、D8And four parasitic capacitances are respectively denoted as C5、C6、C7、C8；
The DC voltage source and the input capacitor CiAfter being connected in parallel, the DC input end of the primary side H bridge is connected, the AC output end a point of the primary side H bridge is connected to one end of a phase-shifting inductor L, and the other end of the phase-shifting inductor L is connected to the dotted end a of the primary side of the high-frequency transformer1Different name terminal b of primary side of high-frequency transformer1Is connected with the point b of the AC output end of the primary side H bridge; end c with same name of secondary side of transformer1The different name end d of the secondary side of the transformer is connected with the point c of the alternating current input end of the secondary side H bridge1An output capacitor C connected with the point d of the AC input end of the secondary H bridgeoThe load resistor R is connected to the secondary side H bridge direct current output end in parallel;
the control method comprises the following steps:
step 1: sampling output voltage U2Input voltage U1Output current i2Load current io；
Step 2: the output power P and the output power per unit value P are obtained according to the following modes:
P＝U2×i0
p＝P/PB
in the formula, PBIs a reference power, PB＝nU1UnV (8fL), n is the transformation ratio of the high-frequency transformer, n is 1, UnFor output voltage rating, f is switching frequency, f is 1/Ts,TsIs the switching period, L is the phase-shifting inductance;
step 3, obtaining the output power per unit value p and the critical output power p according to the step 2maxDetermining the internal phase-shift duty ratio D of the double-active-bridge converter1,
If p < pmax，
If p is greater than or equal to pmax，
In the formula (I), the compound is shown in the specification,k is a voltage conversion ratio defined as an input voltage U1And an output voltage U2The ratio of the product of the transformation ratio n of the high-frequency transformer,and 0<k≤1；
Step 4, the output voltage U of the double-active-bridge converter obtained by sampling in the step 1 is obtained2Sum voltage outer ring set value U*Making difference to obtain a voltage error signal delta U, inputting the voltage error signal delta U into the PI controller 1 to obtain an inner ring given value i2 *Then setting the inner ring to a given value i2 *And the average value of the output currentMaking difference to obtain current error signal delta I, inputting the current error signal delta I into PI controller 2 to obtain duty ratio D of external phase shift2；
Step 4.1, firstly, setting value U of voltage outer ring*And the output voltage U obtained by sampling in the step 12Obtaining a voltage error signal delta U by difference, wherein the delta U is equal to U*-U2Then, the voltage error signal delta U is used as the input of the PI controller 1 to obtain the inner ring set value i2 *(ii) a What is needed isThe PI controller 1 is a proportional-integral controller with a transfer function GPI1(s)The expression of (a) is as follows:
where s is the Laplace operator, kp1For the proportional term coefficient, k, of the PI controller 1i1Is the integral term coefficient of the PI controller 1;
step 4.2, setting the inner ring given value i obtained in the step 4.12 *Subtracting the average value of the output currentObtaining a current error signal Δ I, which is expressed as follows:
in the formula (I), the compound is shown in the specification,is the average value of the output current,m is the number of samplings i2mTo output a current i2The m-th sampling value;
step 4.3, the current error signal delta I obtained in the step 4.2 is used as the input of the PI controller 2 to obtain the duty ratio D of the outer phase shift2(ii) a The PI controller 2 is a proportional-integral controller, and the PI controller transfer function G thereofPI2(s)The expression of (a) is as follows:
wherein k isp2Is the proportional term coefficient, k, of the PI controller 2i2Is the integral term coefficient of the PI controller 2;
step 5, according to the internal phase-shifting duty ratio D obtained in the step 31And the duty ratio D of the outer phase shift obtained in the step 42And order D1、D2The following three conditions are simultaneously satisfied:
0≤D1≤D2/2
0≤D1≤1
0≤D2≤1
according to the dual bidirectional inner phase shift modulation method, the IGBTS is used as a switching tube1Drive signal Q of1For reference, the IGBTS of the respective and switch tube is generated1IGBTS (insulated Gate Bipolar translator) switching tube2IGBTS (insulated Gate Bipolar translator) switching tube3IGBTS (insulated Gate Bipolar translator) switching tube4IGBTS (insulated Gate Bipolar translator) switching tube5IGBTS (insulated Gate Bipolar translator) switching tube6IGBTS (insulated Gate Bipolar translator) switching tube7IGBTS (insulated Gate Bipolar translator) switching tube8Corresponding drive signal Q1、Q2、Q3、Q4、Q5、Q6、Q7、Q8Driving each switch tube IGBT to output voltage U2And maintaining the stability.
Preferably, the specific content of the dual bi-directional intra-phase shift modulation method in step 5 includes:
(1) switching tube IGBTS of primary side H bridge1IGBTS (insulated Gate Bipolar translator) switching tube2IGBTS (insulated Gate Bipolar translator) switching tube3IGBTS (insulated Gate Bipolar translator) switching tube4Drive signal Q of1、Q2、Q3、Q4Switching tube IGBTS of secondary side H bridge5IGBTS (insulated Gate Bipolar translator) switching tube6IGBTS (insulated Gate Bipolar translator) switching tube7IGBTS (insulated Gate Bipolar translator) switching tube8Drive signal Q of5、Q6、Q7、Q8Same frequency, switching tube IGBTS1And a switching tube IGBTS2Complementary conducting, switching tube IGBTS3And a switching tube IGBTS4Complementary conducting, switching tube IGBTS5And a switching tube IGBTS6Complementary conducting, switching tube IGBTS7And a switching tube IGBTS8Conducting complementarily;
(2) switching tube IGBTS4Drive signal Q of4Lags behind the switching tube IGBTS1Drive signal Q of1Switching tube IGBTS3Drive signal Q of3Lags behind the switching tube IGBTS2Drive signal Q of2Lag times are all TΔ1，
(3) Switching tube IGBTS8Drive signal Q of8IGBTS leading to switch tube5Drive signal Q of5Switching tube IGBTS7Drive signal Q of7IGBTS leading to switch tube6Drive signal Q of6The lead time is the same as the lag time in the above (2), i.e., the lead time is also TΔ1，
(4) Switching tube IGBTS5Drive signal Q of5Lags behind the switching tube IGBTS1Drive signal Q of1Switching tube IGBTS6Drive signal Q of6Lags behind the switching tube IGBTS2Drive signal Q of2Lag times are all TΔ2，
The invention discloses a double-active-bridge converter minimum reflux power double-bidirectional internal phase-shifting control method, which can realize minimum reflux power operation when outputting different powers on the basis of double-bidirectional internal phase-shifting modulation, and has the following beneficial effects:
1. on the basis of dual two-way inner phase shift modulation, the minimum reflux power operation can be realized when different powers are output, and particularly the per unit value p of the output power<Critical output power pmaxThe theoretical reflux power is zero. When the per-unit value p of the output power is more than or equal to the critical output power pmaxAnd when the output power requirement is met, the minimum reflux power operation can be realized.
2. Dual bidirectional internally phase-shifted modulation with suppressed externally phase-shifted duty cycle D2Internally phase-shifted duty cycle D due to disturbance1The problem of large amplitude fluctuation. Therefore, the method has better noise immunity when the power is transmitted in a steady state, so that the stability of the control system is improved.
3. Compared with double phase-shift modulation, the method is more suitable for being applied to a system with a double-active-bridge converter as an energy transmission unit, especially for occasions with high requirements on the control stability of the double-active-bridge converter, has very wide prospect and is easy to popularize.
Drawings
Fig. 1 is a schematic diagram of a dual active bridge converter topology.
Fig. 2 is a schematic diagram of a dual phase shift modulation.
Fig. 3 is a schematic diagram of the dual bi-directional phase shift modulation employed in the present invention.
Fig. 4 is a graph of equal output power per unit p based on dual inward phase shift modulation.
Fig. 5 is a graph of the equal return power per unit q-line based on dual inward phase shift modulation.
FIG. 6 is a trace diagram of the internal and external phase-shift duty cycle when the method of the present invention meets the minimum reflux power operation.
FIG. 7 is a control scheme of the method of the present invention.
FIG. 8 is a graph comparing the reflow power per unit q for two methods.
Fig. 9 is a simulation diagram of the power waveform M at a load of 32 Ω without the method of the present invention.
Fig. 10 is a simulation diagram of a power waveform M with a load of 32 Ω when the method of the present invention is employed.
Fig. 11 is a simulation diagram of a power waveform M when the load is 27 Ω without the method of the present invention.
Fig. 12 is a simulation diagram of a power waveform M when the load is 27 Ω by the method of the present invention.
FIG. 13 is the output voltage U of the method of the present invention2And (4) waveform diagrams.
Fig. 14 is a graph of the internal and external phase shift duty cycle variation during power switching in accordance with the present invention.
FIG. 15 is a graph of the variation of the internally shifted duty cycle under the disturbance of the externally shifted duty cycle in the dual phase shift modulation.
FIG. 16 is a graph of the internally shifted duty cycle variation under the disturbance of the externally shifted duty cycle of the dual bi-directional internally phase-shifted modulation.
Detailed Description
Fig. 1 is a schematic diagram of a dual active bridge converter topology in accordance with the present invention. As shown in FIG. 1, the topology of the dual active bridge converter according to the control method comprises a DC voltage source, an input capacitor CiA primary side H bridge, a phase-shifting inductor L, a high-frequency transformer, a secondary side H bridge and an output capacitor CoAnd a load resistor R.
The primary side H-bridge consists of four switching tube IGBTs, four anti-parallel diodes and four parasitic capacitors, wherein the four switching tube IGBTSs are respectively recorded as switching tubes IGBTSs1IGBTS (insulated Gate Bipolar translator) switching tube2IGBTS (insulated Gate Bipolar translator) switching tube3IGBTS (insulated Gate Bipolar translator) switching tube4Four anti-parallel diodes are respectively marked as D1、D2、D3、D4And four parasitic capacitances are respectively denoted as C1、C2、C3、C4(ii) a The secondary side H bridge consists of four switching tube IGBTs, four anti-parallel diodes and four parasitic capacitors, wherein the four switching tube IGBTS are respectively recorded as switching tubes IGBTS5IGBT S of switch tube6IGBTS (insulated Gate Bipolar translator) switching tube7IGBTS (insulated Gate Bipolar translator) switching tube8Four anti-parallel diodes are respectively marked as D5、D6、D7、D8And four parasitic capacitances are respectively denoted as C5、C6、C7、C8。
The DC voltage source and the input capacitor CiAfter being connected in parallel, the DC input end of the primary side H bridge is connected, the AC output end a point of the primary side H bridge is connected to one end of a phase-shifting inductor L, and the other end of the phase-shifting inductor L is connected to the dotted end a of the primary side of the high-frequency transformer1Different name terminal b of primary side of high-frequency transformer1Is connected with the point b of the AC output end of the primary side H bridge; end c with same name of secondary side of transformer1The different name end d of the secondary side of the transformer is connected with the point c of the alternating current input end of the secondary side H bridge1An output capacitor C connected with the point d of the AC input end of the secondary H bridgeoAnd the load resistor R is connected to the direct current output end of the secondary side H bridge in parallel.
The relevant electrical parameters when the invention is implemented are set as follows: input voltage U1400V, rated output voltage Un400V, output voltage U2The average output current of the voltage outer ring is controlled by the inner ring to be stabilized at 400V, namely U2＝UnVoltage conversion ratio k is 1, phase shift inductance L is 120 mu H, and input capacitance Ci110 muF, output capacitance Co110 μ F, and a load resistance R of 32 Ω.
The control method comprises sampling output voltage U2Input voltage U1Output current i2And a load current ioThe method comprises the following steps:
step 1: sampling output voltage U2Input voltage U1Output current i2Load current io。
Step 2: the output power P and the output power per unit value P are obtained according to the following modes:
P＝U2×i0
p＝P/PB
in the formula, PBIs a reference power, PB＝nU1UnV (8fL), n is the transformation ratio of the high-frequency transformer, n is 1, UnFor output voltage rating, f is switching frequency, f is 1/Ts,TsIs the switching period and L is the phase shifting inductance.
In this embodiment, the phase shift inductor L is 120 μ H, and the input voltage U is1400V, rated output voltage Un400V, 20kHz, PB＝8333w，p＝P/8333。
The principle of dual phase shift modulation and dual bi-directional phase shift modulation is shown in FIG. 2 and FIG. 3, wherein U isabIs primary side H bridge AC output voltage, UcdIs a secondary H-bridge AC input voltage iLIs the inductor current. The method adopts a dual bidirectional internal phase shift modulation principle and simultaneously satisfies the condition that D is more than or equal to 01≤D2/2,0≤D1≤1,0≤D2Under the condition of less than or equal to 1, using PBObtaining the output power per unit value p expression as p-2 [3D ] as the reference power1 2+D1(2-4D2)+2D2(D2-1)]The return power per unit value q is expressed as Based on the embodiment, fig. 4 shows a graph of per unit values p of equal output power based on dual inward phase shift modulation, where the per unit values p of output power of all points on the same line are the same; fig. 5 shows a graph of the per-unit values q of the equal return power based on dual inward phase shift modulation, where the per-unit values q of the return power at all points on the same line are the same. In addition, the output power per unit value p and the return power per unit value q increase in the direction of the arrow in the figure.
Step 3, obtaining the output power per unit value p and the critical output power p according to the step 2maxDetermining the internal phase-shift duty ratio D of the double-active-bridge converter1,
If p < pmax，
If p is greater than or equal to pmax，
In the formula (I), the compound is shown in the specification,k is a voltage conversion ratio defined as an input voltage U1And an output voltage U2The ratio of the product of the transformation ratio n of the high-frequency transformer,and 0<k≤1。
In this embodiment, if the voltage conversion ratio k is 1:
step 4, obtaining the output voltage U of the double-active-bridge converter according to the sampling in the step 12And outside of voltageRing set value U*Making difference to obtain a voltage error signal delta U, inputting the voltage error signal delta U into the PI controller 1 to obtain an inner ring given value i2 *Then setting the inner ring to a given value i2 *And the average value of the output currentMaking difference to obtain current error signal delta I, inputting the current error signal delta I into PI controller 2 to obtain duty ratio D of external phase shift2。
Step 4.1, firstly, setting value U of voltage outer ring*And the output voltage U obtained by sampling in the step 12Obtaining a voltage error signal delta U by difference, wherein the delta U is equal to U*-U2Then, the voltage error signal delta U is used as the input of the PI controller 1 to obtain the inner ring set value i2 *(ii) a The PI controller 1 is a proportional-integral controller with a transfer function GPI1(s)The expression of (a) is as follows:
where s is the Laplace operator, kp1For the proportional term coefficient, k, of the PI controller 1i1Is the integral term coefficient of the PI controller 1;
step 4.2, setting the inner ring given value i obtained in the step 4.12 *Subtracting the average value of the output currentObtaining a current error signal Δ I, which is expressed as follows:
in the formula (I), the compound is shown in the specification,is the average value of the output current,m is the number of samplingsNumber, i2mTo output a current i2The m-th sampling value;
step 4.3, the current error signal delta I obtained in the step 4.2 is used as the input of the PI controller 2 to obtain the duty ratio D of the outer phase shift2(ii) a The PI controller 2 is a proportional-integral controller, and the PI controller transfer function G thereofPI2(s)The expression of (a) is as follows:
wherein k isp1For the proportional term coefficient, k, of the PI controller 1i1For PI controller 1 integral term coefficient, kp2Is the proportional term coefficient, k, of the PI controller 2i2Is the integral term coefficient of the PI controller 2.
In the present embodiment, the voltage outer loop given value U*＝400V，kp1＝0.00312，ki1＝5.2，kp2＝0.0096，ki296.15. The internal phase shift duty ratio D obtained by the step 31And the duty ratio D of the outer phase shift obtained in the step 42Satisfies the following conditions: if the output power per unit value D2＝2D1At this time, the reflux power is theoretically zero; if the output power per unit value D2＝0.5D1+0.5. Referring to fig. 6, it is a trace diagram of the internal and external phase-shift duty ratio when the method of the present invention meets the operation of minimum reflux power.
Step 5, according to the internal phase-shifting duty ratio D obtained in the step 31And the duty ratio D of the outer phase shift obtained in the step 42And order D1、D2Simultaneously satisfy the following threeThe following conditions are set out:
0≤D1≤D2/2
0≤D1≤1
0≤D2≤1
according to the dual bidirectional inner phase shift modulation method, the IGBTS is used as a switching tube1Drive signal Q of1For reference, the IGBTS of the respective and switch tube is generated1IGBTS (insulated Gate Bipolar translator) switching tube2IGBTS (insulated Gate Bipolar translator) switching tube3IGBTS (insulated Gate Bipolar translator) switching tube4IGBTS (insulated Gate Bipolar translator) switching tube5IGBTS (insulated Gate Bipolar translator) switching tube6IGBTS (insulated Gate Bipolar translator) switching tube7IGBTS (insulated Gate Bipolar translator) switching tube8Corresponding drive signal Q1、Q2、Q3、Q4、Q5、Q6、Q7、Q8Driving each switch tube IGBT to output voltage U2And maintaining the stability.
The specific content of the dual bi-directional internal phase shift modulation method comprises the following steps:
(1) switching tube IGBTS of primary side H bridge1IGBTS (insulated Gate Bipolar translator) switching tube2IGBTS (insulated Gate Bipolar translator) switching tube3IGBTS (insulated Gate Bipolar translator) switching tube4Drive signal Q of1、Q2、Q3、Q4Switching tube IGBTS of secondary side H bridge5IGBTS (insulated Gate Bipolar translator) switching tube6IGBTS (insulated Gate Bipolar translator) switching tube7IGBTS (insulated Gate Bipolar translator) switching tube8Drive signal Q of5、Q6、Q7、Q8Same frequency, switching tube IGBTS1And a switching tube IGBTS2Complementary conducting, switching tube IGBTS3And a switching tube IGBTS4Complementary conducting, switching tube IGBTS5And a switching tube IGBTS6Complementary conducting, switching tube IGBTS7And a switching tube IGBTS8And conducting complementarily.
(2) Switching tube IGBTS4Drive signal Q of4Lags behind the switching tube IGBTS1Drive signal Q of1Switching tube IGBTS3Drive signal Q of3Lags behind the switching tube IGBTS2Drive signal Q of2Lag times are all TΔ1，
(3) Switching tube IGBTS8Drive signal Q of8IGBTS leading to switch tube5Drive signal Q of5Switching tube IGBTS7Drive signal Q of7IGBTS leading to switch tube6Drive signal Q of6The lead time is the same as the lag time in the above (2), i.e., the lead times are all TΔ1，
(4) Switching tube IGBTS5Drive signal Q of5Lags behind the switching tube IGBTS1Drive signal Q of1Switching tube IGBTS6Drive signal Q of6Lags behind the switching tube IGBTS2Drive signal Q of2Lag times are all TΔ2，
The control structure of the method of the present invention is shown in fig. 7.
FIG. 8 is a graph comparing the reflow power per unit q for two methods. That is, in the present embodiment, a comparison graph of the per unit value q of the reflux power by using the optimization method of the present invention and without using the optimization method of the present invention shows that the reflux power is well suppressed by using the optimization method of the present invention.
The following is a specific observation through simulation of the advantages of the method of the present invention compared with the method without the present invention. It was verified by simulation that fig. 9 is a simulation diagram of the power waveform M at a load of 32 Ω without the method of the present invention, where the portion below the horizontal axis is the return power. FIG. 10 is a simulation diagram of a power waveform M under a load of 32 Ω when the method of the present invention is applied, and the per-unit output power value p is 0.6 corresponding to point A in FIG. 6, because point A is smaller than the critical transmission power pmaxSo the theoretical return power Q is zero. It is obvious that the reflux power is optimized by the method of the invention. At a certain moment, the load is switched, and FIG. 11 shows the work at 27 Ω without the method of the present inventionSimulation plot of rate waveform M. FIG. 12 is a simulation diagram of a power waveform M with a load of 27 Ω according to the method of the present invention, corresponding to point B in FIG. 6, since point B is greater than the critical transmission power pmaxMeanwhile, the tangent point between the line where the output power per unit value p is 0.71 and the line where the reflux power per unit value Q is 0.0012 is the same, and the reflux power Q is theoretically smaller than other points. It can be seen from the comparison between fig. 11 and fig. 12 that the backflow power is reduced by the minimum backflow power control method proposed by the present invention.
Referring to fig. 13, under the control of the voltage outer loop average output current inner loop, the output voltage can be kept stable at different power switching. Referring to fig. 14, it is a dynamic variation process of the internal and external phase shift duty ratio in the dynamic switching process. It can be seen that the internal and external phase-shift duty ratio after stabilization satisfies the relationship of theoretical analysis.
FIG. 15 and FIG. 16 show the inter-phase duty ratio D of the bi-phase modulation and the bi-directional inter-phase modulation employed in the present invention when the per-unit output power p is 0.11Phase-shift-with-outside duty cycle D2Varying waveform pattern giving phase-shifted duty cycle D at steady-state operating point2Minor disturbances Δ D2So that the internal phase shift duty ratio generates disturbance Delta D1It is obvious that the dual bidirectional internally phase-shifted modulation method adopted by the invention at steady state shifts the phase duty ratio D externally2Internal phase shift duty cycle D due to disturbance1The problem of fluctuations has a very good inhibitory effect.
Claims (1)
1. The double-active-bridge converter minimum reflux power double-bidirectional internal phase-shifting control method is characterized in that the topological structure of the double-active-bridge converter related by the control method comprises a direct-current voltage source and an input capacitor CiA primary side H bridge, a phase-shifting inductor L, a high-frequency transformer, a secondary side H bridge and an output capacitor CoAnd a load resistor R; the primary side H-bridge consists of four switching tube IGBTs, four anti-parallel diodes and four parasitic capacitors, wherein the four switching tube IGBTSs are respectively recorded as switching tubes IGBTSs1IGBTS (insulated Gate Bipolar translator) switching tube2IGBTS (insulated Gate Bipolar translator) switching tube3IGBTS (insulated Gate Bipolar translator) switching tube4Four antiparallel diodesThe tubes are denoted respectively by D1、D2、D3、D4And four parasitic capacitances are respectively denoted as C1、C2、C3、C4(ii) a The secondary side H bridge consists of four switching tube IGBTs, four anti-parallel diodes and four parasitic capacitors, wherein the four switching tube IGBTS are respectively recorded as switching tubes IGBTS5IGBTS (insulated Gate Bipolar translator) switching tube6IGBTS (insulated Gate Bipolar translator) switching tube7IGBTS (insulated Gate Bipolar translator) switching tube8Four anti-parallel diodes are respectively marked as D5、D6、D7、D8And four parasitic capacitances are respectively denoted as C5、C6、C7、C8；
The DC voltage source and the input capacitor CiAfter being connected in parallel, the DC input end of the primary side H bridge is connected, the AC output end a point of the primary side H bridge is connected to one end of a phase-shifting inductor L, and the other end of the phase-shifting inductor L is connected to the dotted end a of the primary side of the high-frequency transformer1Different name terminal b of primary side of high-frequency transformer1Is connected with the point b of the AC output end of the primary side H bridge; end c with same name of secondary side of transformer1The different name end d of the secondary side of the transformer is connected with the point c of the alternating current input end of the secondary side H bridge1An output capacitor C connected with the point d of the AC input end of the secondary H bridgeoThe load resistor R is connected to the secondary side H bridge direct current output end in parallel;
the control method comprises the following steps:
step 1: sampling output voltage U2Input voltage U1Output current i2Load current io；
Step 2: the output power P and the output power per unit value P are obtained according to the following modes:
P＝U2×i0
p＝P/PB
in the formula, PBIs a reference power, PB＝nU1UnV (8fL), n is the transformation ratio of the high-frequency transformer, n is 1, UnFor output voltage rating, f is switching frequency, f is 1/Ts,TsIs the switching period, L is the phase-shifting inductance;
step 3, according to the output obtained in step 2Power per unit value p and critical output power pmaxDetermining the internal phase-shift duty ratio D of the double-active-bridge converter1,
If p < pmax，
If p is greater than or equal to pmax，
In the formula (I), the compound is shown in the specification,k is a voltage conversion ratio defined as an input voltage U1And an output voltage U2The ratio of the product of the transformation ratio n of the high-frequency transformer,and 0<k≤1；
Step 4, the output voltage U of the double-active-bridge converter obtained by sampling in the step 1 is obtained2Sum voltage outer ring set value U*Making difference to obtain a voltage error signal delta U, inputting the voltage error signal delta U into the PI controller 1 to obtain an inner ring given value i2 *Then setting the inner ring to a given value i2 *And the average value of the output currentMaking difference to obtain current error signal delta I, inputting the current error signal delta I into PI controller 2 to obtain duty ratio D of external phase shift2；
Step 4.1, firstly, setting value U of voltage outer ring*And the output voltage U obtained by sampling in the step 12Obtaining a voltage error signal delta U by difference, wherein the delta U is equal to U*-U2Then, the voltage error signal delta U is used as the input of the PI controller 1 to obtain the inner ring set value i2 *(ii) a The PI controller 1 is a proportional-integral controller with a transfer function GPI1(s)Expression (2)The following were used:
where s is the Laplace operator, kp1For the proportional term coefficient, k, of the PI controller 1i1Is the integral term coefficient of the PI controller 1;
step 4.2, setting the inner ring given value i obtained in the step 4.12 *Subtracting the average value of the output currentObtaining a current error signal Δ I, which is expressed as follows:
in the formula (I), the compound is shown in the specification,is the average value of the output current,m is the number of samplings i2mTo output a current i2The m-th sampling value;
step 4.3, the current error signal delta I obtained in the step 4.2 is used as the input of the PI controller 2 to obtain the duty ratio D of the outer phase shift2(ii) a The PI controller 2 is a proportional-integral controller, and the PI controller transfer function G thereofPI2(s)The expression of (a) is as follows:
wherein k isp2Is the proportional term coefficient, k, of the PI controller 2i2Is the integral term coefficient of the PI controller 2;
step 5, according to the internal phase-shifting duty ratio D obtained in the step 31And the duty ratio D of the outer phase shift obtained in the step 42And order D1、D2The following three conditions are simultaneously satisfied:
0≤D1≤D2/2
0≤D1≤1
0≤D2≤1
according to the dual bidirectional inner phase shift modulation method, the IGBTS is used as a switching tube1Drive signal Q of1For reference, the IGBTS of the respective and switch tube is generated1IGBTS (insulated Gate Bipolar translator) switching tube2IGBTS (insulated Gate Bipolar translator) switching tube3IGBTS (insulated Gate Bipolar translator) switching tube4IGBTS (insulated Gate Bipolar translator) switching tube5IGBTS (insulated Gate Bipolar translator) switching tube6IGBTS (insulated Gate Bipolar translator) switching tube7IGBTS (insulated Gate Bipolar translator) switching tube8Corresponding drive signal Q1、Q2、Q3、Q4、Q5、Q6、Q7、Q8Driving each switch tube IGBT to output voltage U2Maintaining stability;
the specific contents of the dual bidirectional inner phase shift modulation method comprise:
(1) switching tube IGBTS of primary side H bridge1IGBTS (insulated Gate Bipolar translator) switching tube2IGBTS (insulated Gate Bipolar translator) switching tube3IGBTS (insulated Gate Bipolar translator) switching tube4Drive signal Q of1、Q2、Q3、Q4Switching tube IGBTS of secondary side H bridge5IGBTS (insulated Gate Bipolar translator) switching tube6IGBTS (insulated Gate Bipolar translator) switching tube7IGBTS (insulated Gate Bipolar translator) switching tube8Drive signal Q of5、Q6、Q7、Q8Same frequency, switching tube IGBTS1And a switching tube IGBTS2Complementary conducting, switching tube IGBTS3And a switching tube IGBTS4Complementary conducting, switching tube IGBTS5And a switching tube IGBTS6Complementary conducting, switching tube IGBTS7And a switching tube IGBTS8Conducting complementarily;
(2) switching tube IGBTS4Drive signal Q of4Lags behind the switching tube IGBTS1Drive signal Q of1Switching tube IGBTS3Drive signal Q of3Lags behind the switching tube IGBTS2Drive signal Q of2Lag times are all TΔ1，
(3) Switching tube IGBTS8Drive signal Q of8IGBTS leading to switch tube5Drive signal Q of5Switching tube IGBTS7Drive signal Q of7IGBTS leading to switch tube6Drive signal Q of6The lead time is the same as the lag time in the above (2), i.e., the lead time is also TΔ1，
(4) Switching tube IGBTS5Drive signal Q of5Lags behind the switching tube IGBTS1Drive signal Q of1Switching tube IGBTS6Drive signal Q of6Lags behind the switching tube IGBTS2Drive signal Q of2Lag times are all TΔ2，
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| CN110719030B (en) * | 2019-08-27 | 2022-02-08 | 河北工业大学 | Dual phase-shift modulation method for isolated bidirectional full-bridge DC-DC converter |
| CN111049392B (en) * | 2019-12-27 | 2021-06-11 | 合肥工业大学 | Double-active-bridge expanded phase-shifting minimum reflux power control method based on coordinate transformation |
| CN111416523B (en) * | 2020-04-17 | 2021-07-30 | 合肥科威尔电源系统股份有限公司 | Soft charging control system and method for double-active-bridge DC/DC converter |
| CN111987918B (en) * | 2020-09-25 | 2022-01-21 | 深圳市永联科技股份有限公司 | Bidirectional DC-DC soft switch control method |
| CN112260553B (en) * | 2020-12-18 | 2022-03-11 | 深圳市永联科技股份有限公司 | Control method applied to bidirectional converter |
| CN113141119B (en) * | 2021-04-19 | 2022-04-15 | 中国矿业大学 | Method for optimally controlling backflow power and dynamic performance of dual-active-bridge converter |
| CN113037097B (en) * | 2021-04-29 | 2022-03-25 | 常熟理工学院 | Modulation control method of resonant double-active-bridge converter |
| CN113179024B (en) * | 2021-05-12 | 2022-10-28 | 福州大学 | Discrete iteration prediction control model of three-phase-shift double-active full-bridge direct-current converter |
| CN114157160A (en) * | 2021-12-06 | 2022-03-08 | 西安西电电力系统有限公司 | Converter based on three active bridges and control method thereof |
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