CN108539983B - 一种快速充放电双向反激电路 - Google Patents

一种快速充放电双向反激电路 Download PDF

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CN108539983B
CN108539983B CN201810355503.9A CN201810355503A CN108539983B CN 108539983 B CN108539983 B CN 108539983B CN 201810355503 A CN201810355503 A CN 201810355503A CN 108539983 B CN108539983 B CN 108539983B
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switch tube
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CN108539983A (zh
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王来利
陈阳
田莫帆
卢晓辉
金久洋
孙亚祺
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Xian Jiaotong University
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    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
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    • H02J—CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
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    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
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Abstract

本发明公开的一种快速充放电双向反激电路，包括供电电源Vin、前级升压辅助电路和输出电容Cout，前级升压辅助电路的输入端与供电电源Vin相连，输出端与双向反激电路的输入端相连，输出电容Cout为双向反激电路的输出端；前级升压辅助电路包括与供电电源Vin串联相连的开关管MOSFET Q3和初级高压储能电容CHV；当双向反激电路能量反向流动时，开关管MOSFET Q3没有导通信号，电流不能从漏极流向源级，从而被储存在初级高压储能电容CHV中，高压电容储存电荷后，CHV两端电压升高，会同时减小Tch(双向反激电路充电时间)和Tdisch(双向反激电路放电时间)，从而加快介电弹性体(电容性负载)的重复充放电速度。

Description

一种快速充放电双向反激电路
技术领域
本发明属于电力电子技术在驱动介电弹性体的应用技术领域，具体涉及一种快速充放电双向反激电路。
背景技术
近年来，随着AI技术软硬件上的双重突破，机器人行业又迎来了新的发展春天。其中近期发展突飞猛进的仿生柔性机器人相比于传统刚性结构的机器人在海底勘探、消防救援以及军事上都有明显的优势。
介电弹性体在施加适当电场后可以产生超过100％的形变，并因为其轻型化、低噪声、高效率、高应变，高柔软的特性，在仿生柔性机器人中常被用作人造肌肉。为了适应不同应用场合，介电弹性体已经衍生出性能更加优秀的材料，如高频率响应范围、无需预拉伸的电活性材料。小型鸟类翅膀震动频率较高，蜂鸟翅膀震动频率可高达80Hz，快速响应的介电弹性体为小型仿生鸟的实现提供了可能。
虽然介电弹性体具有优良的特性，但是在应用中存在很多问题。首先，施加在介电弹性体的电场能量在每次转化中仅有小于10％的能量转化为机械能，剩下部分能量需要通过电路回馈电源或者直接通过电阻耗散。而通过电阻耗散会大大降低电路的效率，从而降低整体系统续航；其次，在高速响应介电弹性体应用的场合，需要供电电路达到足够的充放电速度。双向反激拓扑在DCM工作模式下，充电、放电时间与供电电压和原边变压器电流峰值大小成反比例关系，但实际应用中常用锂电池或纽扣电池作为电源，通过DC-DC电路提升到高电压。由于电源电压和能够提供的电流峰值有限，较低的电压和充电电流会使充电速度会受到较大限制。增加变压器匝比也能减小充放电周期时间，但会大大增加变换器的体积和设计难度，并且会增大副边MOSFET的耐压。
因此，为了实现介电弹性体的快速充放电，迫切需要一种改良的双向反激电路提升充放电速度。
发明内容
为了克服上述现有技术存在的缺陷，本发明的目的在于提供一种快速充放电双向反激电路，能够实现在高电压小电容负载应用场合，利用低压锂电池可以实现快速充放电。
本发明是通过以下技术方案来实现：
本发明公开的一种快速充放电双向反激电路，包括供电电源Vin、前级升压辅助电路和输出电容Cout，前级升压辅助电路的输入端与供电电源Vin相连，输出端与双向反激电路的输入端相连，输出电容Cout为双向反激电路的输出端；
所述前级升压辅助电路包括与供电电源Vin串联相连的开关管MOSFET Q3和初级高压储能电容CHV，开关管MOSFET Q3的源级与供电电源Vin的正极相连，漏极分别与初级高压储能电容CHV的正极和双向反激电路的输入端相连；当双向反激电路中的能量反向流动时，开关管MOSFET Q3无信号导通，电流被储存在初级高压储能电容CHV中。
优选地，双向反激电路包括原边电路、副边电路和高频反激变压器，原边电路上设有原边开关管MOSFET Q1，副边电路上设有副边开关管MOSFET Q2，在副边开关管MOSFET Q2上还并联设有阻断二极管D1和外部续流二极管D2；输出电容Cout设置在副边电路上；
高频反激变压器的原边绕组的一端与开关管MOSFET Q3的漏极相连，另一端与原边开关管MOSFET Q1的漏极相连；原边开关管MOSFET Q1的源极与供电电源Vin的负极相连；
高频反激变压器的副边绕组一端与输出电容Cout的正极相连，另一端分别连接阻断二极管D1阳极和外部续流二极管D2阴极，阻断二极管D1的阴极连接副边开关管MOSFET Q2漏极，副边开关管MOSFET Q2源极分别接外部续流二极管D2阳极和输出电容Cout负极，供电电源Vin负极和输出电容Cout负极均接地。
进一步优选地，在充电时间内，副边开关管MOSFET Q2保持关断，原边开关管MOSFET Q1根据控制电路产生的PWM波形进行开通关断操作；其中：
当原边开关管MOSFET Q1开通时，高频反激变压器原边的励磁电流ILm线性增大，高频反激变压器副边感应出上负下正的电压，此时外部续流二极管D2反向截止，副边电路无电流，励磁电流ILm被限制到峰值Ip peak，电流到达Ip peak后开关管MOSFET Q1关断，储存在高频反激变压器内的能量按：输出电容Cout-外部续流二极管D2的流向进行泄放，此时高频反激变压器的励磁变感Lm副边感应出上正下负的电压，副边电流Is从最大值Is peak开始线性增大，同时输出电容Cout上的电压缓慢上升，当副边电流Is下降到零后，开关管MOSFET Q1再次开通，重新开始一个新的周期，直到输出电容Cout上电压达到设定值。
更进一步优选地，在充电时间内，该快速充放电双向反激电路的充电时间按式(1)计算：
Figure GDA0002229486230000031
其中，no为充电过程所需开关周期数；N为高频反激变压器的匝数比；Uin为供电电源Vin的输入电压大小。
进一步优选地，在放电时间内，原边开关管MOSFET Q1保持关断状态，副边开关管MOSFET Q2根据控制电路产生的PWM波形进行开通关断操作；其中：
当副边开关管MOSFET Q2开通时，输出电容Cout的电压施加在高频反激变压器的励磁变感Lm副边，流过高频反激变压器的励磁变感Lm副边的电流Is线性增大，此时原边感应出上负下正的电压，原边开关管MOSFET Q1保持关断状态，其内部寄生二极管保持反向阻断，原边没有感应电流；当控制电路检测到高频反激变压器的励磁变感Lm副边电流Is峰值达到设定值Is peak，副边开关管MOSFET Q2关断，此时原边感应出上正下负的电压，储存在高频反激变压器的励磁变感Lm中的能量按照：原边开关管MOSFET Q1-励磁变感Lm-初级高压储能电容CHV的流向进行泄放，原边励磁电流从峰值Ip peak线性下降，直到为零，此时副边开关管MOSFET Q2再次导通，新的周期反复进行，直到输出电压降低为零。
更进一步优选地，在放电时间内，该快速充放电双向反激电路的放电时间按式(2)计算：
Figure GDA0002229486230000041
nos为放电过程所需的开关周期数；N为高频反激变压器的匝数比；Uin为供电电源Vin的输入电压大小。
优选地，前级升压辅助电路中还包括与初级高压储能电容CHV并联设置的放电电阻R1。
优选地，选用二极管替换开关管MOSFET Q3，实现升压效果。
与现有技术相比，本发明具有以下有益的技术效果：
本发明公开的快速充放电双向反激电路，包括固定输入电压(供电电源)Vin、前级升压辅助电路和输出电容Cout，前级升压辅助电路的输入端与供电电源Vin相连，输出端与双向反激电路的输入端相连，输出电容Cout为双向反激电路的输出端；由于前级升压电路的加入，使得本发明能够利用双向反激电路的能量能够反向流动的特性，加入开关管MOSFET Q3和初级高压储能电容CHV，开关管MOSFET Q3的源级与供电电源Vin的正极相连，漏极与初级高压储能电容CHV的正极相连，当双向反激电路能量反向流动时，开关管MOSFET Q3没有导通信号，电流不能从漏极流向源级，从而被储存在初级高压储能电容CHV中，高压电容储存电荷后，CHV两端电压升高，会同时减小Tch(双向反激电路充电时间)和Tdisch(双向反激电路放电时间)，从而加快介电弹性体(电容性负载)的重复充放电速度。
进一步地，在前级升压电路中还包括与初级高压储能电容CHV并联设置的放电电阻R1，该放电电阻R1的电阻值非常大，作为升压辅助电路的泄放回路，需要说明的是，R1可以根据需要进行取舍。
附图说明
图1是快速充放电双向反激电路示意图；
图2是驱动信号和工作波形示意图；
图3是快速充放电波形示意图；
图4a为t0-t1、t1-t2时段电路工作等效图；
图4b为t2-t3时段电路工作等效图；
图4c为t3-t4时段电路工作等效图；
图4d为t4-t5时段电路工作等效图。
具体实施方式
下面结合具体的实施例对本发明做进一步的详细说明，所述是对本发明的解释而不是限定。
参见图1，为本发明的快速充放电双向反激电路示意图，整体包括三部分：供电电源Vin(固定输入电压)，前级升压辅助电路(虚线部分)和输出电容Cout。前级升压辅助电路的输入端与供电电源Vin相连，输出端与双向反激电路的输入端相连，输出电容Cout为双向反激电路的输出端，双向反激电路工作在BCM工作模式。
其中，前级升压辅助电路由串联在初级回路的开关管MOSFET Q3和初级高压储能电容CHV组成；开关管MOSFET Q3的源级与供电电源Vin的正极相连，漏极分别与初级高压储能电容CHV的正极和双向反激电路的输入端相连；当双向反激电路中的能量反向流动时，开关管MOSFET Q3无信号导通，电流被储存在初级高压储能电容CHV中。
所述双向反激电路中，双向反激电路包括原边电路、副边电路和高频反激变压器，原边电路上设有原边开关管MOSFET Q1，副边电路上设有副边开关管MOSFET Q2，在副边开关管MOSFET Q2上还并联设有阻断二极管D1和外部续流二极管D2；输出电容Cout设置在副边电路上；高频反激变压器的原边绕组的一端与开关管MOSFET Q3的漏极相连，另一端与原边开关管MOSFET Q1的漏极相连；原边开关管MOSFET Q1的源极与供电电源Vin的负极相连；
高频反激变压器的副边绕组一端与输出电容Cout的正极相连，另一端分别连接阻断二极管D1阳极和外部续流二极管D2阴极，阻断二极管D1的阴极连接副边开关管MOSFET Q2漏极，副边开关管MOSFET Q2源极分别接外部续流二极管D2阳极和输出电容Cout负极，供电电源Vin负极和输出电容Cout负极均接地。
采用外部续流二极管D2来代替副边MOSFET性能较差的寄生二极管，可以获得更高的工作效率，由于外部续流二极管D2管压降与副边MOSFET寄生二极管的管压降相同，因此需要阻断二极管D1对寄生二极管进行阻断。
优选地，前级升压辅助电路中还包括电阻R1，电阻R1为放电电阻，电阻值很大，可以根据实际需要进行取舍。
普通双向反激电路在驱动介电弹性体时，将负载电容上未转化为机械能的电能反馈回电源，而本发明由于前级升压辅助电路的加入，反馈的能量被开关管MOSFET Q3阻断，这部分能量储存在初级高压储能电容CHV中，通过选取不同的CHV大小，可以在变压器原边线圈获得不同幅值的高电压(相对于锂电池电压)，产生的高电压能够显著缩短放电时间，同时能够有效提高下一次充电速度。在每个充电周期内，由于机械能的损失，一定有能量从供电电压(固定输入电压)Vin传递到输出，因此开关管MOSFET Q3需要在每个周期内的特定时间开通。
优选地，开关管MOSFET Q3可以用二极管替换，亦可以实现同样的升压效果。
下面结合图2和图3的驱动信号和工作波形对电路的快速充电、放电原理进行详细介绍。
如图2所示，在充电时间内，副边开关管MOSFET Q2保持关断状态，图1中原边开关管MOSFET Q1根据控制电路产生的PWM波形进行开关断，原边开关管MOSFET Q1开通时，高频反激变压器原边的励磁电流ILm线性增大，变压器副边感应出上负下正的电压，此时外部续流二极管D2反向截止，副边电路无电流，励磁电流ILm被限制到峰值Ip peak，电流到达Ip peak后开关管MOSFET Q1关断，储存在高频反激变压器内的能量通过副边电路的电容Cout-外部续流二极管D2的流向进行泄放，此时高频反激变压器副边感应出上正下负的电压，副边电流Is从最大值Is peak开始线性增大，同时电容Cout上的电压缓慢上升，当副边电流Is下降到零后，开关管MOSFET Q1再次开通，重新开始一个新的周期，直到电容Cout上电压达到设定值。
原边开关管MOSFET Q1保持关断状态，副边开关管MOSFET Q2根据控制电路产生的PWM波形进行开通关断操作。其中：当副边开关管MOSFET Q2开通时，负载电容Cout的电压施加在高频反激变压器的副边，流过高频反激变压器副边的电流Is线性增大，此时原边感应出上负下正的电压，原边开关管MOSFET Q1保持关断状态，原边开关管MOSFET Q1的反向并联寄生二极管保持反向阻断，原边没有感应电流；当控制电路检测到高频反激变压器的励磁变感Lm副边电流Is峰值达到设定值Is peak，副边开关管MOSFET Q2关断，此时原边感应出上正下负的电压，储存在高频反激变压器中的能量通过原边电路的原边开关管MOSFET Q1-励磁变感Lm-初级高压储能电容CHV的流向进行泄放，原边励磁电流从峰值Ip peak线性下降，直到为零，此时副边开关管MOSFET Q2再次导通，新的周期反复进行，直到输出电压降低为零。
双向反激电路的充电时间计算公式为：
Figure GDA0002229486230000081
其中，no为充电过程所需开关周期数；N为高频反激变压器的匝数比；Uin为供电电源Vin的输入电压大小；
双向反激电路的放电时间计算公式为：
Figure GDA0002229486230000082
nos为放电过程所需的开关周期数；N为高频反激变压器的匝数比；Uin为供电电源Vin的输入电压大小。
高压电容储存电荷后，CHV两端电压升高，会同时减小Tch和Tdisch，从而加快介电弹性体(电容性负载)的重复充放电速度。
参见图4a至图4d，为各个时段的电路工作等效图，具体工作状态如下：
t0-t1时段：如图4a所示，副边开关管MOSFET Q2保持关断，原边开关管MOSFET Q1开通时，开关管MOSFET Q3开通，电流流向为Vin-Q3-Lm-Q1-Vin，此时刻能量保存在Lm中；原边开关管MOSFET Q1关断后，副边电流流向为Lm s-Cout-D2-Lm s，此时Lm s为原边励磁电流等效于副边的电感。原边、副边的能量流向均为顺时针。第一个充电过程与普通双向反激电路相同。
t1-t2时段：此时电路保持高电压一段时间，称为延迟时间。实际工作中延迟时间根据延迟时间大小决定延时时间内反激电路的工作状态。延时时间较短时，无需考虑负载等效电阻存在导致的电压下降，反激电路在延时时间内不工作。但延时时间较长时，为了维持输出电容电压，在延迟时间内将反激电路设置到稳压输出。现假设延迟时间很短，反激电路在本时间段不工作。
t2-t3时段：如图4b所示，双向反激电路能量反向流动，输出电容Cout开始放电。升压辅助电路内开关管MOSFET Q3保持断开，副边开关管MOSFET Q2根据PWM波形进行开关断，副边开关管MOSFET Q2开通时电流流向为Cout-Lms-D1-Q2-Cout，能量储存在变压器中；副边开关管MOSFET Q2关断后，原边开关管MOSFET Q1采用宽禁带器件，可以利用其自身寄生二极管进行续流，此时电流流向Lm-CHV-Q1-Lm。取放电电阻R1为无穷大，在不考虑能量损失的情况下，介电弹性体转化机械能后残留电场能量全部转移到了原边储能电容CHV中，根据电容储能公式W＝1/2CHVU2，选取不同的电容CHV值，Cout两端电压在完全降为零后，可以在变压器原边获得不同的电压峰值。
t3-t4时段：如图4c所示，合适的电容CHV大小可以在原边获得较高电压而不会损坏控制电路和绝缘设计。根据双向反激电路放电时间计算公式可知，充电时间可以大大减小。此阶段进入下一周期的充电过程，升压辅助电路的开关管MOSFET Q3保持断开，原边开关管MOSFET Q1开通时，原边电流流向为CHV-Lm-Q1-CHV。原边开关管MOSFET Q1关断后，副边电流流向为Lm s-Cout-D2-Lm s，此时Lm s为原边励磁电流等效于副边的电感。原边、副边的能量流向均为顺时针，负载电容Cout两端电压逐渐升高，同时储能电容CHV电压逐渐下降。
t4-t5时段：如图4d所示，在负载电容Cout未达到目标电压，储能电容CHV电压已经降低为输入电压Vin，可以通过传感器采集，在CHV电压降低到输入电压Vin后，开通开关管MOSFET Q3，因而此阶段的能量来自于充电电压源。等效电路图与t1-t2阶段相同。t5时刻，负载电容Cout上电压达到目标值，进入延迟时间阶段，反激电路暂停工作。
特别需要说明的是，本发明所采用的双向反激电路也可以更改为其他双向电路拓扑，只要提升原边电压能够加快充电速度的拓扑均可以使用。
上述虽然结合附图对本发明的具体实施方法进行了描述，但本发明的保护范围不限于此，所属领域技术人员在不脱离本发明原理的前提下做出的各种修改和变形仍在本发明的保护范围之内。

Claims (8)

1.一种快速充放电双向反激电路，其特征在于，包括供电电源Vin、前级升压辅助电路和输出电容Cout，前级升压辅助电路的输入端与供电电源Vin相连，输出端与双向反激电路的输入端相连，输出电容Cout为双向反激电路的输出端；
所述前级升压辅助电路包括与供电电源Vin串联相连的开关管MOSFET Q3和初级高压储能电容CHV，开关管MOSFET Q3的源级与供电电源Vin的正极相连，漏极分别与初级高压储能电容CHV的正极和双向反激电路的输入端相连；当双向反激电路中的能量反向流动时，开关管MOSFET Q3无信号导通，电流被储存在初级高压储能电容CHV中。
2.根据权利要求1所述的快速充放电双向反激电路，其特征在于，双向反激电路包括原边电路、副边电路和高频反激变压器，原边电路上设有原边开关管MOSFET Q1，副边电路上设有副边开关管MOSFET Q2，在副边开关管MOSFET Q2上还并联设有阻断二极管D1和外部续流二极管D2；输出电容Cout设置在副边电路上；
高频反激变压器的原边绕组的一端与开关管MOSFET Q3的漏极相连，另一端与原边开关管MOSFET Q1的漏极相连；原边开关管MOSFET Q1的源极与供电电源Vin的负极相连；
高频反激变压器的副边绕组一端与输出电容Cout的正极相连，另一端分别连接阻断二极管D1阳极和外部续流二极管D2阴极，阻断二极管D1的阴极连接副边开关管MOSFET Q2漏极，副边开关管MOSFET Q2源极分别接外部续流二极管D2阳极和输出电容Cout负极，供电电源Vin负极和输出电容Cout负极均接地。
3.根据权利要求2所述的快速充放电双向反激电路，其特征在于，在充电时间内，副边开关管MOSFET Q2保持关断，原边开关管MOSFET Q1根据控制电路产生的PWM波形进行开通关断操作；其中：
当原边开关管MOSFET Q1开通时，高频反激变压器原边的励磁电流ILm线性增大，高频反激变压器副边感应出上负下正的电压，此时外部续流二极管D2反向截止，副边电路无电流，励磁电流ILm被限制到峰值Ip peak，电流到达Ippeak后开关管MOSFET Q1关断，储存在高频反激变压器的励磁变感Lm内的能量按：输出电容Cout-外部续流二极管D2的流向进行泄放，此时高频反激变压器副边感应出上正下负的电压，副边电流Is从最大值Is peak开始线性增大，同时输出电容Cout上的电压缓慢上升，当副边电流Is下降到零后，开关管MOSFET Q1再次开通，重新开始一个新的周期，直到输出电容Cout上电压达到设定值。
4.根据权利要求3所述的快速充放电双向反激电路，其特征在于，在充电时间内，该快速充放电双向反激电路的充电时间按式(1)计算：
Figure FDA0002229486220000021
其中，no为充电过程所需开关周期数；N为高频反激变压器的匝数比；Uin为供电电源Vin的输入电压大小。
5.根据权利要求2所述的快速充放电双向反激电路，其特征在于，在放电时间内，原边开关管MOSFET Q1保持关断状态，副边开关管MOSFET Q2根据控制电路产生的PWM波形进行开通关断操作；其中：
当副边开关管MOSFET Q2开通时，输出电容Cout的电压施加在高频反激变压器的副边，流过高频反激变压器副边的电流Is线性增大，此时原边感应出上负下正的电压，原边开关管MOSFET Q1保持关断状态，其内部寄生二极管保持反向阻断，原边没有感应电流；当控制电路检测到高频反激变压器副边电流Is峰值达到设定值Is peak，副边开关管MOSFET Q2关断，此时原边感应出上正下负的电压，储存在高频反激变压器中的能量按照：原边开关管MOSFETQ1-励磁变感Lm-初级高压储能电容CHV的流向进行泄放，原边励磁电流从峰值Ippeak线性下降，直到为零，此时副边开关管MOSFET Q2再次导通，新的周期反复进行，直到输出电压降低为零。
6.根据权利要求5所述的快速充放电双向反激电路，其特征在于，在放电时间内，该快速充放电双向反激电路的放电时间按式(2)计算：
Figure FDA0002229486220000031
nos为放电过程所需的开关周期数；N为高频反激变压器的匝数比；Uin为供电电源Vin的输入电压大小。
7.根据权利要求1～6中任意一项所述的快速充放电双向反激电路，其特征在于，前级升压辅助电路中还包括与初级高压储能电容CHV并联设置的放电电阻R1。
8.根据权利要求1～6中任意一项所述的快速充放电双向反激电路，其特征在于，选用二极管替换开关管MOSFET Q3，实现升压效果。
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