CN105141340A - Full-digital receiving method of direct spread MSK signal - Google Patents

Abstract

The invention provides a full-digital receiving method of a direct spread MSK signal. The full-digital receiving method comprises constructing the direct spread MSK signal into an approximate direct spread BPSK signal, acquiring the constructed approximate direct spread BPSK signal by use of a two-dimensional combined acquisition algorithm of pseudo-code phases and carrier Doppler frequency shift based on Doppler effect correction-FFT, tracking the acquired pseudo-code phases and carrier phases, and performing processing, such as de-spreading, demodulation and detection, on a synthesized signal to recover transmitted data.

Description

The digital method of reseptance of DS msk signal

Technical field

The present invention relates to digital communication technology field, the present invention relates to a kind of digital communication technology, particularly a kind of digital method of reseptance of DS msk signal.

Background technology

At present, spread spectrum is many based on BPSK, QPSK modulation system, there is the application of serious nonlinear distortion, Doppler frequency shift and multipath fading, and DS-BPSK signal/QPSK system just cannot be suitable for.DS msk signal combine the advantage such as low intercepting and capturing, the random address ability of multi-user, anti-interference of spread spectrum system and minimum frequency shift keying signal constant-envelope, the availability of frequency spectrum is high, concentration of energy, side lobe attenuation fast, out-of-band radiation power is low, to advantages such as nonlinear distortion are insensitive, be thus widely applied in fields such as tactical data link, civil aviation Ground-to-Air Data Link, missile guidance command, satellite communications.If U.S. army is the joint tactical information distribution system (JTIDS) that Joint Army-Navy-Air Force operation is developed, employing be exactly the working method of " frequency hopping+spread spectrum+MSK modulates ", this system is widely used in multiple weapon platform.

The final goal of DS msk signal receiver process is that despread-and-demodulation goes out to send data, catching of pseudo-code phase and carrier frequency is the precondition of despread-and-demodulation, and receiver is in the past catch and carrier doppler capturing frequency deviation carrying out pseudo-code phase respectively after the DS msk signal down-converted received mostly.Traditional method for acquiring pseudo code has sliding correlation method and matching matrix, between acquisition speed and implementation complexity, there is contradiction.Simultaneously under high dynamic environment, usually there is very high radial velocity between receiver and transmitter, this makes Received signal strength there is tens the KHz even Doppler frequency shift of hundreds of KHz, to spreading code catch and tracking has a huge impact.Therefore, must catch carrier Doppler frequency and compensate, this be also called carrier wave and intercepts and captures.So the two dimension become pseudo-code phase and carrier doppler frequency deviation of catching of pseudo-code signal is caught, this just needs the capture time grown very much.So, under high dynamic environment, how DS msk signal accurately and is rapidly caught and become technological difficulties.

The people such as G.J.R.Povey first proposed the Capturing Models based on numerical portion matched filter and FFT combination algorithm (PMF-FFT), although this method alleviates the impact of Doppler shift on detection threshold to a certain extent, and the two dimension achieving pseudo-code phase and carrier wave frequency deviation is caught, but this method is mainly adapted to mpsk signal, and the capture range of Doppler shift is less, still inapplicable under high dynamic environment.Thus, under high dynamic environment, how to complete spread-spectrum signal accurately and rapidly pseudo-code phase and Doppler shift two dimension are caught, and become the key technology of DS msk signal all-digital receiver.

Summary of the invention

The object of the present invention is to provide the digital method of reseptance of a kind DS msk signal, the DS msk signal of reception can be configured to a kind of approximate DS-BPSK signal signal form, then under large Doppler's Frequency Offset, the two-dimentional joint acquisition algorithm of the pseudo-code phase of DS-BPSK signal signal and carrier doppler frequency deviation can be adopted, realize the catching and following the tracks of under high dynamic environment of DS msk signal, avoid the complex process of directly DS msk signal being caught, follows the tracks of.

A kind of digital method of reseptance of DS msk signal, comprise: DS msk signal is configured to approximate DS-BPSK signal signal, adopt and based on the pseudo-code phase of Doppler effect correction-FFT and the two-dimentional joint acquisition algorithm of carrier doppler frequency deviation, posttectonic approximate DS-BPSK signal signal is caught, pseudo-code phase after catching and carrier phase are followed the tracks of, and by process such as despreading, demodulation, detections, transmission data are recovered to the signal after synchronous.Described approximate DS-BPSK signal signal constructs carrying out extracting also alternate samples after if sampling, the Digital Down Convert DS msk signal after low-pass filtering is multiplied with wave function.The two-dimentional joint acquisition algorithm of the described pseudo-code phase based on Doppler effect correction-FFT and carrier doppler frequency deviation comprises: based on the doppler filtering of FFT; Based on the Doppler effect correction of the local pseudo-code of phase compensating factor and loopy moving; Based on the pseudo-code phase parallel capture of section FFT.

Adopt the digital method of reseptance of above-mentioned DS msk signal, the structure of described approximate DS-BPSK signal signal specifically comprises:

I, Q two-way digital orthogonal baseband signal obtained after low-pass filtering respectively with wave function be multiplied and obtain four tunnel output signals, wherein T _cfor the chip width of spreading code;

P times (P is over-sampling multiple) extraction is carried out to four tunnel output signals and obtains four road signals, be respectively y _i1(k '), y _q1(k '), y _i2(k '), y _q2(k '); Wherein y _i1(k ') and y _q1(k ') be respectively I roadbed band signal, Q roadbed band signal and wave function the signal obtained is extracted, y after being multiplied _i2(k ') and y _q2(k ') be respectively Q roadbed band signal, I roadbed band signal and wave function the signal obtained is extracted after being multiplied; Wherein k '>=0, and be integer, when k ' is y during odd number _i1(k ') and y _q1(k ') be 0, when k ' is y during even number _i2(k '), y _q2(k ') be 0;

To y _i1(k ') and-y _i2(k ') alternate samples as I road, to y _q1(k ') and y _q2the complex signal I+jQ that I, Q two paths of signals combines, as Q road, is approximate DS-BPSK signal signal by (k ') alternate samples.

Adopt the digital method of reseptance of above-mentioned DS msk signal, before pairing approximation DS-BPSK signal signal is caught, carry out following process: to posttectonic approximate DS-BPSK signal signal, gather the spread-spectrum signal of K PN-code capture, PN-code capture is N; K*N sampling point is reordered, obtains the new sequence that N segment length is K.

Adopt the digital method of reseptance of above-mentioned DS msk signal, be K point FFT by the sequence being K to each segment length, carry out doppler filtering.

Adopt the digital method of reseptance of above-mentioned DS msk signal, Doppler shift compensates and comprises: Doppler shift be expressed as f _cfor spread-spectrum code rate, Δ f is inherent spurious frequency deviation, wherein each concrete f _donly corresponding one group (k, m), wherein k and m is integer, k≤K-1, m>=0; By being multiplied by corresponding phase compensating factor to the N section FFT result after doppler filtering compensate wherein i=0,1 ... N-1, j are imaginary part unit; By local pseudo-code being multiplied by one in time domain frequency compensation part, is equivalent to the m position that to be moved to right by the FFT sequence loops of local pseudo-code on frequency domain; By suitably arranging the value of K, inherent spurious frequency deviation Δ f is made to be not more than f _c/ 2KN, thus do not affect pseudorandom codes phase estimation below.

Adopt the digital method of reseptance of above-mentioned DS msk signal, adopt section FFT to carry out pseudo-code phase parallel capture, comprise

Carry out secondary to the output signal after the shifting processing of doppler filtering, phase compensation and pseudo-code FFT sequence to reorder and revert to original order, namely the sequence that K segment length is N is rearranged to, and the pseudo-code phase parallel capture based on FFT is carried out to the sequence that every segment length is N, obtain the correlation output result of pseudo-code;

Based on CFAR criterion, decision threshold is set;

Adopt envelope detected method, mould asked to correlation output result, and peak value and decision threshold are compared:

(1) if peak value exceedes decision threshold, then pseudo-code phase acquisition success;

(2) if peak value does not exceed decision threshold, first paragraph data in K segment data are abandoned, all the other K-1 segment datas move forward successively, continuation is received N point data below, after rearrangement, obtain the sequence of new N segment length for K point, continue to catch pseudo-code phase, until peak value exceedes decision threshold.

DS msk signal two dimension joint acquisition method of the present invention has the following advantages: (1), due to the particularity of DS msk signal form, compared with having a lot of ripe acquisition algorithm with DS-BPSK signal signal, the catching method of DS msk signal is relatively less.After DS msk signal being converted to a kind of approximate DS-BPSK signal signal form, just can use DS msk signal after the ripe algorithm of DS-BPSK signal is modified, reduce development difficulty and the cost of DS MSK all-digital receiver.(2) catching method in the present invention is compensated large Doppler shift by the FFT sequence of the local pseudo-code of loopy moving, doppler range can be expanded easily, and do not reduce the performance that two dimension catches, be thus applicable to very much the various application under high dynamic environment.(3) pseudo-code phase and carrier wave frequency deviation are searched for simultaneously, decrease capture time, the application demand of high data rate spread spectrum communication can be met.

Below in conjunction with Figure of description, the present invention is described further.

Accompanying drawing explanation

Fig. 1 is the inventive method flow chart.

Fig. 2 is the 3-D graphic of the trapping module correlation output of the embodiment of the present invention.

Fig. 3 is the detection probability of the embodiment of the present invention and the false alarm probability change curve with input signal-to-noise ratio.

Fig. 4 is the all-digital receiver system composition diagram of the embodiment of the present invention.

Embodiment

The intermediate-freuqncy signal of the present invention's input can be expressed as

Wherein, γ _ifor the data symbol after i-th spread spectrum of transmission, there is γ _i=± 1, d _ifor the data sequence sent, c _ifor pseudo-code sequence, in | i| _nfor | i| modulo-N arithmetic, represents current chip sequence number, and N is the spreading code cycle, T _c=T/M is the chip width of spreading code, and M is spreading factor, and T is data symbol width before spread spectrum, f _c=1/T _cfor spread-spectrum code rate, f _ifor IF-FRE, ε is the number of chips that the pseudo-code signal received differs with local pseudo-code, ε T _cfor the pseudo-code signal of reception and the phase difference of local pseudo-code, θ is the first phase of Signal reception moment wave function, and θ and ε meets certain corresponding relation, namely for the first phase of Signal reception moment carrier function, f _dfor the difference of the frequency of input signal and local reference signal, produce primarily of Doppler effect, be therefore called Doppler frequency.The object of receiver acquisition obtains ε, f exactly _destimated value.

To the intermediate-freuqncy signal inputted with sample frequency f _s=P/T _cafter (P is over-sampling multiple) carries out if sampling, Digital Down Convert process, obtaining I, Q two-way baseband signal is

Composition graphs 1, a kind of digital method of reseptance of DS msk signal, comprises the following steps:

Step S101, is configured to approximate DS-BPSK signal signal by DS msk signal;

Step S102, adopts and catches posttectonic approximate DS-BPSK signal signal based on the pseudo-code phase of Doppler effect correction-FFT and the two-dimentional joint acquisition algorithm of carrier doppler frequency deviation;

Step S103, follows the tracks of the pseudo-code phase after catching and carrier phase;

Step S104, recovers transmission data to the signal after synchronous by process such as despreading, demodulation, detections.

In step S101, in order to eliminate the impact of modulation waveform function, by this two-way baseband signal respectively with wave function be multiplied, obtain four road signals, next P is carried out to four road signals and doubly extract, obtain with T _cfor the four road signals in sampling interval.Four roadbed band signals can be expressed as:

γ _i, γ _qγ respectively _ithe even order that after carrying out serial to parallel conversion, interpolation twice obtains and odd numbered sequences, and γ _qcompare γ _ipostpone one.Carry out P times (P is over-sampling multiple) extraction to four tunnel output signals and obtain four road signals, the present invention adopts 12 times, is respectively y _i1(k '), y _q1(k '), y _i2(k '), y _q2(k '); Wherein y _i1(k ') and y _q1(k ') be respectively I roadbed band signal, Q roadbed band signal and wave function the signal obtained is extracted, y after being multiplied _i2(k ') and y _q2(k ') be respectively Q roadbed band signal, I roadbed band signal and wave function the signal obtained is extracted after being multiplied.

According to γ _i, γ _qγ respectively _ithe even order that after carrying out serial to parallel conversion, interpolation twice obtains and odd numbered sequences, and γ _qcompare γ _ipostpone the rule of, to y _i1,-y _i2alternate samples as I road, to y _q1, y _q2i, Q two paths of signals, as Q road, is combined complex signal I+jQ by alternate samples, can obtain output signal to be

Formula (8) is the approximate DS-BPSK signal signal converted to by DS msk signal.

In step s 102, known by the autocorrelation performance of Gold sequence, when being mated with pseudo random sequence by r (k), two, the front and back in above formula are independent of each other, and make s _k=γ (k-ε T _c) cos θ+j γ (k-ε T _c-T _c) sin θ carrys out the two dimension how analysis receiver to carry out pseudo-code phase and carrier doppler frequency deviation under large Doppler shift and catch.

Receive K data symbol, comprise the pseudo-code sequence that one-period is N inside each data symbol, namely receiving a length is the sequence of K × N, and sampling point can be expressed as r _0,0, r _1,0... r _k-1,0, r _0,1, r _1,1... r _k-1,1..., r _{0, N-1}, r _{1, N-1}..., r _{k-1, N-1}, be reorder after interval is extracted to this sequence with N, obtain the new sequence that N segment length is K, putting in order as r of new sequence _0,0, r _0,1... r _{0, K-1}, r _1,0, r _1,1... r _{1, K-1}..., r _n-1,0, r _n-1,1..., r _{n-1, K-1}, in order to observe more intuitively, the form of this data sequence matrix A is expressed as

$A=\left[\begin{array}{cccc}{r}_{0,0}& r_{1,0}& ...& r_{N-1,0}\\ r_{0,1}& r_{1,1}& ...& r_{N-1,1}\\ \begin{array}{c}.\\ .\\ .\end{array}& \begin{array}{c}.\\ .\\ .\end{array}& \begin{array}{c}.\\ .\\ .\end{array}& \begin{array}{c}.\\ .\\ .\end{array}\\ r_{0,K-1}& r_{1,K-1}& ...& r_{N-1,K-1}\end{array}\right]=\left[\begin{array}{cccc}{s}_0& s_1{e}^{{\mathrm{j\ω}}_d\frac{1}{f_c}}& ...& s_{N-1}{e}^{{\mathrm{j\ω}}_d\frac{N-1}{f_c}}\\ s_0{e}^{{\mathrm{j\ω}}_d\frac{N}{f_c}}& s_1{e}^{{\mathrm{j\ω}}_d\frac{N+1}{f_c}}& ...& s_{N-1}{e}^{{\mathrm{j\ω}}_d\frac{2N-1}{f_c}}\\ \begin{array}{c}.\\ .\\ .\end{array}& \begin{array}{c}.\\ .\\ .\end{array}& \begin{array}{c}.\\ .\\ .\end{array}& \begin{array}{c}.\\ .\\ .\end{array}\\ s_0{e}^{{\mathrm{j\ω}}_d\frac{\left(K-1\right)N}{f_c}}& s_1{e}^{{\mathrm{j\ω}}_d\frac{\left(K-1\right)N+1}{f_c}}& ...& s_{N-1}{e}^{{\mathrm{j\ω}}_d\frac{KN-1}{f_c}}\end{array}\right]---\left(9\right)$

N section sequence after N row correspondence in matrix reorders, carries out doppler filtering to this N segment data, and namely carry out K point FFT respectively to each row of matrix A and convert, the form of the result matrix B obtained is expressed as

$B=\left[\begin{array}{cccc}B\left(0,0\right)& B\left(1,0\right)& ...& B\left(N-1,0\right)\\ B\left(0,1\right)& B\left(1,1\right)& ...& B\left(N-1,1\right)\\ \begin{array}{c}.\\ .\\ .\end{array}& \begin{array}{c}.\\ .\\ .\end{array}& \begin{array}{c}.\\ .\\ .\end{array}& \begin{array}{c}.\\ .\\ .\end{array}\\ B\left(0,K-1\right)& B\left(1,K-1\right)& ...& B\left(N-1,K-1\right)\end{array}\right]---\left(10\right)$

Each element of formula (10) matrix B can be expressed as

$B(i,k)=\underset{n=0}{\overset{K-1}{\Σ}}{s}_i{e}^{{\mathrm{j\ω}}_d\frac{nN+i}{f_c}}{e}^{-j\frac{2\mathrm{\π}}{K}nk}=s_i{e}^{{\mathrm{j\ω}}_d\frac{i}{f_c}}\underset{n=0}{\overset{K-1}{\Σ}}{e}^{{\mathrm{j\ω}}_d\frac{nN}{f_c}-j\frac{2\mathrm{\π}}{K}nk},i=0,1,...N-1,k=0,1,...K-1---\left(11\right)$

Wherein, ω _d=2 π f _dthe doppler angle frequency of Received signal strength, each concrete ω _donly corresponding one group (k, m) wherein k and m be integer, k≤K-1, m>=0, makes set up, by ω _dbe updated to formula (11) to obtain when Received signal strength Doppler frequency is on FFT frequency, there is Δ ω=0,

The row k of matrix B is regarded as a centre frequency is doppler filtering passage, when Received signal strength frequency exists time neighbouring, through doppler filtering, the major part of signal spectrum exports at a kth filtering channel, only has a very little part can leak into other passages.Visible, on most energy of useful signal are all the concentrated on corresponding Doppler's passage of doppler filtering to received signal, again because the power spectrum of white Gaussian noise is all equally distributed over the entire frequency band, its energy can be uniformly distributed in each channel, and therefore doppler filtering also has certain noise removal function.But filtered output still comprises Doppler shift, the present invention adopts the method for phase compensation to process each passage, and concrete way is exported by Doppler filter to be multiplied by corresponding phase compensating factor.Phase compensating factor Matrix C can be expressed as

$C=\left[\begin{array}{cccc}1& 1& ...& 1\\ 1& e^{-j\frac{2\mathrm{\π}}{KN}}& ...& e^{-\frac{2\mathrm{\π}}{KN}(N-1)}\\ \begin{array}{c}.\\ .\\ .\end{array}& \begin{array}{c}.\\ .\\ .\end{array}& \begin{array}{c}.\\ .\\ .\end{array}& \begin{array}{c}.\\ .\\ .\end{array}\\ 1& e^{-j\frac{2\mathrm{\π}(K-1)}{KN}}& ...& e^{-j\frac{2\mathrm{\π}(K-1)}{KN}(N-1)}\end{array}\right]---\left(12\right)$

Output matrix D=B*C after phase compensation is

$D=\left[\begin{array}{cccc}B\left(0,0\right)& B\left(1,0\right)& ...& B\left(N-1,0\right)\\ B\left(0,1\right)& B\left(1,1\right)e^{-j\frac{2\mathrm{\π}}{KN}}& ...& B\left(N-1,1\right)e^{-j\frac{2\mathrm{\π}}{KN}\left(N-1\right)}\\ \begin{array}{c}.\\ .\\ .\end{array}& \begin{array}{c}.\\ .\\ .\end{array}& \begin{array}{c}.\\ .\\ .\end{array}& \begin{array}{c}.\\ .\\ .\end{array}\\ B\left(0,K-1\right)& B\left(1,K-1\right)e^{-j\frac{2\mathrm{\π}\left(K-1\right)}{KN}}& ...& B\left(N-1,K-1\right)e^{-j\frac{2\mathrm{\π}\left(K-1\right)}{KN}\left(N-1\right)}\end{array}\right]---\left(13\right)$

In doppler filtering process, the frequency resolution of FFT conversion is Δ f=f _c/ KN is [0, f without blurred bandwidth _c/ N], when Doppler shift drop on without within the scope of blurred bandwidth time, by doppler filtering and phase compensation, if occur that relevant peaks exports on a kth passage, according to described k and ω above _done-to-one relationship, just can obtain the Doppler frequency of Received signal strength.But under high dynamic environment, Doppler shift generally can reach tens even hundreds of kHz, beyond without blurred bandwidth, high dynamically under Doppler shift can be expressed as $f_d=\frac{k}{KN}{f}_c+\frac{m}{N}{f}_c+\mathrm{\Δ}f,m\≠0,$ for inherent spurious frequency deviation fixing after doppler filtering and phase compensation, the correlation peak that its existence can reduce pseudorandom codes phase estimation causes capturing correct pseudo-code phase, and local pseudo-code is multiplied by one by time domain frequency component can eliminate this fixing inherent spurious frequency deviation, be equivalent to the m position that to be moved to right by the FFT sequence loops of local pseudo-code on frequency domain, the output always had after process like this in a passage is only left to be not more than f _cthe frequency difference Δ f of/2KN, so just can make this residual frequency Δ f can not affect pseudorandom codes phase estimation below by the value suitably arranging K.Visible, hunting zone is relevant with the displacement situation of local pseudo-code FFT transformation results, if maximum mobile n left _lposition, maximum mobile n to the right _rposition, so effective hunting zone of frequency becomes the Doppler shift hunting zone that supposing the system requires is [-f _d, f _d], so can get wherein for rounding up, represent and round downwards.

There is data symbol saltus step in posttectonic approximate DS-BPSK signal signal, in a PN-code capture, the polarity of some pseudo noise code changes and destroys its autocorrelation performance, thus catch pseudo-code phase and have an impact.In order to eliminate the impact of symbol saltus step, posttectonic signal is carried out delay multiplication.Signal Doppler frequency after delay multiplication becomes original twice, so the dynamic range of Doppler frequency also will become original twice when parameters, and will be only real Doppler frequency except 2 after estimating Doppler frequency.

After the shifting processing of doppler filtering, phase compensation and pseudo-code FFT sequence, kth the multi-channel output signal corresponding with signal Doppler frequency is

$D(i,k)=s_i{e}^{j\left(\frac{2{\mathrm{\πmf}}_c}{N}+\mathrm{\Δ}\mathrm{\ω}\right)\frac{i}{f_c}}\underset{n=0}{\overset{K-1}{\Σ}}{e}^{j\mathrm{\Δ}\mathrm{\ω}\frac{nN}{f_c}},i=0,1,...,N-1---\left(14\right)$

Next, carry out secondary to the output signal after process to reorder and revert to original order, namely the sequence that K segment length is N is rearranged to, and the pseudo-code phase parallel capture based on FFT is carried out to the sequence that every segment length is N, namely the whole sequence pseudo-code phase parallel capture done based on section FFT is obtained to the correlation output result of pseudo-code.

Pseudo-code phase parallel capture based on FFT adopts twice FFT to add an IFFT to replace matched filtering, local pseudo-code sequence { c _ifFT loopy moving to the right the result of position can regard sequence as carry out the result of FFT.The final two dimension based on Doppler effect correction-FFT is caught Output rusults and is:

$\begin{array}{c}z\left(\mathrm{\ϵ},k,\hat{m}\right)=IFFT\left\{FFT\left\{B\left(i,k\right)e^{-j\frac{2\mathrm{\π}\left(k-1\right)}{KN}\left(i-1\right)}\right\}\·{\mathrm{FFT}}^{*}\left\{c_i{e}^{j\frac{2\mathrm{\π}\hat{m}{f}_c}{N}\·\frac{i}{f_c}}\right\}\right\}\\ =\underset{i=0}{\overset{N-1}{\Σ}}{s}_{i-\mathrm{\ϵ}}{e}^{j\left(\frac{2{\mathrm{\πmf}}_c}{N}+\mathrm{\Δ}\mathrm{\ω}\right)\frac{i}{f_c}}{c}_i{e}^{-j\frac{2\mathrm{\π}\hat{m}{f}_c}{N}\·\frac{i}{f_c}}\underset{n=0}{\overset{K-1}{\Σ}}{e}^{j\mathrm{\Δ}\mathrm{\ω}\frac{nN}{f_c}}\end{array}---\left(15\right)$

When time, have

$z\left(\mathrm{\τ},k,m\right)=\underset{i=0}{\overset{N-1}{\Σ}}{s}_{i-\mathrm{\ϵ}}{c}_i{e}^{j\mathrm{\Δ}\mathrm{\ω}\frac{i}{f_c}}\underset{n=0}{\overset{K-1}{\Σ}}{e}^{j\mathrm{\Δ}\mathrm{\ω}\frac{nN}{f_c}}=\underset{n=0}{\overset{K-1}{\Σ}}\underset{i=0}{\overset{N-1}{\Σ}}{s}_{i-\mathrm{\ϵ}}{c}_i{e}^{j\mathrm{\Δ}\mathrm{\ω}\frac{i+nN}{f_c}}=\underset{i^{\′}=0}{\overset{KN-1}{\Σ}}{s}_{i^{\′}-\mathrm{\ϵ}}{c}_{i^{\′}}{e}^{j\mathrm{\Δ}\mathrm{\ω}\frac{i^{\′}}{f_c}}---\left(16\right)$

The method that acquiring pseudo code in native system adopts envelope detected and CFAR thresholding to arrange.Mould is asked to the Output rusults of IFFT, and correlation peak and decision threshold are compared, if correlation peak is greater than decision threshold, then represent pseudo-code phase acquisition success, frequency residing for peak value is exactly the estimated value of Doppler frequency, and residing phase place is exactly the estimated value of pseudo-code phase; If correlation peak does not exceed decision threshold, then represent there is no acquisition success, first paragraph data in K segment data are abandoned, all the other K-1 segment datas move forward successively, after continuation being received the rearrangement of N point data below, obtain new N segment length and proceed fast Acquisition process for the sequence of K point, until correlation peak is greater than decision threshold, completes pseudo-code phase and catch.

In step S103 and step S104, above-mentioned fast Acquisition processing module has carried out rough estimate to pseudo-code phase and Doppler shift, want despreading from Received signal strength, demodulate transmission data, also to carry out essence synchronous, namely pseudo-code phase and carrier phase are followed the tracks of, the tracking that the present invention adopts in advance-hysteresis delay phase-locked loop, Costas loop realizes pseudo-code phase and carrier phase respectively.After completing pseudo-code phase and carrier frequency synchronization, enter into despreading, separate mediation detection module, finally recover transmission data symbol.

The present invention adopts the system shown in Fig. 4 to receive DS msk signal, system sampling frequency f _s=245.52MHz, IF-FRE is 76.725MHz, over-sampling multiple P=12, and spread-spectrum code rate is 20.46Mchip/s, data rate is 20kbps, spreading code adopts Gold sequence, code length N=1023, the code periodicity of two-dimensional search, namely K value is 32, Doppler frequency range is [-200KHz, 200KHz], therefore n _l=20, n _r=20, frequency search stepping is 625Hz, and pseudo-code searching stepping is a chip.

Composition graphs 2, input signal-to-noise ratio SNR=-15dB, pseudo-code phase ε=478.4chip, Doppler frequency f _dduring=169.85KHz, carry out the two-dimensional search of pseudo-code phase and carrier doppler frequency deviation, obtain the 3-D graphic of normalization correlation output.As can be seen from the figure the estimated value of pseudo-code phase differs 0.4chip with actual value, within the scope of half-chip; The estimated value of carrier wave frequency deviation differs 162.5Hz with actual value, in the scope of maximum residual frequency difference.The digital method of reseptance of visible the present invention's design can carry out catching fast and accurately to the pseudo-code phase of DS msk signal and carrier wave frequency deviation under high dynamic environment.

Composition graphs 3, input signal-to-noise ratio is SNR=[-30dB ,-5dB], f _d=40KHz, false alarm probability P _fwhen=0.01, carry out 5000 times and catch, obtain detection probability and the false alarm probability change curve with signal to noise ratio.Found that after signal to noise ratio reaches-17dB, detection probability levels off to 1, and owing to have employed CFAR thresholding design standard, false alarm probability is hardly by the impact of signal to noise ratio.

Claims (8)

1. a digital method of reseptance for DS msk signal, is characterized in that, comprise

DS msk signal is configured to approximate DS-BPSK signal signal,

Adopt and based on the pseudo-code phase of Doppler effect correction-FFT and the two-dimentional joint acquisition algorithm of carrier doppler frequency deviation, posttectonic approximate DS-BPSK signal signal caught,

Pseudo-code phase and carrier phase are followed the tracks of,

By process such as despreading, demodulation, detections, transmission data are recovered to the signal after synchronous.

Described approximate DS-BPSK signal signal constructs carrying out extracting also alternate samples after if sampling, the Digital Down Convert DS msk signal after low-pass filtering is multiplied with wave function;

The two-dimentional joint acquisition algorithm of the described pseudo-code phase based on Doppler effect correction-FFT and carrier doppler frequency deviation comprises:

Based on the doppler filtering of FFT;

Based on the Doppler effect correction of the local pseudo-code of phase compensating factor and loopy moving;

Based on the pseudo-code phase parallel capture of section FFT.

2. the digital method of reseptance of DS msk signal according to claim 1, is characterized in that, the structure of described approximate DS-BPSK signal signal specifically comprises:

I, Q two-way digital orthogonal baseband signal obtained after low-pass filtering respectively with wave function be multiplied and obtain four tunnel output signals, wherein T _cfor the chip width of spreading code;

Carry out P to four tunnels output signals doubly to extract (P is over-sampling multiple) and obtain four road signals, be respectively y _i1(k '), y _q1(k '), y _i2(k '), y _q2(k '); Wherein y _i1(k ') and y _q1(k ') be respectively I roadbed band signal, Q roadbed band signal and wave function the signal obtained is extracted, y after being multiplied _i2(k ') and y _q2(k ') be respectively Q roadbed band signal, I roadbed band signal and wave function the signal obtained is extracted after being multiplied; Wherein k '>=0, and be integer, when k ' is y during odd number _i1(k ') and y _q1(k ') be 0, when k ' is y during even number _i2(k '), y _q2(k ') be 0;

To y _i1(k ') and-y _i2(k ') alternate samples as I road, to y _q1(k ') and y _q2the complex signal I+jQ that I, Q two paths of signals combines, as Q road, is approximate DS-BPSK signal signal by (k ') alternate samples.

3. the digital method of reseptance of DS msk signal according to claim 2, is characterized in that, before pairing approximation DS-BPSK signal signal is caught, carries out following process:

To posttectonic approximate bpsk signal, gather the spread-spectrum signal of K PN-code capture, PN-code capture is N;

K*N sampling point is reordered, obtains the new sequence that N segment length is K.

4. the digital method of reseptance of DS msk signal according to claim 3, is characterized in that, is K point FFT, carries out doppler filtering by the sequence being K to each segment length.

5. the digital method of reseptance of DS msk signal according to claim 3, it is characterized in that, Doppler effect correction comprises:

Doppler shift is expressed as f _cfor spread-spectrum code rate, Δ f is inherent spurious frequency deviation, wherein each concrete f _donly corresponding one group (k, m), wherein k and m is integer, k≤K-1, m>=0;

By being multiplied by corresponding phase compensating factor to the N section FFT result after doppler filtering compensate wherein i=0,1 ... N-1, j are imaginary part unit;

By local pseudo-code being multiplied by one in time domain frequency compensation be equivalent to the m position that to be moved to right by the FFT sequence loops of local pseudo-code on frequency domain;

By suitably arranging the value of K, inherent spurious frequency deviation Δ f is made to be not more than f _c/ 2KN, thus do not affect pseudorandom codes phase estimation below.

6. the digital method of reseptance of DS msk signal according to claim 3, is characterized in that, adopts section FFT to carry out pseudo-code phase parallel capture, comprising:

Carry out secondary to the output signal after the shifting processing of doppler filtering, phase compensation and pseudo-code FFT sequence to reorder and revert to original order, namely the sequence that K segment length is N is rearranged to, and the pseudo-code phase parallel capture based on FFT is carried out to the sequence that every segment length is N, obtain the correlation output result of pseudo-code;

Based on CFAR criterion, decision threshold is set;

Adopt envelope detected method, mould asked to correlation output result, and peak value and decision threshold are compared:

(1) if peak value exceedes decision threshold, then pseudo-code phase acquisition success;

(2) if peak value does not exceed decision threshold, first paragraph data in K segment data are abandoned, all the other K-1 segment datas move forward successively, continuation is received N point data below, after rearrangement, obtain the sequence of new N segment length for K point, continue to catch pseudo-code phase, until peak value exceedes decision threshold.

7. the digital method of reseptance of DS msk signal according to claim 1, is characterized in that, adopts advanced-hysteresis delay locking ring, tracking that Costas loop realizes pseudo-code phase and carrier phase respectively.

8. the digital method of reseptance of DS msk signal according to claim 1, is characterized in that, employing coherent demodulation, feedback judgement detection method are carried out process to the msk signal after despreading and recovered transmission data.