CN104536016A - GNSS new-system signal capturing device and method - Google Patents

Abstract

The invention discloses a GNSS new-system signal capturing device and method. The device comprises a difference correlator module, a carrier wave NCO generator, a local PN code generator, a correlator, a frequency mixer, a segment correlation zero-fill FFT module and a capture detection control module. The difference correlator module determines the PN code phase offset of a received satellite signal via a digital intermediate-frequency input signal, a difference output signal and a local carrier wave are mixed and demodulated to the base band, an in-phase branch I and an orthogonal branch Q are obtained via the correlator and the local PN code generator, the segment correlation zero-fill FFT module carries out coherent integration and segment noncoherent accumulation on operation results, and after zero-filling of the R-point accumulation result, S-point FFT is carried out to complete rapidly capture the GNSS new-system signals. The PN code phase offset of the received satellite signal can be rapidly determined, the Doppler frequency estimation precision can be improved via segment correlation zero-fill FFT, the device is simple, the false capture rate is low, and new-system signals can be rapidly captured.

Description

A kind of GNSS New System signal capture device and method

Technical field

The invention belongs to signal of global navigation satellite system processing technology field, be specifically related to a kind of GNSS New System signal capture device and method.

Background technology

GLONASS (Global Navigation Satellite System) (Global Navigation Satellite System, GNSS) all plays extremely important status in military and civilian field.GNSS can be receiver and provides location, navigation and time service service, occupies critical role in military and civilian field.Along with the development of navigational system, multi-system information merges positioning calculation and has become current development trend, New System signal is widely used in the Navsat such as the third generation GPS of the U.S., the Galileo of European Union, and the code length of New System signal is 10230 chips, bit rate reaches 10.23Mcps, is the old system navigation signal code length of tradition and bit rate 10 times.If adopt traditional slip serial to be correlated with time-frequency two-dimensional searching algorithm, parallel frequency search algorithm or parallel code phase search algorithm, computational complexity can be there is larger, search time is longer, affect the problems such as receiver primary positioning time and warm start, particularly in high dynamic application environment, constrain the capture rate of receiver.

Therefore, be necessary the quick catching device and the method that propose a kind of GNSS New System signal, reduce the computational complexity in receiver signal acquisition procedure, improve the work efficiency of receiver, become a kind of new technical need.

Summary of the invention

In order to solve the problems referred to above that prior art exists, the invention provides a kind of GNSS New System signal capture device and method.

According to an aspect of the present invention, provide a kind of GNSS New System signal capture device, comprise differential correlator module (101), zero padding FFT module (108) and Acquisition Detection control module (109) are correlated with in frequency mixer I (102), frequency mixer Q (103), carrier wave NCO (104), correlator I (105), correlator Q (106), PN code generator (107), segmentation.

Preferably, described differential correlator module (101) is for determining received digital intermediate frequency input signal PN code phase offset amount; The output signal of differential correlator module (101) is connected with frequency mixer I (102) and frequency mixer Q (103) simultaneously, and the sine produced with described carrier wave NCO (104) respectively, cosine signal are multiplied; Frequency mixer I (102) is connected with correlator I (105), and frequency mixer Q (103) is connected with correlator Q (106); PN code generator (107) is connected with correlator I (105) and correlator Q (106), for the PN code in the satellite-signal that relevant elimination receives simultaneously; Correlator I (105) is connected with correlator Q (106) zero padding FFT relevant to segmentation module (108), for determining the Doppler frequency deviation of receiving satellite signal; Acquisition Detection control module (109) zero padding FFT relevant to segmentation module (108) connects, and to FFT result modulo operation, and compares with the Acquisition Detection amount threshold value preset, and carries out catching result output.

Preferably, described digital intermediate frequency input signal is the digital medium-frequency signal of satellite-signal after down coversion mixing, intermediate-freuqncy signal filter and amplification, analog to digital conversion that satellite navigation receiver receives.

Preferably, described differential correlator module (101) comprises signal lag module (201), multiplier (202) and signal lag control module (203).

Preferably, described signal lag module (201) is connected with multiplier (202), for time delay satellite digital IF input signals.

Preferably, described multiplier (202) is connected with signal lag control module (203).Multiplier (202) is multiplied with the digital intermediate frequency input signal after time delay for the digital medium-frequency signal inputted.Signal lag control module (203) for carrying out square operation to the output signal of multiplier (202), and differentiates result, to determine the PN code phase offset amount of described digital medium-frequency signal.

Preferably, described segmentation is correlated with zero padding FFT module (108) for plural for M point correlated results I+jQ is divided into R section, every segment length is M/R, and respectively noncoherent accumulation is carried out to each Piecewise Correlator, composition R point complex result, and carry out S point FFT computing after mending S-R individual 0, wherein plural correlated results I+jQ is digital medium-frequency signal respectively through formed the afterwards complex result of correlator I (105) and correlator Q (106), j is imaginary unit, M is the number of chips of tracking satellite, the span of R is 2 ~ 512, the span of S is 512 ~ 4096.

Optionally, during when R is the integral multiple of 2 and with S relatively, can not zero padding operation be carried out, directly carry out FFT computing.

According to an aspect of the present invention, provide a kind of GNSS New System signal acquisition methods, described method comprises the steps:

Step S1: satellite navigation receiver receiving satellite signal, carries out down coversion mixing, intermediate frequency filtering amplifies and analog to digital conversion, obtain digital IF input signals;

Step S2: digital intermediate frequency input signal, through differential correlator module, obtains the PN code phase offset amount of tracking satellite;

Step S3: in-phase branch I and quadrature branch signal Q, through comprising the frequency mixer of local carrier NCO, the correlator of local PN code generator, is demodulated into complex number type baseband signal by digital intermediate frequency differential output signal;

Step S4: the complex number type baseband signal in step S3 to be correlated with zero padding FFT module through segmentation, and to Output rusults delivery;

Step S5: delivery is carried out to the result that FFT exports, and compares with the detection threshold value preset, determine Doppler frequency deviation and the PN code phase offset amount of institute's tracking satellite.

Optionally, in described step S1, analog to digital conversion can adopt the ADC of any digit between 4 ~ 12.

Preferably, in described step S2, differential correlator module comprises time delay module, multiplier and signal lag control module, for the sampled point number of order delayed digital IF input signals, be multiplied with original digital intermediate frequency input signal after digital medium-frequency signal after time delay gets conjugation, differentiate after modulo operation, obtain the PN code phase offset amount of institute's tracking satellite.

Preferably, in described step S3, Acquisition Detection control module (109) controls carrier wave NCO frequency mixer (104) and PN code generator (107), generates and mobile local PN code phase offset for controlling local carrier.

Preferably, in described step S4, segmentation is correlated with zero padding FFT module for plural for M point correlated results I+jQ is divided into R section, every segment length is M/R, and respectively noncoherent accumulation is carried out to each Piecewise Correlator, composition R point complex result, and carry out S point FFT computing after mending S-R individual 0, wherein plural correlated results I+jQ is digital medium-frequency signal respectively through formed the afterwards complex result of correlator I (105) and correlator Q (106), j is imaginary unit, M is the number of chips of tracking satellite, the span of R is 2 ~ 512, the span of S is 512 ~ 4096.

Optionally, in described step S5, the detection threshold value preset sets according to the PN code length of tracking satellite signal and received signal power.

The present invention can produce positive beneficial effect, the PN code phase offset amount of receiving satellite signal can be determined fast according to difference correlation module, Doppler frequency accuracy of detection is improve by the segmentation zero padding FFT that is correlated with, whole signal capture device is simple, computational complexity is low, it is low that void catches rate, is applicable to the fast Acquisition application of New System signal.

Accompanying drawing explanation

Fig. 1 shows the GNSS New System signal capture structure drawing of device of the preferred embodiment of the present invention;

Fig. 2 shows the differential correlator function structure chart of the preferred embodiment of the present invention;

The segmentation that Fig. 3 shows the preferred embodiment of the present invention is correlated with zero padding FFT function structure chart;

Fig. 4 shows the GNSS New System signal acquisition methods process flow diagram of the preferred embodiment of the present invention;

Fig. 5 shows the GNSS New System signal capture design sketch of the specific embodiment of the invention.

Embodiment

For making the object, technical solutions and advantages of the present invention clearly understand, below in conjunction with embodiment also with reference to accompanying drawing, the present invention is described in more detail.Should be appreciated that, these describe just exemplary, and do not really want to limit the scope of the invention.In addition, in the following description, the description to known features and technology is eliminated, to avoid unnecessarily obscuring concept of the present invention.

Fig. 1 shows the GNSS New System signal capture structure drawing of device of the preferred embodiment of the present invention.

As shown in Figure 1, the invention provides a kind of GNSS New System signal capture device, this device comprises differential correlator module (101), zero padding FFT module (108) and Acquisition Detection control module (109) are correlated with in frequency mixer I (102), frequency mixer Q (103), carrier wave NCO (104), correlator I (105), correlator Q (106), PN code generator (107), segmentation.

Differential correlator module (101) is for determining received digital intermediate frequency input signal PN code phase offset amount; The output signal of differential correlator module (101) is connected with frequency mixer I (102) and frequency mixer Q (103) simultaneously, and the sine produced with described carrier wave NCO (104) respectively, cosine signal are multiplied; Frequency mixer I (102) is connected with correlator I (105), and frequency mixer Q (103) is connected with correlator Q (106); PN code generator (107) is connected with correlator I (105) and correlator Q (106), for the PN code in the satellite-signal that relevant elimination receives simultaneously; Correlator I (105) is connected with correlator Q (106) zero padding FFT relevant to segmentation module (108), for determining the Doppler frequency deviation of receiving satellite signal; Acquisition Detection control module (109) zero padding FFT relevant to segmentation module (108) connects, and to FFT result modulo operation, and compares with the Acquisition Detection amount threshold value preset, and carries out catching result output.

As shown in table 1 below, GNSS New System signal mainly comprises B1/L1/E1 frequency, and centre frequency is 1575.42MHz, and adopt MBOC modulation system, bit rate is 1.023Mcps, and bandwidth is ± 7.161MHz; B2/E5 frequency adopts frequency centered by 1191.795MHz, and with (TD-) AltBOC for modulation system, bit rate is 10.23Mcps, and bandwidth is ± 25.575MHz; L5/B2a/E5a frequency adopts frequency centered by 1176.45MHz, and with (TDDM-) QPSK for modulation system, bit rate is 10.23Mcps, and bandwidth is ± 10.23MHz; B2b/E5b frequency adopts frequency centered by 1207.14MHz, and with (TDDM-) QPSK for modulation system, bit rate is 10.23Mcps, and bandwidth is ± 10.23MHz; In above frequency, represent BD with the frequency of B beginning, represent GPS with the frequency of L beginning, represent Galileo with the frequency of E beginning.When bit rate reaches 10.23Mcps, traditional acquisition algorithm computational complexity is comparatively large, and capture rate is lower, be difficult to meet high dynamically wait environment catch needs.

Table 1 New System signal madulation parameter

Fig. 2 shows the differential correlator function structure chart of the preferred embodiment of the present invention.

As shown in Figure 2, differential correlator module (101) comprises signal lag module (201), multiplier (202) and signal lag control module (203).Wherein signal lag module (201) is connected with multiplier (202), for time delay satellite digital IF input signals, and is connected with signal lag control module (203).Signal lag control module (203) is connected with signal lag module (201), for the time delay sampled point quantity of control figure IF input signals, and modulo operation and outcome evaluation are carried out to the signal multiplication result through multiplier (202).

Digital intermediate frequency input signal S _iFn () is the digital medium-frequency signal that satellite-signal that satellite navigation receiver receives obtains after down coversion mixing, intermediate-freuqncy signal filter and amplification, analog to digital conversion, after differential correlator module (101), be equivalent to realization wherein, m is S _iFthe number of chips of (n) time delay.Suppose that the complex number type digital medium-frequency signal of certain satellite is wherein, the amplitude of A representation signal, x (n) represents PN code, D (n) representative data code or NH code, f _drepresent carrier doppler frequency deviation, n represents number epoch time of digital medium-frequency signal, T _sthe sampling period of representation signal, then

$\begin{array}{c}{S}_{\mathrm{dif}}\left(n\right)=S_{\mathrm{IF}}\left(n\right)\stackrel{\‾}{S_{\mathrm{IF}}(n-m)}\\ =\mathrm{Ax}\left(n\right)D\left(n\right)e^{j2\mathrm{\π}{f}_{d}n{T}_s}\·\stackrel{\‾}{\mathrm{Ax}(n-m)D(n-m)e^{j2\mathrm{\π}{f}_d(n-m)T_s}}\\ =A^{2}x\left(n\right)x(n-m)D\left(n\right)D(n-m)e^{j2\mathrm{\π}{f}_{d}n{T}_s}\·e^{-j2\mathrm{\π}{f}_d(n-m)T_s}\\ =A^{2}x\left(n\right)x(n-m)D\left(n\right)D(n-m)e^{j2\mathrm{\π}{f}_{d}m{T}_s}\end{array}$

The probability that data bit edges saltus step occurs in acquisition phase for numeric data code or NH code D (n) D (n-m) is 0.001%, so can ignore its impact on difference correlated results.From above formula, to S _difafter (n) result delivery, can judge whether PN code phase offset amount m now aligns with the digital medium-frequency signal received, otherwise by signal lag control module (203) the adjustment phase pushing figure m in differential correlator module (101), till the PN code of the satellite-signal received aligns with the PN code after time delay, the result S now exported _difn () has maximal value.

The segmentation that Fig. 3 shows the preferred embodiment of the present invention is correlated with zero padding FFT function structure chart.

As shown in Figure 3, segmentation is correlated with zero padding FFT module (108) for plural for M point correlated results I+jQ is divided into R section, every segment length is M/R, and respectively noncoherent accumulation is carried out to each Piecewise Correlator, composition R point complex result, and carry out S point FFT computing after mending S-R individual 0, wherein plural correlated results I+jQ is digital medium-frequency signal respectively through formed the afterwards complex result of correlator I (105) and correlator Q (106), j is imaginary unit, and concrete signal calculating process is as described below.

Suppose that Received signal strength is after Digital Down Convert, the digital baseband signal obtained is:

$r\left(k\right)=D\left(k\right)\mathrm{PN}\left(k\right)e^{j(2\mathrm{\π}{f}_{d}k{T}_s-\mathrm{\θ})}$

Wherein D (k) is binary modulated information (as bit information or NH code), and PN (k) is spreading code, i.e. PN code, f _dfor Doppler frequency deviation, T _sfor the sampling period, θ is signal phase.

Be that the baseband signal of M is divided into R section by chip lengths, every segment length is P, P=M/R.I-th section of (i=1,2,3 ... R) signal sampling point is relevant to the local PN code sampling point of corresponding section, when temporarily not considering binary modulated informational influence, obtains correlation:

$\begin{array}{c}C\left(i\right)=\underset{k=(i-1)*P+1}{\overset{i*P}{\mathrm{\Σ}}}\mathrm{PN}\left(k\right)\mathrm{PN}(k+\mathrm{\Δ})e^{j(2\mathrm{\π}{f}_{d}k{T}_s-\mathrm{\θ})}\\ =\underset{k=(i-1)*P+1}{\overset{i*P}{\mathrm{\Σ}}}R\left(\mathrm{\Δ}\right)e^{j(2\mathrm{\π}{f}_{d}k{T}_s-\mathrm{\θ})}\\ =R\left(\mathrm{\Δ}\right)\·e^{j[(2\mathrm{\π}{f}_d(i-1)*P+1)T_s-\mathrm{\θ}]}\frac{1-e^{j\left(2\mathrm{\π}{f}_{d}P{T}_s\right)}}{1-e^{j\left(2\mathrm{\π}{f}_d{T}_s\right)}}\end{array}$

Wherein, the autocorrelation function that R (Δ) is PN code, Δ is the code phase difference of Received signal strength and local PN code.After differential correlator module (101), PN code phase offset amount is determined in this locality, so place only need carry out once-through operation, can estimate Doppler frequency deviation and PN code phase offset value.To carry out the FFT computing of S point after individual for R relevant value complement S-R 0 (under normal circumstances R < S), obtaining S FFT output is:

$\begin{array}{c}{Z}_c\left(m\right)+j*Z_s\left(m\right)=\underset{i=1}{\overset{S}{\mathrm{\&Sigma;}}}C\left(i\right)e^{-j\frac{2\mathrm{\&pi;}}{S}\mathrm{im}}=\underset{i=1}{\overset{R}{\mathrm{\&Sigma;}}}C\left(i\right)e^{-j\frac{2\mathrm{\&pi;}}{S}\mathrm{im}}\\ =\underset{i=1}{\overset{R}{\mathrm{\&Sigma;}}}R\left(\mathrm{\&Delta;}\right)\&CenterDot;\frac{1-e^{j\left(2\mathrm{\&pi;}{f}_{d}P{T}_s\right)}}{1-e^{j\left(2\mathrm{\&pi;}{f}_d{T}_s\right)}}\&CenterDot;e^{-j\left\{2\mathrm{\&pi;}{f}_d\right[(i-1)P+1]T_s-\frac{2\mathrm{\&pi;}}{S}\mathrm{im}-\mathrm{\&theta;}\}}\\ =R\left(\mathrm{\&Delta;}\right)\&CenterDot;\frac{1-e^{j\left(2\mathrm{\&pi;}{f}_d{\mathrm{PT}}_s\right)}}{1-e^{j\left(2\mathrm{\&pi;}{f}_d{T}_s\right)}}\&CenterDot;e^{j[2\mathrm{\&pi;}{f}_d(1-p)T_s-\mathrm{\&theta;}]}\&CenterDot;\frac{1-e^{j2\mathrm{\&pi;R}(f_d{\mathrm{PT}}_s-\frac{m}{S})}}{1-e^{j2\mathrm{\&pi;}(f_s{\mathrm{PT}}_s-\frac{m}{S})}}\\ =R\left(\mathrm{\&Delta;}\right)\&CenterDot;e^{j[2\mathrm{\&pi;}{f}_d(1-p)T_s-\mathrm{\&theta;}]}\&CenterDot;\frac{1-e^{j\left(2\mathrm{\&pi;}{f}_d{\mathrm{PT}}_s\right)}}{1-e^{j\left(2\mathrm{\&pi;}{f}_d{T}_s\right)}}\&CenterDot;\frac{1-e^{j2\mathrm{\&pi;R}(f_d{\mathrm{PT}}_s-\frac{m}{S})}}{1-e^{j2\mathrm{\&pi;}(f_d{\mathrm{PT}}_s-\frac{m}{S})}}(m=\mathrm{1,2},\&CenterDot;\&CenterDot;\&CenterDot;S)\end{array}$

Output rusults delivery is obtained:

$\begin{array}{c}Z\left(m\right)=|{Z^2}_c\left(m\right)+{Z^2}_s\left(m\right)|={|R\left(\mathrm{\&Delta;}\right)\&CenterDot;e^{j[2\mathrm{\&pi;}{f}_d(1-p)T_s-\mathrm{\&theta;}]}\&CenterDot;\frac{1-e^{j\left(2\mathrm{\&pi;}{f}_{d}P{T}_s\right)}}{1-e^{j\left(2\mathrm{\&pi;}{f}_d{T}_s\right)}}\&CenterDot;\frac{1-e^{j2\mathrm{\&pi;R}(f_{d}P{T}_s-\frac{m}{s})}}{1-e^{j2\mathrm{\&pi;}(f_{d}P{T}_s-\frac{m}{S})}}|}^2\\ ={|R\left(\mathrm{\&Delta;}\right)\&CenterDot;\frac{\sin \left(\mathrm{\&pi;}{f}_{d}P{T}_s\right)}{\sin \left(\mathrm{\&pi;}{f}_d{T}_s\right)}\&CenterDot;\frac{\sin \left[\mathrm{\&pi;R}(f_{d}P{T}_s-\frac{m}{S})\right]}{\sin \left[\mathrm{\&pi;}(f_d{\mathrm{PT}}_s-\frac{m}{S})\right]}|}^2\\ =R{\left(\mathrm{\&Delta;}\right)}^2\&CenterDot;{\left|\frac{\sin \left(\frac{M}{R}\mathrm{\&pi;}{f}_d{T}_s\right)}{\sin \left(\mathrm{\&pi;}{f}_d{T}_s\right)}\right|}^2\&CenterDot;{\left|\frac{\sin [\mathrm{\&pi;R}(f_d\frac{M}{R}{T}_s-\frac{m}{S})}{\sin \left[\mathrm{\&pi;}(f_d\frac{M}{R}{T}_s-\frac{m}{S})\right]}\right|}^2\end{array}$

When PN code phase is not synchronous, S the Acquisition Detection amount that R (Δ) ≈ 0, FFT exports all is no more than threshold value, at this moment needs adjustment local code phase place to continue search; And in this programme, differential correlator module (101) has pre-estimated out PN code phase offset amount, so time there is R (Δ) ≈ 1, Acquisition Detection amount is as follows:

$Z\left(m\right)={\left|\frac{\sin \left(\frac{M}{R}\mathrm{\&pi;}{f}_d{T}_s\right)}{\sin \left(\mathrm{\&pi;}{f}_d{T}_s\right)}\right|}^2\&CenterDot;{\left|\frac{\sin \left[\mathrm{\&pi;R}(f_d\frac{M}{R}{T}_s-\frac{m}{S})\right]}{\sin \left[\mathrm{\&pi;}(f_d\frac{M}{R}{T}_s-\frac{m}{S})\right]}\right|}^2$

In detection limit, Section 1 is the snr loss that in integration section, frequency deviation causes, and has nothing to do with m value.Second item size changes with m, when time, corresponding section 2 is made to reach maximum, now prize judgment amount Z (m) ≈ M ², estimate that the Doppler frequency deviation obtained is in addition, for the ease of receiver hardware implementing, the usual value of Piecewise Correlator R is the usual value of 256, S is herein 512.

Fig. 4 shows the GNSS New System signal acquisition methods process flow diagram of the preferred embodiment of the present invention.

As shown in Figure 4, the GNSS New System signal acquisition methods of the preferred embodiment of the present invention comprises the steps:

Step S1: satellite navigation receiver receiving satellite signal, carries out down coversion mixing, intermediate frequency filtering amplifies and analog to digital conversion, obtain digital IF input signals;

Step S2: digital intermediate frequency input signal is through differential correlator module, obtain the PN code phase offset amount of tracking satellite, wherein differential correlator module comprises time delay module, multiplier and signal lag control module, for the sampled point number of order delayed digital IF input signals, be multiplied with original digital intermediate frequency input signal after digital medium-frequency signal after time delay gets conjugation, differentiate after modulo operation, obtain the PN code phase offset amount of institute's tracking satellite;

Step S3: digital intermediate frequency differential output signal is through comprising the frequency mixer of local carrier NCO, the correlator of local PN code generator, and in-phase branch I and quadrature branch signal Q is assembled into complex number type baseband signal, wherein Acquisition Detection control module (109) controls carrier wave NCO frequency mixer (104) and PN code generator (107), generates and mobile local PN code phase offset for controlling local carrier;

Step S4: the complex number type baseband signal in step S3 to be correlated with zero padding FFT module through segmentation, and to Output rusults delivery, wherein segmentation is correlated with zero padding FFT module for plural for M point correlated results I+jQ is divided into R section, every segment length is M/R, and respectively noncoherent accumulation is carried out to each Piecewise Correlator, composition R point complex result, and carry out S point FFT computing after mending S-R individual 0, wherein plural correlated results I+jQ is digital medium-frequency signal respectively through formed the afterwards complex result of correlator I (105) and correlator Q (106), j is imaginary unit, M is the number of chips of tracking satellite, the span of R is 2 ~ 512, the span of S is 512 ~ 4096.

Step S5: delivery is carried out to the result that FFT exports, and compares with the detection threshold value preset, determine Doppler frequency deviation and the PN code phase offset amount of institute's tracking satellite.

Fig. 5 shows the GNSS New System signal capture design sketch of the specific embodiment of the invention.

As shown in Figure 5, in the specific embodiment of the invention, simulated data adopts Galileo E1 signal, and concrete simulation parameter configuration is as follows: the code length of PN code is 4092, and bit rate is 1.023MHz, over-sampling multiple is 5, receive intermediate frequency and be set to 46.42MHz, carrier-to-noise ratio is 40dB/Hz, and base band signal process process Satellite signal maximum Doppler frequency offset is 4000Hz, the Doppler frequency deviation preset is 500Hz, and code phase offset amount is 120 chips.According to the ICD document structure tree PN code of Galileo, after adding intermediate frequency carrier, form digital intermediate frequency input signal.Dispense the mixing of satellite-signal down coversion in this concrete real example, intermediate frequency filtering has amplified and analog-digital conversion process.This digital intermediate frequency input signal is after differential correlator module (101), and the chip phase side-play amount estimated is 120.Suppose that the hardware register of receiver in each calculating process stores a code length of PN code and the digital signal of over-sampling multiple, i.e. 4092*5=20460 point data.20460 I roads and Q road correlation data, after frequency mixer and correlator, after weeding out PN code and intermediate frequency impact, are divided into R section by signal.R=256 is supposed in this concrete real example, S=512, then the length of every section is about 20460/256=80, in each section 80 data are added, form 256 complex number type correlated results, carry out FFT computing after supplementing S-R (i.e. 512-256=256) individual zero, finally to S point processing result delivery, the frequency corresponding to maximal value is the Doppler frequency deviation of Received signal strength.The Doppler frequency deviation estimated in this concrete real example is 505.8347Hz, and analog digital intermediate-freuqncy signal with the addition of white Gaussian noise.In Fig. 5, x-axis and y-axis represent code phase and Doppler frequency deviation respectively, and z-axis represents normalized FFT delivery result, and code corresponding to maximal value is 120chip mutually, and Doppler frequency deviation is 505.8347Hz.

Should be understood that, above-mentioned embodiment of the present invention only for exemplary illustration or explain principle of the present invention, and is not construed as limiting the invention.Therefore, any amendment made when without departing from the spirit and scope of the present invention, equivalent replacement, improvement etc., all should be included within protection scope of the present invention.In addition, claims of the present invention be intended to contain fall into claims scope and border or this scope and border equivalents in whole change and modification.

Claims (9)

1. a GNSS New System signal capture device, comprising differential correlator module (101), zero padding FFT module (108) and Acquisition Detection control module (109) are correlated with in frequency mixer I (102), frequency mixer Q (103), carrier wave NCO (104), correlator I (105), correlator Q (106), PN code generator (107), segmentation, it is characterized in that: differential correlator module (101) is for determining received digital intermediate frequency input signal PN code phase offset amount; The output signal of differential correlator module (101) is connected with frequency mixer I (102) and frequency mixer Q (103) simultaneously, and the sine produced with described carrier wave NCO (104) respectively, cosine signal are multiplied; Frequency mixer I (102) is connected with correlator I (105), and frequency mixer Q (103) is connected with correlator Q (106); PN code generator (107) is connected with correlator I (105) and correlator Q (106), for the PN code in the satellite-signal that relevant elimination receives simultaneously; Correlator I (105) is connected with correlator Q (106) zero padding FFT relevant to segmentation module (108), for determining the Doppler frequency deviation of receiving satellite signal; Acquisition Detection control module (109) zero padding FFT relevant to segmentation module (108) connects, and to FFT result modulo operation, and compares with the Acquisition Detection amount threshold value preset, and carries out catching result output.

2. GNSS New System signal capture device according to claim 1, is characterized in that: described digital intermediate frequency input signal is the digital medium-frequency signal of satellite-signal after down coversion mixing, intermediate-freuqncy signal filter and amplification, analog to digital conversion that satellite navigation receiver receives.

3. GNSS New System signal capture device according to claim 1, is characterized in that: described differential correlator module (101) comprises signal lag module (201), multiplier (202) and signal lag control module (203).

4. GNSS New System signal capture device according to claim 3, is characterized in that: described signal lag module (201) is connected with multiplier (202), for time delay satellite digital IF input signals; Multiplier (202) is connected with signal lag control module (203), and multiplier (202) is multiplied with the digital intermediate frequency input signal after time delay for the digital medium-frequency signal inputted; Signal lag control module (203) carries out square operation for the output signal of multiplier (202), and differentiates result, to determine the PN code phase offset amount of described digital medium-frequency signal.

5. GNSS New System signal capture device according to claim 1, it is characterized in that: described segmentation is correlated with zero padding FFT module (108) for plural for M point correlated results I+jQ is divided into R section, every segment length is M/R, and respectively noncoherent accumulation is carried out to each Piecewise Correlator, composition R point complex result, and carry out S point FFT computing after mending S-R individual 0, wherein plural correlated results I+jQ is digital medium-frequency signal respectively through formed the afterwards complex result of correlator I (105) and correlator Q (106), j is imaginary unit, M is the number of chips of tracking satellite, the span of R is 2 ~ 512, the span of S is 512 ~ 4096.

6. a GNSS New System signal acquisition methods, is characterized in that, described method comprises the steps:

Step S1: satellite navigation receiver receiving satellite signal, carries out down coversion mixing, intermediate frequency filtering amplifies and analog to digital conversion, obtain digital IF input signals;

Step S2: digital intermediate frequency input signal, through differential correlator module, obtains the PN code phase offset amount of tracking satellite;

Step S3: in-phase branch I and quadrature branch signal Q through comprising the frequency mixer of local carrier NCO, the correlator of local PN code generator, and is demodulated into complex number type baseband signal by digital intermediate frequency differential output signal;

Step S4: the complex number type baseband signal in step S3 to be correlated with zero padding FFT module through segmentation, and to Output rusults delivery;

Step S5: delivery is carried out to the result that FFT exports, and compares with the detection threshold value preset, determine Doppler frequency deviation and the PN code phase offset amount of institute's tracking satellite.

7. GNSS New System signal acquisition methods according to claim 6, it is characterized in that: in described step S2, differential correlator module comprises time delay module, multiplier and signal lag control module, for the sampled point number of order delayed digital IF input signals, be multiplied with original digital intermediate frequency input signal after digital medium-frequency signal after time delay gets conjugation, differentiate after modulo operation, obtain the PN code phase offset amount of institute's tracking satellite.

8. GNSS New System signal acquisition methods according to claim 6, it is characterized in that: in described step S3, Acquisition Detection control module (109) controls carrier wave NCO frequency mixer (104) and PN code generator (107), generates and mobile local PN code phase offset for controlling local carrier.

9. GNSS New System signal acquisition methods according to claim 6, it is characterized in that: in described step S4, segmentation is correlated with zero padding FFT module for plural for M point correlated results I+jQ is divided into R section, every segment length is M/R, and respectively noncoherent accumulation is carried out to each Piecewise Correlator, composition R point complex result, and carry out S point FFT computing after mending S-R individual 0, wherein plural correlated results I+jQ is digital medium-frequency signal respectively through formed the afterwards complex result of correlator I (105) and correlator Q (106), j is imaginary unit, M is the number of chips of tracking satellite, the span of R is 2 ~ 512, the span of S is 512 ~ 4096.