CN105790788A - Pseudocode-Doppler combined capturing method of direct sequence spread spectrum MSK signal - Google Patents

Abstract

The invention provides a pseudocode-Doppler combined capturing method of a direct sequence spread spectrum MSK signal. The method comprises the following steps of converting the direct sequence spread spectrum MSK signal to an appropriate direct sequence spread spectrum BPSK signal through intermediate-frequency matching processing; performing Doppler compensation on sampling data of the appropriate direct sequence spread spectrum BPSK signal; capturing the code phase error, the Doppler channel and Doppler displacement value of the BPSK signal after Doppler compensation through a pseudocode parallel capturing method, and acquiring a Doppler frequency offset according to the Doppler channel and the pseudocode displacement value. The pseudocode-Doppler combined capturing method can realize capturing of the direct sequence spread spectrum MSK signal in a high dynamic condition and furthermore finishes combined estimation for the pseudocode phase and the Doppler frequency offset. On the condition of relatively low signal-to-noise ratio, not only is relatively low Doppler frequency offset estimation error realized, but also high pseudocode phase capturing performance is obtained, thereby ensuring relatively good coarse synchronization of a received signal before entering a signal tracking processing module.

Description

The pseudo-code of a kind of DS msk signal-Doppler's joint acquisition method

Technical field

The present invention relates to a kind of digital communication technology, the pseudo-code of a kind of DS msk signal-Doppler's joint acquisition Method.

Background technology

At present, spread spectrum system many employings BPSK, QPSK modulation system, but both modulation systems cannot be applicable to exist sternly Weight non-linear distortion, Doppler frequency shift and the application of multipath fading.DS MSK communication system has spread spectrum system simultaneously Advantage and the constant-envelope of msk signal, the availability of frequency spectrums such as low intercepting and capturing, the random address ability of multi-user, strong anti-interference performance High, energy is concentrated, side lobe attenuation is fast, out-of-band radiation power is low, to advantages such as non-linear distortion are insensitive, tactical data link, The fields such as civil aviation Ground-to-Air Data Link, missile guidance instruction transmission, satellite communication are widely applied.So, DS MSK The field that signal cannot be suitable at DS-BPSK signal/QPSK signal still has good application prospect.

DS msk signal carries out the synchronization that the condition of accurate despread-and-demodulation is signal, including pseudo-code and carrier synchronization.Signal Synchronization is divided into thick synchronization and essence synchronization, and the present invention lays particular emphasis on signal and slightly synchronizes, i.e. pseudo-code and Doppler frequency deviation capture.Classical Msk signal catching method has two kinds, and a kind of is based on the relevant Code acquisition that slides, and this method realizes simple but capture time very Long；Another kind is Code acquisition based on matched filtering.The capture relevant peaks of both approaches is easily affected by Doppler frequency deviation, Be simply not proposed to high dynamically under capture.Scholar is had to propose to utilize the method that difference is relevant how general first to estimate on this basis Strangle frequency deviation and compensate, then doing Code acquisition with matched filtering, but the resultant error that the method estimates under low signal-to-noise ratio is relatively Greatly, more difficult acquirement balance and between Doppler's estimation range and estimation difference.So, DS MSK under high dynamically low signal-to-noise ratio The signal of communication system synchronizes to become technological difficulties.

First G.J.R.Povey et al. proposes to combine with FFT the catching of (PMF-FFT) based on numerical portion matched filtering device Obtain model, although this method alleviates the Doppler frequency deviation impact on acquiring pseudo code performance to a certain extent, and realize The two dimension capture of pseudo-code phase and carrier wave frequency deviation, but this method is mainly adapted to mpsk signal, and Doppler frequency deviation Capture range less, the most inapplicable under high dynamic environment.Thus, how to complete accurately and rapidly under high dynamic environment The pseudo-code phase of spread-spectrum signal and the capture of Doppler frequency deviation two dimension, become the key technology of DS MSK all-digital receiver.

Summary of the invention

It is an object of the invention to provide pseudo-code-Doppler's joint acquisition method of a kind of DS msk signal, it is achieved high dynamic Under state, the capture of DS msk signal, completes the Combined estimator of pseudo-code phase and Doppler frequency deviation, under relatively low signal-to-noise ratio, no simultaneously Only there is less Doppler frequency deviation estimation difference performance, and there is good pseudo-code phase acquisition performance, thus ensure Receive signal and the most obtain the thickest synchronization before entering signal follows the tracks of (essence synchronizes) processing module.This method includes following Step:

Step 1, is converted to approximate DS-BPSK signal signal by intermediate frequency matching treatment to DS msk signal；

Step 2, the sampled data of pairing approximation DS-BPSK signal signal carries out Doppler effect correction；

Step 3, to the bpsk signal after Doppler effect correction by pseudo-code parallel capturing method catch code phase error, how general Strangle passage and Doppler displacement value, and obtain Doppler frequency deviation according to Doppler's passage and Doppler displacement value.

Use said method, the production method of described DS-BPSK signal signal particularly as follows:

Step 1.1, uses wave filter h (t) to receive DS msk signal, and the impulse response of wave filter h (t) is

$h\left(t\right)=g(T-t)=\left\{\begin{array}{cc}-sin2{\mathrm{\πf}}_{2}t& 0\≤t\≤T\\ 0& others\end{array}\right.;$

Step 1.2, the signal after reception is successively through f₁Down coversion, low-pass filtering, K times extract after obtain DS-BPSK signal base band Signal

Wherein, wave filter h (t) and switched filter g (t) producing DS msk signal match,

$g\left(t\right)=\left\{\begin{array}{cc}sin2{\mathrm{\πf}}_{2}t& 0\≤t\≤T\\ 0& others\end{array}\right.,$

f_cFor carrier frequency, T is the spread spectrum code cycle,

τ is code phase error, and τ initial value is 0,For initial phase, R_cFor spread-spectrum code rate, f_dFor Doppler frequency deviation, n_i For the multiple white noise of Gauss, d_iFor transmission symbol, c_iFor spread spectrum code.

Using said method, pairing approximation DS-BPSK signal signal sample data carries out the detailed process of Doppler effect correction and is:

Step 2.1, to the pseudo-code sequence that each cycle is N in each data symbol in M the data symbol received Carry out extraction with N for interval to obtain one N × M and tie up matrix A, wherein

Sampling point sequence is

$c_0,c_1{e}^{j2\mathrm{\π}2f_d\frac{1}{R_c}},...,c_{N-1}{e}^{j2\mathrm{\π}2f_d\frac{N-1}{R_c}},c_0{e}^{j2\mathrm{\π}2f_d\frac{N}{R_c}},...,c_{N-1}{e}^{j2\mathrm{\π}2f_d\frac{2N-1}{R_c}},...,c_0{e}^{j2\mathrm{\π}2f_d\frac{(M-1)N}{R_c}},...c_{N-1}{e}^{j2\mathrm{\π}2f_d\frac{MN-1}{R_c}},$

Matrix A is

f_dFor Doppler frequency deviation, R_cFor spread-spectrum code rate, c_iFor spread spectrum code；

Step 2.2, is M point FFT to the often row of matrix A, obtains N × M and tie up matrix, and ignore Doppler's estimated value errorObtain matrix F

$F(i,k)==c_i{e}^{j2\mathrm{\π}2f_d\frac{i}{R_c}}\underset{n=0}{\overset{M-1}{\Σ}}{e}^{j2\mathrm{\π}\mathrm{\Δ}f\frac{nN}{R_c}},(i=0,1,...,N-1;k=0,1,...,M-1)$

Step 2.3, arranges phase compensating factor D of N × M dimension, and the acquisition Doppler that matrix F is multiplied with compensating factor D mends Matrix B after repaying, wherein

$D(i,k)=e^{-j\frac{2\mathrm{\π}}{MN}i\·k},(i=0,1,...,N-1;k=0,1,...,M-1)$

$B(i,k)=c_i{e}^{j2\mathrm{\π}\frac{m}{N}i+j2\mathrm{\π}\mathrm{\Δ}f\frac{i}{R_c}}\underset{n=0}{\overset{M-1}{\Σ}}{e}^{j2\mathrm{\π}\mathrm{\Δ}f\frac{nN}{R_c}},(i=0,1,...,N-1;k=0,1,...,M-1).$

Using said method, described pseudo-code parallel capturing method detailed process is:

Step 3.1, does N point FFT by the signal after Doppler effect correction；

Step 3.2, does shifting processing after this locality pseudo-code N point FFT, and shift range is i.e.Value is [-n_l,n_r], take Value is that negative indication is to leftValue translates to the right for just representing

Step 3.3, is multiplied step 3.1 with the result of step 3.2 and does N point IFFT conversion again, and capture obtains

$\begin{array}{c}y(\mathrm{\τ},k,\hat{m})=IFFT\left\{FFT\right\{B(i.k)\}\·FFT\{c_i{e}^{j\frac{2\mathrm{\π}\hat{m}{R}_c}{N}\·\frac{i}{R_c}}\left\}\right\}\\ =\underset{i=0}{\overset{N-1}{\Σ}}{c}_{i-\mathrm{\τ}}{e}^{j\frac{2\mathrm{\π}mi}{N}+j2\mathrm{\π}\mathrm{\Δ}f\frac{i}{R_c}}\·c_i{e}^{-j\frac{2\mathrm{\π}\hat{m}}{N}\·i}\underset{n=0}{\overset{M-1}{\Σ}}{e}^{j2\mathrm{\π}\mathrm{\Δ}f\frac{nN}{R_c}}\end{array},$

Wherein, τ is code phase error, and k is Doppler's passage, and m is pseudo-code shift value；

Step 3.4, obtains Doppler frequency deviation estimated value according to Doppler's passage and Doppler displacement valueWhereinIt is respectively Doppler's passage and the estimated value of Doppler displacement value.

Use said method, a decision threshold is set, if the peak value of envelope detected is more than decision gate in pseudo-code parallel capture Limit, then pseudo-code phase acquisition success；Otherwise, the first paragraph data of the sampled data of bpsk signal being abandoned, remaining segment data depends on Secondary move forward, and continue to the data in a spread spectrum code cycle, new data are re-started joint acquisition and processes, until peak Value is more than till decision threshold.

The present invention compared with prior art, has the advantage that (1) replaces convolution computing with FFT, substantially reduces The complexity of algorithm；(2) scanning for pseudo-code phase and carrier wave frequency deviation, capture time is M spread spectrum code cycle, capture simultaneously Speed is improved；(3) by loopy moving this locality pseudo-code FFT sequence, large doppler frequency deviation is compensated, can facilitate Ground expands doppler range, it is adaptable to high dynamic environment；(4) the relevant accumulation interval in the present invention is M pseudo-code In the cycle, introduce the snr gain of M times, be particularly suitable for the applied environment of low signal-to-noise ratio；(5) in DS msk signal being carried out Frequently matching treatment, is converted into DS-BPSK signal signal, solves the problem that msk signal form is complicated, can will be applicable to DS The quick capturing method of bpsk signal is applied to DS msk signal.

Below in conjunction with Figure of description, the invention will be further described.

Accompanying drawing explanation

Fig. 1 is the method flow diagram of the present invention.

Fig. 2 is the system composition diagram of the embodiment of the present invention.

Fig. 3 is the 3-D graphic of the trapping module correlation output of the embodiment of the present invention.

Fig. 4 is the detection probability of the embodiment of the present invention and the false-alarm probability change curve with input signal-to-noise ratio.

Detailed description of the invention

In conjunction with Fig. 1, the invention mainly includes steps:

Step 1, is converted to approximate DS-BPSK signal signal by intermediate frequency matching treatment to DS msk signal；

Step 2, the sampled data of pairing approximation DS-BPSK signal signal carries out Doppler effect correction；

Step 3, to the bpsk signal after Doppler effect correction based on FFT by pseudo-code parallel capturing method catch code phase place by mistake Difference, Doppler's passage and Doppler displacement value.

Specifically:

DS msk signal can use serial mode to produce, i.e. signal after spread spectrum and carrier wave cos (2 π f₁T) BPSK is carried out Modulation obtains DS-BPSK signal signal, more converted wave filter g (t) can produce DS msk signal.The impulse of switched filter rings Should be

$g\left(t\right)=\left\{\begin{array}{cc}sin2{\mathrm{\πf}}_{2}t& 0\≤t\≤T\\ 0& others\end{array}\right.---\left(1\right)$

Wherein, f_cFor carrier frequency, T is the spread spectrum code cycle.

For this serial producing method of DS msk signal, use at receiving terminal and match with switched filter g (t) Wave filter h (t) be received, the impulse response of corresponding matched filtering device is

$h\left(t\right)=g(T-t)=\left\{\begin{array}{cc}-sin2{\mathrm{\πf}}_{2}t& 0\≤t\≤T\\ 0& others\end{array}\right.---\left(2\right)$

Signal after coupling is again through f₁Down coversion, low-pass filtering, then carry out K times and extract, obtain DS-BPSK signal baseband signal. This extracting multiple is consistent with over-sampling multiple, and its output signal of now matched filtering is represented by

Wherein, τ is code phase error；For initial phase；R_cFor spread-spectrum code rate；f_dFor Doppler frequency deviation；n_iFor Gauss Multiple white noise；Analyze for convenience, set initial phaseCode phase error τ=0.Due to posttectonic approximation DS There is data symbol saltus step in bpsk signal, destroys its autocorrelation performance, thus catch pseudo-code phase and Doppler frequency deviation Obtain generation impact.In order to eliminate the impact of symbol saltus step, can be by posttectonic signal and one signal obtained of self delay It is multiplied.At this moment, the Doppler frequency deviation of signal becomes original 2 times.And negligible transmission symbol d during lower surface analysis_iImpact, only Analyze spread spectrum code c_iEffect.

Now Doppler effect correction module receives M data symbol, and comprising a cycle inside each data symbol is the puppet of N Code sequence, sampling point is represented by

$c_0,c_1{e}^{j2\mathrm{\π}2f_d\frac{1}{R_c}},...,c_{N-1}{e}^{j2\mathrm{\π}2f_d\frac{N-1}{R_c}},c_0{e}^{j2\mathrm{\π}2f_d\frac{N}{R_c}},...,c_{N-1}{e}^{j2\mathrm{\π}2f_d\frac{2N-1}{R_c}},...,c_0{e}^{j2\mathrm{\π}2f_d\frac{(M-1)N}{R_c}},...c_{N-1}{e}^{j2\mathrm{\π}2f_d\frac{MN-1}{R_c}}---\left(4\right)$

This sequence carried out extraction with N for interval obtain N × M dimension matrix A be

$A(i,n)=c_i{e}^{j2\mathrm{\π}2f_d\frac{nN+i}{R_c}},(i=0,1,...,N-1;n=0,1,...,M-1)---\left(5\right)$

The each column of A is a spread spectrum code cycle, is represented by more intuitively

To A often row be M point FFT, obtain N × M dimension matrix F:

$\begin{array}{c}F(i,k)=\underset{n=0}{\overset{M-1}{\Σ}}{c}_i{e}^{j2\mathrm{\π}2f_d\frac{nN+i}{R_c}}{e}^{-j\frac{2\mathrm{\π}}{M}nk}=c_i{e}^{j2\mathrm{\π}2f_d\frac{i}{R_c}}\underset{n=0}{\overset{M-1}{\Σ}}{e}^{j2\mathrm{\π}2f_d\frac{nN}{R_c}-j\frac{2\mathrm{\π}}{M}nk}---\left(7\right)\\ (i=0,1,...,N-1;k=0,1,...,M-1)\end{array}$

The kth of F is arranged and regards asDoppler's passage, to arbitrary f_d, all there is k and m so that:

$f_d=\frac{{\mathrm{kR}}_c}{2MN}+\frac{{\mathrm{mR}}_c}{2N}+\mathrm{\Δ}f---\left(8\right)$

Wherein,Analyze for convenience, below ignore this impact.Thus can determine that DOPPLER RESOLUTIONWithout blurred bandwidth it isThis up-to-date style (7) is rewritable is

$\begin{array}{c}F(i,k)==c_i{e}^{j2\mathrm{\π}2f_d\frac{i}{R_c}}\underset{n=0}{\overset{M-1}{\Σ}}{e}^{j2\mathrm{\π}\mathrm{\Δ}f\frac{nN}{R_c}}\\ (i=0,1,...,N-1;k=0,1,...,M-1)\end{array}---\left(9\right)$

If Doppler frequency f of signal_dIt is less than without blurred bandwidth, i.e. during m=0, can be according to the passage at FFT peak value place Number k estimates f_d.But still there is Doppler frequency deviation on kth passage, can be solved by phase compensation, introduce N × M dimension here Phase compensating factor D:

$\begin{array}{c}D(i,k)=e^{-j\frac{2\mathrm{\π}}{MN}i\·k}\\ (i=0,1,...,N-1;k=0,1,...,M-1)\end{array}---\left(10\right)$

Being multiplied corresponding for F with D, obtaining the result after Doppler effect correction is

$\begin{array}{c}B(i,k)=c_i{e}^{j2\mathrm{\π}\frac{m}{N}i+j2\mathrm{\π}\mathrm{\Δ}f\frac{i}{R_c}}\underset{n=0}{\overset{M-1}{\Σ}}{e}^{j2\mathrm{\π}\mathrm{\Δ}f\frac{nN}{R_c}}\\ (i=0,1,...,N-1;k=0,1,...,M-1)\end{array}---\left(11\right)$

But, work as f_dWhen exceeding without blurred bandwidth, i.e. during m ≠ 0, remain a fixed frequency difference itemThis fixed frequency difference Item can be estimated by pseudo-code parallel capture below.The concrete grammar used is backward for this locality pseudo-code is done N point FFT LeftPosition:

$C(k-\hat{m})=\underset{i=0}{\overset{N-1}{\Σ}}{c}_i{e}^{-j\frac{2\mathrm{\π}}{N}i(k-\hat{m})}=\underset{i=0}{\overset{N-1}{\Σ}}{c}_i{e}^{j2\mathrm{\π}\frac{\hat{m}}{N}i}{e}^{-j\frac{2\mathrm{\π}}{N}nk}=FFT\left\{c_i{e}^{j2\mathrm{\π}\frac{\hat{m}}{N}i}\right\}---\left(12\right)$

Be equivalent to the sequence of time domainDo N point FFT, and this displacement item just can estimate fixing frequency DifferenceRemaining frequency difference itemIt is the frequency deviation estimation difference of the present invention, can be by suitably adjusting the M that counts of FFT Reduce its value so that this frequency deviation estimation difference does not affect pseudo-code phase capture below.

The detailed process of pseudo-code parallel capturing method is: the signal after Doppler effect correction does N point FFT, will simultaneously Do shifting processing after local pseudo-code N point FFT, the result after two resume module is multiplied and does N point IFFT conversion again, here Shift range is i.e.Value [-n_l,n_r], value is that negative indication is to leftValue translates to the right for just representingDuring analysis Whole process can be regarded as B (i, k) andConvolution effect.Now, by start setting up be 0 code phase error τ weight New introduction, the result of pseudo-code parallel capture is

$\begin{array}{c}y(\mathrm{\τ},k,\hat{m})=IFFT\left\{FFT\right\{B(i.k)\}\·FFT\{c_i{e}^{j\frac{2\mathrm{\π}\hat{m}{R}_c}{N}\·\frac{i}{R_c}}\left\}\right\}\\ =\underset{i=0}{\overset{N-1}{\Σ}}{c}_{i-\mathrm{\τ}}{e}^{j\frac{2\mathrm{\π}mi}{N}+j2\mathrm{\π}\mathrm{\Δ}f\frac{i}{R_c}}\·c_i{e}^{-j\frac{2\mathrm{\π}\hat{m}}{N}\·i}\underset{n=0}{\overset{M-1}{\Σ}}{e}^{j2\mathrm{\π}\mathrm{\Δ}f\frac{nN}{R_c}}\end{array}---\left(13\right)$

WhenTime, have

$\begin{array}{c}y(\mathrm{\τ},k,m)=\underset{i=0}{\overset{N-1}{\Σ}}{c}_{i-\mathrm{\τ}}{c}_i{e}^{j2\mathrm{\π}\mathrm{\Δ}f\frac{i}{R_c}}\underset{n=0}{\overset{M-1}{\Σ}}{e}^{j2\mathrm{\π}\mathrm{\Δ}f\frac{nN}{R_c}}\\ =\underset{i^{\′}=0}{\overset{MN-1}{\Σ}}{c}_{i^{\′}-\mathrm{\τ}}{c}_{i^{\′}}{e}^{j2\mathrm{\π}\mathrm{\Δ}f\frac{i^{\′}}{R_c}}\end{array}---\left(14\right)$

Wherein, i'=i+nN, and pseudo-code sequence { c_iWith N as cycle, so c_i'=c_i+nN=c_i。

The judging module of the present invention uses the method that envelope detected and CFAR thresholding are arranged.The correlated results of parallel capture Output is to judging module, if correlation peak is more than decision threshold, then it represents that pseudo-code phase acquisition success；If correlation peak does not has Having more than decision threshold, then it represents that do not have acquisition success, the first paragraph data in M segment data abandoned, remaining M-1 segment data depends on Secondary move forward, after will continue to the data in a spread spectrum code cycle, obtain the sequence that new M segment length is N point continue into Row τ-f_dTwo dimension joint acquisition processes, until correlation peak is more than decision threshold, completes pseudo-code phase capture.

Capture and obtain three estimated values about relevant peaks:Wherein,It is pseudorandom codes phase estimation value, and how general The estimated value strangled is

$\widehat{f_d}=\frac{(\hat{k}-1)R_c}{2MN}+\frac{(\hat{m}-1)R_c}{2N}---\left(15\right)$

Doppler's estimated value error of the present invention is

$\mathrm{\Δ}f\≤\frac{R_c}{2MN}---\left(16\right)$

Doppler's estimation range is

$\[-\frac{n_L{R}_c}{2N},\frac{n_R{R}_c}{2N}+\frac{(M-1)R_c}{2MN}\]---\left(17\right)$

Embodiment one

System sampling frequency f_s=245.52MHz, over-sampling multiple K=12, IF-FRE f_c=76.725MHz, spreads code Speed is R_c=20.46Mchip/s, data rate is 20kbps, and spread spectrum code uses Gold sequence, and code length N=1023, two dimension is searched The code periodicity of rope, i.e. receives data symbol number M=32, n_L=50, n_R=50.

The intermediate frequency DS msk signal received carries out coupling through the matched filtering device that impulse response is h (t) and receives.Coupling The impulse response of wave filter is

$h\left(t\right)=\left\{\begin{array}{cc}-sin2{\mathrm{\πf}}_{2}t& 0\≤t\≤T\\ 0& others\end{array}\right.---\left(18\right)$

Wherein, For the spread spectrum code cycle.The frequency response of matched filtering device is

$D\left(f\right)=-\frac{{\mathrm{sin\πT}}_c(f-f_2)}{{\mathrm{\πT}}_c(f-f_2)}---\left(19\right)$

Wave filter of the present invention be designed with convex optimisation technique, it is necessary first to filter design problem is converted into convex optimization Problem, can be established as convex Optimized model by the peakedness coefficient of matched filtering device:

$\min imize\underset{\mathrm{\ω}\∈\[0,\mathrm{\π}\]}{sup}|H\left(\mathrm{\ω}\right)-D\left(\mathrm{\ω}\right)|---\left(20\right)$

Wherein, sup is infimum；ω=2 π f is angular frequency；D (ω) is given frequency response function；H (ω) is institute The frequency response of the wave filter of design,H (n) is filter coefficient, N₀For filter order.

Filter coefficient h (n) of actual application is tried to achieve by the cvx workbox of Matlab software.

Signal after coupling is again through f₁Down coversion, whereinLow-pass filtering after down coversion, then K times extracted, this Extracting multiple is consistent with over-sampling multiple.Now obtaining match output signal is

Wherein, τ is code phase error, analyzes for convenience and is set to 0；Also 0 it is set to for initial phase；f_dFor Doppler's frequency Partially；n_iFor the multiple white noise of Gauss；In order to eliminate the impact of symbol saltus step, posttectonic signal can be obtained with self delay one The signal multiplication arrived.At this moment, the Doppler frequency deviation of signal becomes original 2 times.So ignoring transmission symbol d during lower surface analysis_i Impact, only analyze spread spectrum code c_iEffect.

Doppler effect correction module receives M data symbol, comprises the pseudo-code sequence that cycle is N inside each data symbol Row.Extracting with N this sequence for interval, often M data of extraction are as a line, obtain N × M dimension matrix A:

$A(i,n)=c_i{e}^{j2\mathrm{\π}2f_d\frac{nN+i}{R_c}},(i=0,1,...,N-1;n=0,1,...,M-1)---\left(22\right)$

The often row of A is M point FFT, obtains the matrix F of N × M:

$\begin{array}{c}F(i,k)=\underset{n=0}{\overset{M-1}{\Σ}}{c}_i{e}^{j2\mathrm{\π}2f_d\frac{nN+i}{R_c}}{e}^{-j\frac{2\mathrm{\π}}{M}nk}=c_i{e}^{j2\mathrm{\π}2f_d\frac{i}{R_c}}\underset{n=0}{\overset{M-1}{\Σ}}{e}^{j2\mathrm{\π}2f_d\frac{nN}{R_c}-j\frac{2\mathrm{\π}}{M}nk}\\ (i=0,1,...,N-1;k=0,1,...,M-1)\end{array}---\left(23\right)$

To arbitrary f_d, all k and m of existence anduniquess so that:

$f_d=\frac{{\mathrm{kR}}_c}{2MN}+\frac{{\mathrm{mR}}_c}{2N}+\mathrm{\Δ}f---\left(24\right)$

Wherein,The impact of Δ f is ignored during lower surface analysis.The kth of F is arranged and regards asDoppler Passage, then without blurred bandwidth be

Below with regard to f_dExceed and discuss in two kinds of situation without blurred bandwidth with being less than.

(1) it is less than without blurred bandwidth, i.e. m=0.Now f_dFall on kth passage, and the estimated value of k can be according to FFT Peak value determines.But still there is Doppler frequency deviation on kth passage, can be solved by phase compensation, introduce a phase compensation here Factor D:

$D(i,k)=e^{-j\frac{2\mathrm{\π}}{MN}i\·k},(i=0,1,...,N-1;k=0,1,...,M-1)---\left(25\right)$

It is multiplied corresponding for F with D, obtains the result after Doppler effect correction:

$B(i,k)=c_i{e}^{j2\mathrm{\π}\frac{m}{N}i+j2\mathrm{\π}\mathrm{\Δ}f\frac{i}{R_c}}\underset{n=0}{\overset{M-1}{\Σ}}{e}^{j2\mathrm{\π}\mathrm{\Δ}f\frac{nN}{R_c}},(i=0,1,...,N-1;k=0,1,...,M-1)---\left(26\right)$

(2) exceed without blurred bandwidth, i.e. m ≠ 0.Still a fixed frequency difference is remained after carrying out Doppler effect correction in (1) ?The method that the present invention uses is to estimate with FFT pseudo-code parallel capture.Concrete grammar for carrying out N point by this locality pseudo-code FFT, then to right translationPosition, the operation of this two step is equivalent to the sequence of time domainDo N point FFT, institute The m value in fixed frequency difference just can be estimated with this shift value.

After Doppler effect correction and pseudo-code parallel capture, remaining frequency difference itemThe frequency deviation being the present invention is estimated Meter error, can reduce its value by the M value of counting suitably adjusting FFT.

The detailed process of pseudo-code parallel capturing method is: the signal after Doppler effect correction does N point FFT, will simultaneously Do shifting processing after local pseudo-code N point FFT, the result after two resume module is multiplied and does N point IFFT conversion again, shift model Enclose for [-n_L,n_R], i.e.Value [-n_L,n_R].HereValue is that negative indication is to leftValue translates to the right for just representingWhole process can be regarded as during analysis B (i, k) andConvolution effect.Now, by start setting up be 0 code-phase Position error τ is introduced again, and the result of pseudo-code parallel capture is

$\begin{array}{c}y(\mathrm{\τ},k,\hat{m})=IFFT\left\{FFT\right\{B(i.k)\}\·FFT\{c_i{e}^{j\frac{2\mathrm{\π}\hat{m}{R}_c}{N}\·\frac{i}{R_c}}\left\}\right\}\\ =\underset{i=0}{\overset{N-1}{\Σ}}{c}_{i-\mathrm{\τ}}{e}^{j\frac{2\mathrm{\π}mi}{N}+j2\mathrm{\π}\mathrm{\Δ}f\frac{i}{R_c}}\·c_i{e}^{-j\frac{2\mathrm{\π}\hat{m}}{N}\·i}\underset{n=0}{\overset{M-1}{\Σ}}{e}^{j2\mathrm{\π}\mathrm{\Δ}f\frac{nN}{R_c}}\end{array}---\left(27\right)$

WhenTime:

$\begin{array}{c}y(\mathrm{\τ},k,m)=\underset{i=0}{\overset{N-1}{\Σ}}{c}_{i-\mathrm{\τ}}{c}_i{e}^{j2\mathrm{\π}\mathrm{\Δ}f\frac{i}{R_c}}\underset{n=0}{\overset{M-1}{\Σ}}{e}^{j2\mathrm{\π}\mathrm{\Δ}f\frac{nN}{R_c}}\\ =\underset{i^{\′}=0}{\overset{MN-1}{\Σ}}{c}_{i^{\′}-\mathrm{\τ}}{c}_{i^{\′}}{e}^{j2\mathrm{\π}\mathrm{\Δ}f\frac{i^{\′}}{R_c}}\end{array}---\left(28\right)$

Wherein, i'=i+nN, and pseudo-code sequence { c_iWith N as cycle, so c_i'=c_i+nN=c_i。

Judging module uses the method that envelope detected and CFAR thresholding are arranged.Capture obtains estimating about three of relevant peaks Evaluation:Wherein,It is pseudorandom codes phase estimation value, and the estimated value of Doppler is:

$\widehat{f_d}=\frac{(\hat{k}-1)R_c}{2MN}+\frac{(\hat{m}-1)R_c}{2N}---\left(29\right)$

Doppler's estimated value error of the present invention is

$\mathrm{\Δ}f\≤\frac{R_c}{2MN}---\left(30\right)$

Doppler's estimation range is

$\[-\frac{n_L{R}_c}{2N},\frac{n_R{R}_c}{2N}+\frac{(M-1)R_c}{2MN}\]---\left(31\right)$

Fig. 2 is for receiving data symbol number M=32, signal to noise ratio snr=-10dB, Doppler frequency deviation f_d=160.589kHz, During pseudo-code difference τ=474.7chip, carry out pseudo-code phase relevant defeated with the normalization that carrier doppler frequency deviation two-dimensional search obtains The 3-D graphic gone out.As can be seen from the figure the estimation of pseudo-code phase differs 0.3chip with actual value, in 0.5chip；Carrier wave The estimated value of frequency deviation is 160.625kHz, differs 36Hz with actual value, in the range of maximum residual frequency difference.

The emulation each estimated value of gained is:Calculating formula (29), (30), (31) can obtain:Doppler's estimated value error delta f≤312.5Hz；Doppler's estimation range is [-500000,509687.5] Hz.Pseudo-code-Doppler's joint acquisition method of visible present invention design can be to pseudo-code phase and carrier wave under high dynamic environment Frequency deviation captures accurately.

Fig. 3 is input signal-to-noise ratio SNR=[-30dB ,-5dB], f_d=40kHz, false-alarm probability P_fWhen=0.001, carry out Capture the detection probability obtained and the false-alarm probability change curve with signal to noise ratio for 5000 times.Reach when signal to noise ratio as seen from the figure During to-20dB, detection probability levels off to 1, and owing to using CFAR thresholding design standard, false-alarm probability is little affected by signal to noise ratio Impact.

It can thus be seen that pseudo-code-Doppler's joint acquisition method of the DS msk signal of present invention design, with existing Catching method compare, have algorithm complex little, simultaneously complete pseudo-code phase and Doppler frequency deviation capture, estimation difference little, Expand the advantages such as processing gain, be highly suitable for the applied environment of high dynamically low signal-to-noise ratio, there is stronger practical value.

Claims (5)

1. the pseudo-code of a DS msk signal-Doppler's joint acquisition method, it is characterised in that comprise the following steps:

Be converted to approximate DS-BPSK signal signal by intermediate frequency matching treatment to DS msk signal；

The sampled data of pairing approximation DS-BPSK signal signal carries out Doppler effect correction；

To the bpsk signal after Doppler effect correction by pseudo-code parallel capturing method catch code phase error, Doppler's passage and many General Le shift value, and obtain Doppler frequency deviation according to Doppler's passage and Doppler displacement value.

Method the most according to claim 1, it is characterised in that the production method of described DS-BPSK signal signal particularly as follows:

Using wave filter h (t) to receive DS msk signal, the impulse response of wave filter h (t) is

$h\left(t\right)=g(T-t)=\left\{\begin{array}{cc}-sin2{\mathrm{\πf}}_{2}t& 0\≤t\≤T\\ 0& others\end{array}\right.;$

Signal after reception is successively through f₁Down coversion, low-pass filtering, K times extract after obtain DS-BPSK signal baseband signal

Wherein, wave filter h (t) and switched filter g (t) producing DS msk signal match,

$g\left(t\right)=\left\{\begin{array}{cc}sin2{\mathrm{\πf}}_{2}t& 0\≤t\≤T\\ 0& others\end{array}\right.$

f_cFor carrier frequency, T is the spread spectrum code cycle,

τ is code phase error, and τ initial value is 0,For initial phase, R_cFor spread-spectrum code rate, f_dFor Doppler frequency deviation, n_iFor height This multiple white noise, d_iFor transmission symbol, c_iFor spread spectrum code.

Method the most according to claim 1, it is characterised in that pairing approximation DS-BPSK signal signal sample data carries out Doppler The detailed process compensated is:

Step 2.1, to the pseudo-code sequence that each cycle is N in each data symbol in M the data symbol received with N be Interval carries out extraction and obtains one N × M and tie up matrix A, wherein

Sampling point sequence is

$c_0,c_1{e}^{j2\mathrm{\π}2f_d\frac{1}{R_c}},...,c_{N-1}{e}^{j2\mathrm{\π}2f_d\frac{N-1}{R_c}},c_0{e}^{j2\mathrm{\π}2f_d\frac{N}{R_c}},...,c_{N-1}{e}^{j2\mathrm{\π}2f_d\frac{2N-1}{R_c}},...,c_0{e}^{j2\mathrm{\π}2f_d\frac{\left(M-1\right)N}{R_c}},...c_{N-1}{e}^{j2\mathrm{\π}2f_d\frac{MN-1}{R_c}},$

Matrix A is(i=0,1 ..., N-1；N=0,1 ..., M-1),

f_dFor Doppler frequency deviation, R_cFor spread-spectrum code rate, c_iFor spread spectrum code；

Step 2.2, is M point FFT to the often row of matrix A, obtains N × M and tie up matrix, and ignore Doppler's estimated value errorObtain matrix F

$F(i,k)==c_i{e}^{j2\mathrm{\π}2f_d\frac{i}{R_c}}\underset{n=0}{\overset{M-1}{\Σ}}{e}^{j2\mathrm{\π}\mathrm{\Δ}f\frac{nN}{R_c}},\left(i=0,1,...,N-1;k=0,1,...,M-1\right)$

Step 2.3, arranges phase compensating factor D of N × M dimension, after the acquisition Doppler effect correction that matrix F is multiplied with compensating factor D Matrix B, wherein

$D(i,k)=e^{-j\frac{2\mathrm{\π}}{MN}i\·k},(i=0,1,...,N-1;k=0,1,...,M-1)$

$B(i,k)=c_i{e}^{j2\mathrm{\π}\frac{m}{N}i+j2\mathrm{\π}\mathrm{\Δ}f\frac{i}{R_c}}\underset{n=0}{\overset{M-1}{\Σ}}{e}^{j2\mathrm{\π}\mathrm{\Δ}f\frac{nN}{R_c}},\left(i=0,1,...,N-1;k=0,1,...,M-1\right).$

Method the most according to claim 1, it is characterised in that described pseudo-code parallel capturing method detailed process is:

Step 3.1, does N point FFT by the signal after Doppler effect correction；

Step 3.2, does shifting processing after this locality pseudo-code N point FFT, and shift range is i.e.Value is [-n_l,n_r], value is Negative indication is to leftValue translates to the right for just representing

Step 3.3, is multiplied step 3.1 with the result of step 3.2 and does N point IFFT conversion again, and capture obtains

$\begin{array}{c}y(\mathrm{\τ},k,\hat{m})=\mathrm{IFFT}\left\{\mathrm{FFT}\right\{B(i.k)\}\·\mathrm{FFT}\{c_i{e}^{j\frac{2\mathrm{\π}\hat{m}{R}_c}{N}\·\frac{i}{R_c}}\left\}\right\}\\ =\underset{i=0}{\overset{N-1}{\mathrm{\Σ}}}{c}_{i-\mathrm{\τ}}{e}^{j\frac{2\mathrm{\πmi}}{N}+j2\mathrm{\π\Δf}\frac{i}{R_c}}\·c_i{e}^{-j\frac{2\mathrm{\π}\hat{m}}{N}\·i}\underset{n=0}{\overset{M-1}{\mathrm{\Σ}}}{e}^{j2\mathrm{\π\Δf}\frac{\mathrm{nN}}{R_c}}\end{array},$

Wherein, τ is code phase error, and k is Doppler's passage, and m is Doppler displacement value；

Step 3.4, obtains Doppler frequency deviation estimated value according to Doppler's passage and pseudo-code shift value WhereinIt is respectively Doppler's passage and the estimated value of pseudo-code shift value.

Method the most according to claim 1 a, it is characterised in that decision threshold is set, if envelope in pseudo-code parallel capture The peak value of detection is more than decision threshold, then pseudo-code phase acquisition success；Otherwise, by the first hop count of the sampled data of bpsk signal According to abandoning, remaining segment data moves forward successively, and continues to the data in a spread spectrum code cycle, re-starts new data Joint acquisition processes, till peak value is more than decision threshold.