CN103647461A

CN103647461A - Control method and apparatus of AC-DC series resonance matrix converter

Info

Publication number: CN103647461A
Application number: CN201310655501.9A
Authority: CN
Inventors: 张政权; 刘庆想; 李伟
Original assignee: Southwest Jiaotong University
Current assignee: Li Wei; Liu Qingxiang; Zhang Zhengquan; Southwest Jiaotong University
Priority date: 2013-12-06
Filing date: 2013-12-06
Publication date: 2014-03-19
Anticipated expiration: 2033-12-06
Also published as: CN103647461B

Abstract

The invention relates to matrix converter control technology and high-frequency AC link technology, in particular to a control method and device for an AC-DC series resonant matrix converter for high-voltage direct current loads. The method of the present invention proposes a control strategy in which the excitation voltage is first switched from low-line voltage to high-line voltage and then switched to zero voltage within half a cycle of high-frequency current, thereby realizing the instantaneous synthesis of three voltages. While the equivalent excitation voltage is adjusted, the average value of the input line current of each phase is proportional to the phase voltage, and only a small filter inductance value can achieve high power factor and low harmonic current. The beneficial effect of the invention is that it can realize the control of the AC-DC series resonant matrix converter with stable voltage output featuring high efficiency, high power factor, low harmonic wave and low peak current. The invention is especially applicable to AC-DC series resonant matrix converters.

Description

A control method and device for an AC-DC series resonant matrix converter

技术领域technical field

本发明涉及矩阵变换器控制技术和高频交流连接技术，具体的说是涉及一种高压直流负载用AC-DC串联谐振矩阵变换器的控制方法及装置。The invention relates to matrix converter control technology and high-frequency AC connection technology, in particular to a control method and device for an AC-DC series resonant matrix converter for high-voltage direct current loads.

背景技术Background technique

高压直流电源在连续波以及长脉冲调制器高功率微波系统中有着广泛应用。为了满足未来高技术战争的军事需求，高功率微波系统在朝着高功率、小型化、轻量化的方向发展，这就要求其电源有更高的功率密度，效率和功率因数。目前普遍使用的电源一般采用存在中间直流储能环节的DC-Link技术，中间储能环节的存在必然会增加电源系统的体积和重量，降低了电源的功率密度；另外，这种电源在其电网输入端的电能质量不高，功率因数较低、谐波含量较大，为了进行校正或抑制，必然需要引入额外的电力电子器件，这样又进一步降低了供电系统的功率密度和效率，为了解决上述问题，研究新拓扑结构与控制技术的电源，提高电源的效率，功率密度和功率因数就变得尤为重要。High voltage DC power supplies are widely used in continuous wave and long pulse modulator high power microwave systems. In order to meet the military needs of future high-tech warfare, high-power microwave systems are developing in the direction of high power, miniaturization, and light weight, which requires their power supplies to have higher power density, efficiency, and power factor. At present, the commonly used power supply generally adopts DC-Link technology with an intermediate DC energy storage link. The existence of the intermediate energy storage link will inevitably increase the volume and weight of the power supply system and reduce the power density of the power supply; in addition, this kind of power supply in its power grid The power quality at the input end is not high, the power factor is low, and the harmonic content is large. In order to correct or suppress, it is necessary to introduce additional power electronic devices, which further reduces the power density and efficiency of the power supply system. In order to solve the above problems , it becomes especially important to study the power supply with new topology and control technology, and to improve the efficiency, power density and power factor of the power supply.

矩阵变换器具有能量双向流通、正弦输入输出电流、输入功率因数可控、输出电压幅值和相位可控、无中间储能环节和结构紧凑等诸多优点，将矩阵变换器应用于高压直流电源将显著提高电源的功率密度。The matrix converter has many advantages such as bidirectional flow of energy, sinusoidal input and output current, controllable input power factor, controllable output voltage amplitude and phase, no intermediate energy storage link, and compact structure. Applying the matrix converter to a high-voltage DC power supply will Significantly increases the power density of the power supply.

目前矩阵变换器的调制算法主要分为AV调制算法，瞬时电压合成算法和空间矢量调制算法。这些调制算法相对复杂，计算量较大，更重要的是不能适用在高频(几十kHz)输出场合。当前矩阵变换器的换流策略主要分为电压型和电流型换流策略，为实现可靠的换流输入需串联较大电感防止输入短路，输出需采用箝位电路防止输出开路，而这些方法也不适用于当前高频工作的拓扑电路。At present, the modulation algorithms of matrix converters are mainly divided into AV modulation algorithms, instantaneous voltage synthesis algorithms and space vector modulation algorithms. These modulation algorithms are relatively complex, with a large amount of calculation, and more importantly, they cannot be applied to high-frequency (tens of kHz) output occasions. The current commutation strategies of matrix converters are mainly divided into voltage-type and current-type commutation strategies. In order to achieve reliable commutation input, a large inductance must be connected in series to prevent input short-circuit, and a clamp circuit must be used to prevent output from being open. These methods also Not suitable for current topological circuits operating at high frequencies.

应用在高频输出场合的矩阵变换器称为高频交流链，随着矩阵开关后接的主回路类型以及工作模式的不同，矩阵开关的控制策略、换流策略均不同，不能借鉴已有的方法；为了减小电源的输出纹波、提高输入侧功率因数、同时减小主回路电流峰值，需研究应用适用于当前拓扑电路和工作模式的新型控制策略和换流策略。The matrix converter used in high-frequency output occasions is called a high-frequency AC link. With the difference in the type of main circuit connected behind the matrix switch and the working mode, the control strategy and commutation strategy of the matrix switch are different, and the existing ones cannot be used for reference. Methods: In order to reduce the output ripple of the power supply, improve the power factor of the input side, and reduce the peak current of the main circuit at the same time, it is necessary to study and apply a new control strategy and commutation strategy suitable for the current topology circuit and working mode.

发明内容Contents of the invention

本发明所要解决的，就是针对上述常规矩阵变换器的不足，提出一种实现稳压输出的AC-DC串联谐振矩阵变换器的控制方法及装置。What the present invention aims to solve is to propose a control method and device for an AC-DC series resonant matrix converter that realizes voltage-stabilized output in view of the shortcomings of the above-mentioned conventional matrix converter.

本发明解决上述技术问题所采用的技术方案是：一种AC-DC串联谐振矩阵变换器的控制方法，其特征在于，包括以下步骤：The technical solution adopted by the present invention to solve the above-mentioned technical problems is: a kind of control method of AC-DC series resonant matrix converter, it is characterized in that, comprises the following steps:

a.实时采集负载电压V₀和三相电压源的三相输入相电压u_a，u_b，u_c；a. Collect the load voltage V ₀ and the three-phase input phase voltage u _a , u _b , u _c of the three-phase voltage source in real time;

b.根据实时采集到的三相输入相电压u_a，u_b，u_c的相对大小关系，将每个输入相电压周期划分为12个区间，每个区间内相电压的极性和大小确定，且保持单调变化，所述12个区间具体为：b. According to the relative size relationship of the three-phase input phase voltage u _a , u _b , u _c collected in real time, each input phase voltage cycle is divided into 12 intervals, and the polarity and magnitude of the phase voltage in each interval are determined , and maintain a monotonous change, the 12 intervals are specifically:

区间Ⅰ：u_a＞u_c＞u_b，U_P=u_a，U_M=u_c，U_N=u_b；Interval Ⅰ: u _a > u _c > u _b , U _P = u _a , U _M = u _c , U _N = u _b ;

区间Ⅱ：u_a＞u_b＞u_c，U_P=u_a，U_M=u_b，U_N=u_c；Interval II: u _a > u _b > u _c , U _P = u _a , U _M = u _b , U _N = u _c ;

区间Ⅲ：u_a＞u_b＞u_c，U_P=u_c，U_M=u_b，U_N=u_a；Interval III: u _a > u _b > u _c , U _P = u _c , U _M = u _b , U _N = u _a ;

区间Ⅳ：u_b＞u_a＞u_c，U_P=u_c，U_M=u_a，U_N=u_b；Interval IV: u _b >u _a >u _c , U _P =u _c , U _M =u _a , U _N =u _b ;

区间Ⅴ：u_b＞u_a＞u_c，U_P=u_b，U_M=u_a，U_N=u_c；Interval Ⅴ: u _b > u _a > u _c , U _P = u _b , U _M = u _a , U _N = u _c ;

区间Ⅵ：u_b＞u_c＞u_a，U_P=u_b，U_M=u_c，U_N=u_a；Interval VI: u _b >u _c >u _a , U _P =u _b , U _M =u _c , U _N =u _a ;

区间Ⅶ：u_b＞u_c＞u_a，U_P=u_a，U_M=u_b，U_N=u_c；Interval VII: u _b > u _c > u _a , U _P = u _a , U _M = u _b , U _N = u _c ;

区间Ⅷ：u_c＞u_b＞u_a，U_P=u_a，U_M=u_b，U_N=u_c；Interval Ⅷ: u _c > u _b > u _a , U _P = u _a , U _M = u _b , U _N = u _c ;

区间Ⅸ：u_c＞u_b＞u_a，U_P=u_c，U_M=u_b，U_N=u_a；Interval IX: u _c > u _b > u _a , U _P = u _c , U _M = u _b , U _N = u _a ;

区间Ⅹ：u_c＞u_a＞u_b，U_P=u_c，U_M=u_a，U_N=u_b；Interval X: u _c > u _a > u _b , U _P = u _c , U _M = u _a , U _N = u _b ;

区间Ⅺ：u_c＞u_a＞u_b，U_P=u_b，U_M=u_a，U_N=u_c；Interval Ⅺ: u _c > u _a > u _b , U _P = u _b , U _M = u _a , U _N = u _c ;

区间Ⅻ：u_a＞u_c＞u_b，U_P=u_b，U_M=u_c，U_N=u_a；Interval Ⅻ: u _a > u _c > u _b , U _P = u _b , U _M = u _c , U _N = u _a ;

c.采用低线电压U_k、高线电压U_j以及0电压共同参与的组合方式完成激励，即采用6过程的工作模式，谐振电流正半周和负半周均进行2次换流且均包含3个工作过程，正负半周激励电压的极性相反，具体为：第1个工作过程采用低线电压U_k，第2个工作过程采用高线电压U_j，第3个工作过程采用0电压，第4个工作过程采用低线电压-U_k，第5个工作过程采用高线电压-U_j，第6个工作过程采用0电压；假设在1-2工作过程中，从U_M相流出电荷量为Q₁,从U_N相流出电荷量为Q₂，在4-5工作过程中，流出U_M相的电荷量为Q₃,流出U_N相的电荷量为Q₄，根据电荷量精确分配的调制策略，在一个谐振电流半周期内，使不同相流出或流入的电荷量之比等于各自的相电压绝对值之比，可得电荷分配比例：

c. The combination of low-line voltage U _k , high-line voltage U _j and zero voltage is used to complete the excitation, that is, the working mode of 6 processes is adopted, and the positive half cycle and negative half cycle of the resonant current are commutated twice and include 3 In each working process, the polarities of positive and negative half-cycle excitation voltages are opposite, specifically: the first working process uses low-line voltage U _k , the second working process uses high-line voltage U _j , and the third working process uses 0 voltage, The 4th working process adopts the low-line voltage- _Uk , the 5th working process adopts the high-line voltage- _Uj , and the 6th working process adopts 0 voltage; assuming that in the 1-2 working process, the charge flows out from the U _M phase The amount of charge flowing out from the U _N phase is Q ₁ , the amount of charge flowing out from the U N phase is Q ₂ , during the 4-5 working process, the amount of charge flowing out of the U _M phase is Q ₃ , and the amount of charge flowing out of the U _N phase is Q ₄ The distribution modulation strategy makes the ratio of the outflow or inflow of charges in different phases equal to the ratio of the absolute value of the respective phase voltages in a half cycle of the resonant current, and the charge distribution ratio can be obtained:

d.根据谐振电容电压峰值uc_max和负载电压V₀，获取顺序接入的低线电压U_k、高线电压U_j和0电压中每个电压需要接入的时间以及三个电压的切换时间点；d. According to the peak value uc _max of the resonant capacitor voltage and the load voltage V ₀ , obtain the time required for each of the sequentially connected low-line voltage U _k , high-line voltage U _j and 0 voltage and the switching time of the three voltages point;

e.根据工作时刻电网相电压所处的区间以及需要输出的电流方向，并根据步骤c所述的工作过程，分配对应的功率开关的开关状态组合；e. According to the interval of the phase voltage of the power grid at the working time and the current direction to be output, and according to the working process described in step c, assign the corresponding switching state combination of the power switch;

f.根据步骤d所得的三个电压的切换时间点生成通用的时序控制信号，控制各工作过程之间的切换；f. According to the switching time points of the three voltages obtained in step d, a general sequence control signal is generated to control the switching between each working process;

g.根据步骤f的控制完成三相相间的选择和切换，判断工作是否结束，若是，则退出，若否，则回到步骤a。g. Complete the selection and switching of the three phases according to the control of step f, judge whether the work is over, if yes, exit, if not, return to step a.

具体的，步骤d的具体方法为：Specifically, the specific method of step d is:

根据串联谐振变换器工作特性，以谐振电容电压为横轴、谐振电流i与特征阻抗Z的乘积值为纵轴构建平面直角坐标系，谐振电路特征阻抗

其中Lr为谐振电感值，Cr为谐振电容值，假设谐振电流正半周的3个工作过程对应的轨迹为分别以O₁、O₂、O₃为圆心并分别以R₁、R₂、R₃为半径的相连接的圆弧，连接点为P₁和P₂，激励电压分别为高线电压U_j，低线电压U_k和0电压，定义O₁=U_k-V₀，O₂=U_j-V₀，O₃=-V₀，在电流正半周期内，假设第1和第2工作过程对应的谐振电容电压变化量分别为Δuc₁和Δuc₂，Δuc₁和Δuc₂的比值与和这两过程对应的电荷量Q₁和Q₂的比值相等，电容电压峰值为uc_max，稳态工作时，谐振电容电压正最大值与负最大值相等，因而在电流为0的开始时刻对应谐振电容起始电压可定为-uc_max，设连接点P₁和P₂分别对应的横坐标值为u₁和u₂，即u₁为第1过程结束后谐振电容电压，u₂为第2过程结束后谐振电容电压，通过公式：According to the working characteristics of the series resonant converter, a rectangular coordinate system is constructed with the resonant capacitor voltage as the horizontal axis and the product value of the resonant current i and the characteristic impedance Z as the vertical axis, and the characteristic impedance of the resonant circuit

Among them, Lr is the resonant inductance value, Cr is the resonant capacitance value, assuming that the trajectories corresponding to the three working processes of the positive half cycle of the resonant current are respectively centered on O ₁ , O ₂ , O ₃ and centered on R ₁ , R ₂ , R ₃ is the radius of the connected circular arc, the connection points are P ₁ and P ₂ , the excitation voltages are the high-line voltage U _j , the low-line voltage U _k and the zero voltage respectively, define O ₁ =U _k -V ₀ , O ₂ = U _j -V ₀ , O ₃ =-V ₀ , in the positive half cycle of the current, it is assumed that the voltage variation of the resonant capacitor corresponding to the first and second working processes is the ratio of Δuc ₁ to Δuc ₂ , Δuc ₁ to Δuc ₂ It is equal to the ratio of the charges Q ₁ and Q ₂ corresponding to these two processes, and the peak value of the capacitor voltage is uc _max . In steady state operation, the positive and negative maximum values of the resonant capacitor voltage are equal, so at the beginning moment when the current is 0 The initial voltage of the corresponding resonant capacitor can be set as -uc _max , and the abscissa values corresponding to the connection points P ₁ and P ₂ are respectively u ₁ and u ₂ , that is, u ₁ is the voltage of the resonant capacitor after the first process, and u ₂ is After the end of the second process, the resonant capacitor voltage, through the formula:

$\{\begin{matrix} {R R}_{22}^{22} - - {(({O o}_{22} - - {u u}_{11}))}^{22} + + {(({O o}_{11} - - {u u}_{11}))}^{22} = = {R R}_{11}^{22} \\ {R R}_{22}^{22} - - {(({u u}_{22} - - {O o}_{22}))}^{22} + + {(({u u}_{22} - - {O o}_{33}))}^{22} = = {R R}_{33}^{22} \\ {uc uc}_{max max} = = {R R}_{11} - - {O o}_{11} \\ {R R}_{33} = = {uc uc}_{max max} - - {O o}_{33} = = {R R}_{11} - - {O o}_{11} - - {O o}_{33} \end{matrix}$

前两项可得：The first two are available:

$\{\begin{matrix} {(({u u}_{11} - - {O o}_{22}))}^{22} - - {(({u u}_{11} - - {O o}_{11}))}^{22} = = {R R}_{22}^{22} - - {R R}_{11}^{22} \\ {(({u u}_{22} - - {O o}_{33}))}^{22} - - {(({u u}_{22} - - {O o}_{22}))}^{22} = = {(({R R}_{11} - - {O o}_{11} - - {O o}_{33}))}^{22} - - {R R}_{22}^{22} \end{matrix}$

简化后得：After simplification:

${u u}_{11} = = [[\frac{{R R}_{22}^{22} - - {R R}_{11}^{22}}{{O o}_{11} - - {O o}_{22}} + + (({O o}_{11} + + {O o}_{22}))]] / / 22$

${u u}_{22} = = [[\frac{{(({R R}_{11} - - {O o}_{11} - - {O o}_{33}))}^{22} - - {R R}_{22}^{22}}{{O o}_{22} - - {O o}_{33}} + + (({O o}_{22} + + {O o}_{33}))]] / / 22$

根据电荷分配约束条件：According to the charge distribution constraints:

$K K = = \frac{{Q Q}_{11}}{{Q Q}_{22}} = = \frac{CΔ CΔ {uc uc}_{11}}{CΔ CΔ {uc uc}_{22}} = = \frac{{u u}_{11} - - ((- - {uc uc}_{max max}))}{{u u}_{22} - - {u u}_{11}}$

可得：Available:

${R R}_{22} = = \sqrt{\frac{K K (({O o}_{11} - - {O o}_{22})) (({uc uc}^{22}_{max max} + + {O o}_{22}^{22} - - 22 {uc uc}_{max max} {O o}_{33})) + + ((11 + + K K)) (({O o}_{22} - - {O o}_{33})) (({uc uc}^{22}_{max max} = = {O o}_{22}^{22} + + {22 uc uc}_{max max} {O o}_{11})) - - 22 {uc uc}_{max max} (({O o}_{11} - - {O o}_{22})) (({O o}_{22} - - {O o}_{33}))}{(({O o}_{22} - - {O o}_{33})) + + K K (({O o}_{11} - - {O o}_{33}))}}$

根据uc_max、O₁、O₂、O₃和K值可得到u₁和u₂的值；The values of u ₁ and u ₂ can be obtained according to uc _max , O ₁ , O ₂ , O ₃ and K;

设第一工作过程轨迹对应的弧度为θ₁、第二工作过程轨迹对应的弧度为θ₂、第三工作过程轨迹对应的弧度为θ₃，其对应的表达式分别为：Assuming that the radian corresponding to the first working process trajectory is θ ₁ , the corresponding radian of the second working process trajectory is θ ₂ , and the corresponding radian of the third working process trajectory is θ ₃ , the corresponding expressions are respectively:

${θ θ}_{11} = = {cos cos}^{- - 11} ((\frac{{O o}_{11} - - {u u}_{11}}{{R R}_{11}}))$

${θ θ}_{22} = = π π - - {cos cos}^{- - 11} (({θ θ}_{22__11})) - - {cos cos}^{- - 11} (({θ θ}_{22__22})) = = π π - - {cos cos}^{- - 11} ((\frac{{u u}_{22} - - {O o}_{22}}{{R R}_{22}})) - - {cos cos}^{- - 11} ((\frac{{O o}_{22} - - {u u}_{11}}{{R R}_{22}}))$

${θ θ}_{33} = = {cos cos}^{- - 11} ((\frac{{u u}_{22} - - {O o}_{33}}{{R R}_{33}}))$

根据θ=ωt，可得：第一工作过程结束时间t₁＝θ₁/ω，第二工作过程结束时间t₂＝(θ₁+θ₂)/ω，第三工作过程结束时间t₃＝(θ₁+θ₂+θ₃)/ω；其中ω为谐振角频率，

可分别得到切换点时间t₁、t₂和t₃。According to θ=ωt, it can be obtained: the end time of the first working process t ₁ = θ ₁ /ω, the ending time of the second working process t ₂ =(θ ₁ +θ ₂ )/ω, the ending time of the third working process t ₃ = (θ ₁ +θ ₂ +θ ₃ )/ω; where ω is the resonant angular frequency,

Switching point times t ₁ , t ₂ and t ₃ can be obtained respectively.

一种AC-DC串联谐振矩阵变换器的控制装置，包括三相电源2、滤波器3、EMI滤波器4、整流硅堆17和滤波电路18，其特征在于，还包括开关矩阵1、触发驱动电路16、故障保护电路、过零比较器电路5、相位检测单元6、电压采集模块15、负载电压采集电路11、闭环控制单元10、控制参数计算单元9、时序生成单元8和开关状态控制单元7，所述故障保护电路包括电网故障检测电路12、过流检测电路13和过温检测电路14；所述三相电源2的相电压通过滤波器3与开关矩阵1连接，三相电源2的零电压和相电压通过EMI滤波器4分别与电网故障检测电路12、过零比较器电路5和电压采集模块15连接；过零比较器电路5与相位检测单元6的输入端连接，相位检测单元6的一个输出端和电压采集模块15的输出端与控制参数计算单元9的输入端连接，控制参数计算单元9的输出端与时序生成单元8的输入端连接，时序生成单元8的输出端、相位检测单元6的另一个输出端以及过流检测电路13和过温检测电路14的输出端与开关状态控制单元7的输入端连接，开关状态控制单元7的输出端与触发驱动电路16的输入端连接，开关矩阵1连接触发驱动电路16的输出端、整流硅堆17的一端和过流检测电路13的输入端，过温检测电路14的输入端连接整流硅堆17，整流硅堆17的另一端连接滤波电路18和负载电压采集电路11，负载电压采集电路11的输出端连接闭环控制器10的输入端，闭环控制器10的输出端连接控制参数计算单元9的输入端；其中，A control device for an AC-DC series resonant matrix converter, including a three-phase power supply 2, a filter 3, an EMI filter 4, a rectifying silicon stack 17 and a filter circuit 18, is characterized in that it also includes a switch matrix 1, a trigger drive Circuit 16, fault protection circuit, zero-crossing comparator circuit 5, phase detection unit 6, voltage acquisition module 15, load voltage acquisition circuit 11, closed-loop control unit 10, control parameter calculation unit 9, timing generation unit 8 and switch state control unit 7. The fault protection circuit includes a power grid fault detection circuit 12, an overcurrent detection circuit 13 and an overtemperature detection circuit 14; the phase voltage of the three-phase power supply 2 is connected to the switch matrix 1 through a filter 3, and the three-phase power supply 2 Zero voltage and phase voltage are respectively connected with grid fault detection circuit 12, zero-crossing comparator circuit 5 and voltage acquisition module 15 through EMI filter 4; Zero-crossing comparator circuit 5 is connected with the input end of phase detection unit 6, and phase detection unit An output end of 6 and an output end of the voltage acquisition module 15 are connected to the input end of the control parameter calculation unit 9, the output end of the control parameter calculation unit 9 is connected to the input end of the timing generation unit 8, the output end of the timing generation unit 8, The other output terminal of the phase detection unit 6 and the output terminals of the overcurrent detection circuit 13 and the overtemperature detection circuit 14 are connected to the input terminal of the switch state control unit 7, and the output terminal of the switch state control unit 7 is connected to the input of the trigger drive circuit 16 The switch matrix 1 is connected to the output end of the trigger drive circuit 16, one end of the rectifier silicon stack 17 and the input end of the overcurrent detection circuit 13, the input end of the overtemperature detection circuit 14 is connected to the rectification silicon stack 17, and the rectification silicon stack 17 The other end is connected to the filter circuit 18 and the load voltage acquisition circuit 11, the output end of the load voltage acquisition circuit 11 is connected to the input end of the closed-loop controller 10, and the output end of the closed-loop controller 10 is connected to the input end of the control parameter calculation unit 9; wherein,

过零比较器电路5将三相电源2输入的各相电压通过与零线的过零比较转变为与各相电压极性一致的数字信号，该数字信号经数字滤波之后传输给相位检测单元6和开关状态控制单元7；The zero-crossing comparator circuit 5 converts each phase voltage input by the three-phase power supply 2 into a digital signal with the same polarity as each phase voltage through zero-crossing comparison with the zero line, and the digital signal is transmitted to the phase detection unit 6 after digital filtering and switch state control unit 7;

负载电压采集电路11采集实际输出到负载部分的电压并传递到闭环控制单元10，闭环控制单元10根据定值与实际输出值比较和计算得到控制量并输入给控制参数计算单元9；The load voltage acquisition circuit 11 collects the voltage actually output to the load part and transmits it to the closed-loop control unit 10, and the closed-loop control unit 10 compares and calculates the control quantity according to the fixed value and the actual output value and inputs it to the control parameter calculation unit 9;

相位检测单元6对电网极性信号进行跟踪和同步，对极性信号宽度进行测量以识别电网是否有故障，根据极性信号的前后变化得到电网相序，同步计数器值与电网的相位对应，根据此值可以间接得到电网的相位，用于辅助处理同一电网极性下的不同区间，并得到不同时刻需要的电荷分析比例K，相位检测单元6得到的数据传输给控制参数计算单元9和开关状态控制单元7；The phase detection unit 6 tracks and synchronizes the polarity signal of the power grid, measures the width of the polarity signal to identify whether there is a fault in the power grid, and obtains the phase sequence of the power grid according to the change of the polarity signal before and after, and the value of the synchronization counter corresponds to the phase of the power grid. This value can indirectly obtain the phase of the power grid, which is used to assist in the processing of different intervals under the same grid polarity, and to obtain the charge analysis ratio K required at different times. The data obtained by the phase detection unit 6 is transmitted to the control parameter calculation unit 9 and the switch state control unit 7;

控制参数计算单元9根据接收到的相位检测单元6和电压采集模块15传输的数据，得出当前状态下的电流周期和需要切换相的时间，将得到的时间控制量传输到时序生成单元8；The control parameter calculation unit 9 obtains the current cycle in the current state and the time to switch phases according to the received data transmitted by the phase detection unit 6 and the voltage acquisition module 15, and transmits the obtained time control amount to the timing generation unit 8;

时序生成单元8根据接收到的信号产生控制开关的时序信号，并将控制信号传输到开关状态控制单元7；The timing generation unit 8 generates a timing signal for controlling the switch according to the received signal, and transmits the control signal to the switch state control unit 7;

开关状态控制单元7根据接收到的时序信号和相位检测单元6提供的数据，判断电网所处区间，以及正负交替的电流输出方向选择与时序生成单元8输出的12路信号相对应的开关编号，开关选择模块7输出的与实际开关位置相对应的控制信号经开关驱动电路16驱动开关矩阵1完成变换器主电路的控制。The switch state control unit 7 judges the interval of the power grid according to the received timing signal and the data provided by the phase detection unit 6, and selects the switch number corresponding to the 12-way signal output by the timing generating unit 8 for the positive and negative alternating current output direction The control signal corresponding to the actual switch position output by the switch selection module 7 drives the switch matrix 1 through the switch drive circuit 16 to complete the control of the main circuit of the converter.

本发明的有益效果为，将矩阵变换器用于大功率直流电源，提高了整体的功率密度。采用串联谐振过谐振的工作模式与同等功率下采用断续模式相比：满足同样的纹波需求时需要的滤波电容容量更小，调整速率更快；而且主回路电流峰值降低一半，降低了开关的电流应力；同时当前商品化封装好的双向开关的电流为400A等级，可以正好适用采用当前方案的100kW的装置；同时提出了在高频电流半个周期内，采用激励电压先从低线电压切换到高线电压，然后再切换到0电压的控制策略，实现了3电压的瞬时合成，在实现等效激励电压调节的同时也使得每相输入线电流的平均值正比于相电压，只需较小滤波电感值即可实现高的功率因数和低谐波的电流，意味着电感的尺寸、重量和损耗大为减小。The beneficial effect of the invention is that the matrix converter is used for a high-power direct-current power supply, and the overall power density is improved. Compared with the discontinuous mode at the same power, the working mode of series resonance and over-resonance requires smaller filter capacitor capacity and faster adjustment speed to meet the same ripple demand; and the peak value of the main circuit current is reduced by half, reducing switching At the same time, the current of the current commercially packaged bidirectional switch is 400A, which can be just suitable for the 100kW device using the current scheme; at the same time, it is proposed that the excitation voltage should start from the low-line voltage within half a cycle of the high-frequency current. The control strategy of switching to high line voltage and then switching to 0 voltage realizes the instantaneous synthesis of 3 voltages. While realizing the regulation of equivalent excitation voltage, it also makes the average value of the input line current of each phase proportional to the phase voltage. Small filter inductor values can achieve high power factor and low harmonic current, which means that the size, weight and loss of the inductor are greatly reduced.

附图说明Description of drawings

图1是AC-DC矩阵变换器的拓扑结构；Figure 1 is the topology of the AC-DC matrix converter;

图2是正半周工作状态图；Fig. 2 is a positive half-cycle working state diagram;

图3是包含6个工作过程的示意图；Fig. 3 is a schematic diagram comprising 6 working processes;

图4是一个周期内的开关控制和换流时序图；Figure 4 is a sequence diagram of switch control and commutation within one cycle;

图5是电网相电压工作区间划分图；Fig. 5 is a division diagram of the grid phase voltage working interval;

图6是本发明的控制方法FPGA实现框图；Fig. 6 is a control method FPGA realization block diagram of the present invention;

图7是本发明的控制装置结构示意图。Fig. 7 is a schematic structural diagram of the control device of the present invention.

具体实施方式Detailed ways

下面结合附图，详细描述本发明的技术方案：Below in conjunction with accompanying drawing, describe technical scheme of the present invention in detail:

对于串联谐振变换器而言，由输出滤波电容与负载并联组成的负载回路与谐振回路串联，谐振电流完全流经负载回路，通过对谐振电流的调节从而实现输出电压的调节与稳定。由于谐振电流完全流经谐振电容，且电容电压变化量与电流积分值成正比，假设稳态时电流周期时间基本恒定，那么每个周期电流平均值也与电容电压变化量成正比，本发明以谐振电容电压峰值(uc_max)作为控制量表征谐振回路的工作状态；闭环控制根据实时采集到的负载电压与设定电压进行闭环控制运算得到需要的控制量uc_max。For the series resonant converter, the load circuit composed of the output filter capacitor connected in parallel with the load is connected in series with the resonant circuit, the resonant current flows completely through the load circuit, and the output voltage is adjusted and stabilized by adjusting the resonant current. Since the resonant current flows completely through the resonant capacitor, and the capacitance voltage change is proportional to the current integral value, assuming that the current cycle time is basically constant in a steady state, then the average value of the current in each cycle is also proportional to the capacitance voltage change. The present invention uses The peak value of the resonant capacitor voltage (uc _max ) is used as the control quantity to characterize the working state of the resonant circuit; the closed-loop control is based on the real-time collected load voltage and the set voltage to perform closed-loop control calculation to obtain the required control quantity uc _max .

本发明的具体控制方法为：Concrete control method of the present invention is:

①因为串接在回路中的负载电压(等效为电压源)已经通过测量得到，而谐振参数一定，为了实现本发明所述期望的工作状态，就需要调节等效的激励电压；当前可用于激励的电压为电网相电压的组合，根据实时采集到的三相输入相电压u_a，u_b，u_c的相对大小关系，将每个输入相电压周期划分为12个区间，每个区间内相电压的极性和大小确定，且保持单调变化，所述12个区间具体为：①Because the load voltage (equivalent to a voltage source) connected in series in the loop has been obtained by measurement, and the resonance parameter is fixed, in order to achieve the desired working state of the present invention, it is necessary to adjust the equivalent excitation voltage; currently available for The excitation voltage is a combination of grid phase voltages. According to the relative size relationship of the three-phase input phase voltage u _a , u _b , u _c collected in real time, each input phase voltage cycle is divided into 12 intervals. The polarity and magnitude of the phase voltage are determined and kept changing monotonously. The 12 intervals are specifically:

②采用低线电压(U_k)、高线电压(U_j)以及0电压共同参与的组合方式完成激励，即采用6过程的工作模式，谐振电流正半周和负半周均进行2次换流且均包含3个工作过程，正负半周激励电压的极性相反，具体为：第1个工作过程采用低线电压U_k，第2个工作过程采用高线电压U_j，第3个工作过程采用0电压，第4个工作过程采用低线电压-U_k，第5个工作过程采用高线电压-U_j，第6个工作过程采用0电压；假设在1-2工作过程中，从U_M相流出电荷量为Q₁,从U_N相流出电荷量为Q₂，在4-5工作过程中，流出U_M相的电荷量为Q₃,流出U_N相的电荷量为Q₄，根据电荷量精确分配的调制策略，在一个谐振电流半周期内，使不同相流出或流入的电荷量之比等于各自的相电压绝对值之比，比例K值由查表得到，电荷分配比例为：②The combination of low-line voltage (U _k ), high-line voltage (U _j ) and zero voltage is used to complete the excitation, that is, the working mode of 6 processes is adopted, and the positive half cycle and negative half cycle of the resonant current are commutated twice and Both include 3 working processes, and the polarities of positive and negative half-cycle excitation voltages are opposite, specifically: the first working process uses low-line voltage U _k , the second working process uses high-line voltage U _j , and the third working process uses 0 voltage, the 4th working process adopts low line voltage -U _k , the 5th working process adopts high line voltage -U _j , and the 6th working process adopts 0 voltage; suppose in 1-2 working process, from U _M The amount of charge flowing out of phase U is Q ₁ , and the amount of charge flowing out from phase U _N is Q ₂ . During the 4-5 working process, the amount of charge flowing out of phase U _M is Q ₃ , and the amount of charge flowing out of phase U _N is Q ₄ The modulation strategy for the precise distribution of charge, within a half cycle of the resonant current, makes the ratio of the outflow or inflow of different phases equal to the ratio of the absolute value of the respective phase voltages. The ratio K value is obtained from the look-up table, and the charge distribution ratio is:

$\frac{{Q Q}_{11}}{{Q Q}_{22}} = = \frac{{Q Q}_{33}}{{Q Q}_{44}} = = K K . .$

③采用状态图作为分析串联谐振3电压瞬时合成作为激励源的方法，并用此方法得到控制参数。③Use the state diagram as a method to analyze the instantaneous synthesis of the series resonance 3 voltage as the excitation source, and use this method to obtain the control parameters.

变量说明：激励电压可选择的有三个：高线电压U_j，低线电压U_k和0电压；负载电压等效到变压器初级为V₀，谐振电容电压为uc，u₁为第1过程结束后谐振电容电压，u₂为第2过程结束后谐振电容电压，谐振电容电压峰值为uc_max，谐振电流为i，谐振电路特征阻抗为Z，谐振角频率为ω，相位角为θ，工作持续时间为t，Lr为谐振电感值，Cr为谐振电容值。Variable description: There are three options for the excitation voltage: high line voltage U _j , low line voltage U _k and 0 voltage; the load voltage is equivalent to the primary of the transformer as V ₀ , the resonant capacitor voltage is uc, and u ₁ is the end of the first process After the resonant capacitor voltage, u ₂ is the resonant capacitor voltage after the second process, the peak value of the resonant capacitor voltage is uc _max , the resonant current is i, the characteristic impedance of the resonant circuit is Z, the resonant angular frequency is ω, the phase angle is θ, and the work continues The time is t, Lr is the resonant inductance value, and Cr is the resonant capacitance value.

其中 $ω = 1 / \sqrt{Lr \cdot Cr}, Z = \sqrt{Lr / Cr};$ in $ω = 1 / \sqrt{Lr &Center Dot; Cr}, Z = \sqrt{Lr / Cr};$

定义O₁=U_k-V₀，O₂=U_j-V₀，O₃=-V₀； (1)Define O ₁ =U _k -V ₀ , O ₂ =U _j -V ₀ , O ₃ =-V ₀ ; (1)

电流正半周工作过程对应的状态图如图2所示，横轴为谐振电容电压，纵轴为谐振电流i与特征阻抗Z的乘积值；l₁，l₂，l₃为电流正半周的三个工作过程对应的轨迹，分别以O₁，O₂，O₃为圆心，分别以R₁，R₂，R₃为半径的相连接的圆弧；在电流正半周期内，Δuc₁和Δuc₂分别为第1和第2工作过程对应的谐振电容电压变化量，比值与和这两过程对应的电荷量Q₁和Q₂的比值相等。The state diagram corresponding to the working process of the positive half cycle of the current is shown in Figure _2. _The _horizontal axis is the resonant capacitor voltage, and the vertical axis is the product value of the resonant current i and the characteristic impedance Z; The trajectories corresponding to each working process are connected circular arcs with O ₁ , O ₂ , and O ₃ as centers and R ₁ , R ₂ , and R ₃ as radii; in the positive half cycle of the current, Δuc ₁ and Δuc ₂ are the voltage variations of the resonant capacitor corresponding to the first and second working processes, and the ratio is equal to the ratio of the charges Q ₁ and Q ₂ corresponding to the two processes.

状态图(图2)上圆心(O₁，O₂，O₃)参数根据b所述方法和表达式(1)得到且是基本稳定的已知量，电容电压峰值uc_max是闭环控制得到控制量也是的已知量；稳态工作时，谐振电容电压正最大值与负最大值相等，因而在电流为0的开始时刻对应谐振电容起始电压可定为-uc_max。那么图2中半径R₁和R₃可知，如果半径R₂也确定，那么圆弧交点P₁和P₂就确定，从而状态图参数全部确定；The parameters of the circle center (O ₁ , O ₂ , O ₃ ) on the state diagram (Figure 2) are obtained according to the method described in b and expression (1) and are basically stable known quantities. The peak value of the capacitor voltage uc _max is controlled by closed-loop control The quantity is also a known quantity; in steady state operation, the positive and negative maximum values of the resonant capacitor voltage are equal, so the corresponding initial voltage of the resonant capacitor at the beginning of the current is 0 can be set as -uc _max . Then the radii R ₁ and R ₃ in Fig. 2 show that if the radius R ₂ is also determined, then the arc intersection points P ₁ and P ₂ are determined, so that the parameters of the state diagram are all determined;

R₂值的关系着P₁和P₂的位置(u₁和u₂)，而这两点的位置关系受到电荷分配条件(Δuc₁和Δuc₂存在一定比例关系)的限制，采用R₂表示u₁和u₂，再代入到比例限制条件中便可计算得到R₂，状态图确定后，根据状态图上的几何关系可以计算得到每段圆弧(过程)对应的相位角度θ，根据θ=ωt从而获得时间控制参数3个时间节点t₁～t₃。The value of R ₂ is related to the positions of P ₁ and P ₂ (u ₁ and u ₂ ), and the positional relationship between these two points is limited by the charge distribution conditions (there is a certain proportional relationship between Δuc ₁ and Δuc ₂ ), which is represented by R ₂ u ₁ and u ₂ can be calculated by substituting u 1 _{and u 2} into the proportional constraints. After the state diagram is determined, the phase angle θ corresponding to each arc (process) can be calculated according to the geometric relationship on the state diagram. According to θ =ωt to obtain three time nodes t ₁ ˜t ₃ of time control parameters.

状态图稳定的几何约束关系：The geometric constraints of state diagram stability:

$\{\begin{matrix} {R R}_{22}^{22} - - {(({O o}_{22} - - {u u}_{11}))}^{22} + + {(({O o}_{11} - - {u u}_{11}))}^{22} = = {R R}_{11}^{22} \\ {R R}_{22}^{22} - - {(({u u}_{22} - - {O o}_{22}))}^{22} + + {(({u u}_{22} - - {O o}_{33}))}^{22} = = {R R}_{33}^{22} \\ {uc uc}_{max max} = = {R R}_{11} - - {O o}_{11} \\ {R R}_{33} = = {uc uc}_{max max} - - {O o}_{33} = = {R R}_{11} - - {O o}_{11} - - {O o}_{33} \end{matrix} - - - - - - ((22))$

式(2)前两项整理后可得：After rearranging the first two terms of formula (2), we can get:

$\{\begin{matrix} {(({u u}_{11} - - {O o}_{22}))}^{22} - - {(({u u}_{11} - - {O o}_{11}))}^{22} = = {R R}_{22}^{22} - - {R R}_{11}^{22} \\ {(({u u}_{22} - - {O o}_{33}))}^{22} - - {(({u u}_{22} - - {O o}_{22}))}^{22} = = {(({R R}_{11} - - {O o}_{11} - - {O o}_{33}))}^{22} - - {R R}_{22}^{22} \end{matrix} - - - - - - ((33))$

对式(3)简化后可得表达式：The expression (3) can be obtained after simplification:

${u u}_{11} = = [[\frac{{R R}_{22}^{22} - - {R R}_{11}^{22}}{{O o}_{11} - - {O o}_{22}} + + (({O o}_{11} + + {O o}_{22}))]] / / 22 - - - - - - ((44))$

${u u}_{22} = = [[\frac{{(({R R}_{11} - - {O o}_{11} - - {O o}_{33}))}^{22} - - {R R}_{22}^{22}}{{O o}_{22} - - {O o}_{33}} + + (({O o}_{22} + + {O o}_{33}))]] / / 22 - - - - - - ((55))$

电荷分配约束条件：Charge distribution constraints:

$K K = = \frac{{Q Q}_{11}}{{Q Q}_{22}} = = \frac{CΔ CΔ {uc uc}_{11}}{CΔ CΔ {uc uc}_{22}} = = \frac{{u u}_{11} - - ((- - {uc uc}_{max max}))}{{u u}_{22} - - {u u}_{11}} - - - - - - ((66))$

把式(4)和式(5)带入到式(6)中，经整理后可得R₂表达式(7)如下：Bringing formula (4) and formula (5) into formula (6), after arrangement, the expression (7) of _R2 can be obtained as follows:

其中uc_max已知；O₁，O₂，O₃由式(1)得到；R₁=uc_max+O₁，R₃=uc_max-O₃；K由查表得到，因而根据式(6)可计算得到关键中间量R₂。Among them, uc _max is known; O ₁ , O ₂ , O ₃ are obtained by formula (1); R ₁ = uc _max + O ₁ , R ₃ = uc _max -O ₃ ; K is obtained by looking up the table, so according to formula (6 ) can be calculated to obtain the key intermediate quantity R ₂ .

将式(7)得到的R₂值代入式(5)可得u₂值；Substituting the _R2 value obtained by formula (7) into formula (5) can obtain the _u2 value;

将式(6)整理后可得：After rearranging formula (6), we can get:

${u u}_{11} = = \frac{{Ku Ku}_{22} - - {u u}_{c c__max max}}{11 + + K K} - - - - - - ((88))$

将式(5)得到的u₂值代入式(8)可得u₁；Substituting the u ₂ value obtained in formula (5) into formula (8) can get u ₁ ;

状态图上各圆弧(l₁～l₃)对应角度的表达式如下：The expression of the angle corresponding to each arc (l ₁ ~ l ₃ ) on the state diagram is as follows:

${θ θ}_{11} = = {cos cos}^{- - 11} ((\frac{{O o}_{11} - - {u u}_{11}}{{R R}_{11}})) - - - - - - ((99))$

${θ θ}_{22} = = π π - - {cos cos}^{- - 11} (({θ θ}_{22__11})) - - {cos cos}^{- - 11} (({θ θ}_{22__22})) = = π π - - {cos cos}^{- - 11} ((\frac{{u u}_{22} - - {O o}_{22}}{{R R}_{22}})) - - {cos cos}^{- - 11} ((\frac{{O o}_{22} - - {u u}_{11}}{{R R}_{22}})) - - - - - - ((1010))$

${θ θ}_{33} = = {cos cos}^{- - 11} ((\frac{{u u}_{22} - - {O o}_{33}}{{R R}_{33}})) - - - - - - ((1111))$

根据θ=ωt可得：According to θ=ωt:

t₁＝θ₁/ω (12)t ₁ =θ ₁ /ω (12)

t₂＝(θ₁+θ₂)/ω (13)t ₂ =(θ ₁ +θ ₂ )/ω (13)

t₃＝(θ₁+θ₂+θ₃)/ω (14)t ₃ =(θ ₁ +θ ₂ +θ ₃ )/ω (14)

根据式(12)～(14)求得控制所需的切换点时间。According to the formula (12) ~ (14) to obtain the switching point time required for control.

④根据工作时刻电网相电压所处的区间，按②所述的工作过程，按照③求解得到切换点时间完成一个电流周期的控制；④According to the interval of the phase voltage of the power grid at the working time, according to the working process described in ②, according to ③ to obtain the switching point time to complete the control of a current cycle;

除同极性之间切换的短时间外，矩阵开关在其他时刻对谐振回路均保持双向连通；设“1”表示开通，“0”表示关断，在不同条件下，有如下表1-表4的开关信号。Except for the short time of switching between the same polarity, the matrix switch maintains two-way connection to the resonant circuit at other times; set "1" to indicate opening, and "0" to indicate off. Under different conditions, there are the following table 1-table 4 switching signals.

表1第1到3区间双向功率开关状态组合次序表(开关位置见附图1)Table 1 Combination sequence of bidirectional power switch states in intervals 1 to 3 (see attached drawing 1 for switch positions)

表2第4到6区间双向功率开关状态组合次序表Table 2 Combination sequence table of bidirectional power switch states in the 4th to 6th intervals

表3第7到9区间双向功率开关状态组合次序表Table 3 Combination sequence table of bidirectional power switch states in the 7th to 9th intervals

表4第10到12区间双向功率开关状态组合次序表Table 4 Combination sequence table of bidirectional power switch states in the 10th to 12th intervals

⑤采用电压型“两步换流”和“四步换流”策略针对不同工作过程进行切换，谐振回路的激励电压是由上臂电压(H₁点)和下臂电压(H₂)差实现的(见图1和图3)，对谐振回路不同的激励电压对应着不同的上臂和下臂电压的组合；换流可以以臂为单位进行三相相间的选择和切换，上臂由S₁～S₆组成，下臂由S₇～S₁₂组成，电网相电压工作区间划分如图5所示，以电网状态在第1区间，完整电流周期为例，上臂和下臂包含换流的开关状态如下：⑤Using voltage-type "two-step commutation" and "four-step commutation" strategies to switch between different working processes, the excitation voltage of the resonant circuit is realized by the difference between the upper arm voltage (H ₁ point) and the lower arm voltage (H ₂ ) (See Figure 1 and Figure 3), different excitation voltages for the resonant circuit correspond to different combinations of upper and lower arm voltages; the commutation can be selected and switched between three phases in units of arms, and the upper arm consists of S ₁ to S ₆ components, the lower arm is composed of S ₇ ~ S ₁₂ , the phase voltage working interval of the grid is shown in Figure 5, taking the grid state in the first interval and the complete current cycle as an example, the switch states of the upper arm and the lower arm including commutation are as follows :

上臂：upper arm:

过程1～过程3：(S₁+S₂+S₄+S₆)；不需要换流；Process 1～Process 3: (S ₁ +S ₂ +S ₄ +S ₆ ); no commutation required;

过程3(S₁+S₂+S₄+S₆)→(S₁+S₆)→过程4(S₁+S₅+S₆)；需要2步完成换流；Process 3(S ₁ +S ₂ +S ₄ +S ₆ )→(S ₁ +S ₆ )→Process 4(S ₁ +S ₅ +S ₆ ); two steps are required to complete the commutation;

过程4(S₁+S₅+S₆)→(S₁+S₅)→(S₁+S₃+S₅)→(S₁+S₃)→过程5(S₁+S₃+S₄)；需要4步完成换流；Process 4(S ₁ +S ₅ +S ₆ )→(S ₁ +S ₅ )→(S ₁ +S ₃ +S ₅ )→(S ₁ +S ₃ )→Process 5(S ₁ +S ₃ +S ₄ ); 4 steps are required to complete the commutation;

过程5(S₁+S₃+S₄)→(S₁+S₄)→过程6(S₁+S₂+S₄+S₆)；需要2步完成换流；Process 5(S ₁ +S ₃ +S ₄ )→(S ₁ +S ₄ )→Process 6(S ₁ +S ₂ +S ₄ +S ₆ ); two steps are required to complete the commutation;

下臂：lower arm:

过程1：(S₈+S₁₁+S₁₂)→(S₈+S₁₂)→(S₈+S₁₀+S₁₂)→(S₈+S₁₀)→过程2(S₈+S₉+S₁₀)；需要4步完成换流；Process 1: (S ₈ +S ₁₁ +S ₁₂ )→(S ₈ +S ₁₂ )→(S ₈ +S ₁₀ +S ₁₂ )→(S ₈ +S ₁₀ )→Process 2 (S ₈ +S ₉ + S ₁₀ ); need 4 steps to complete the commutation;

过程2(S₈+S₉+S₁₀)→(S₈+S₉)→过程3(S₇+S₈+S₉+S₁₁)；需要2步完成换流；Process 2(S ₈ +S ₉ +S ₁₀ )→(S ₈ +S ₉ )→Process 3(S ₇ +S ₈ +S ₉ +S ₁₁ ); two steps are required to complete the commutation;

过程3～过程6：(S₇+S₈+S₉+S₁₁)；不需要换流；Process 3～Process 6: (S ₇ +S ₈ +S ₉ +S ₁₁ ); no commutation required;

为了便于开关状态的控制，把开关状态控制功能模块分为两个子模块：时序生成模块和开关选择模块；时序生成模块根据d所述计算得到的t₁～t₃生成与电网状态和电流方向无关的，但包含有换流操作的12路时序信号，如图4所示；开关选择模块根据当前电网所处区间，以及电流方向选择相应的开关与上述12路信号相连接，以第一区间为例，开关选择模块的选择结果如下：In order to facilitate the control of the switch state, the switch state control function module is divided into two sub-modules: the timing generation module and the switch selection module; the generation of t ₁ ~ t ₃ obtained by the timing generation module according to the calculation described in d has nothing to do with the grid state and current direction However, it includes 12 channels of timing signals for commutation operations, as shown in Figure 4; the switch selection module selects the corresponding switch to connect to the above 12 channels of signals according to the section of the current power grid and the direction of the current, taking the first section as For example, the selection result of the switch selection module is as follows:

上臂：S₁=up_max₂,S₂=up_max₁,S₃=up_mid₁,S₄=up_mid₂,S₅=up_min₁,S₆=up_min₂；Upper arm: S ₁ =up_max ₂ , S ₂ =up_max ₁ , S ₃ =up_mid ₁ , S ₄ =up_mid ₂ , S ₅ =up_min ₁ , S ₆ =up_min ₂ ;

下臂：S₇=dn_max₁,S₈=dn_max₂,S₉=dn_mid₂,S₁₀=dn_mid₁,S₁₁=dn_min₂,S₁₂=dn_min₁；Lower arm: S ₇ =dn_max ₁ , S ₈ =dn_max ₂ , S ₉ =dn_mid ₂ , S ₁₀ =dn_mid ₁ , S ₁₁ =dn_min ₂ , S ₁₂ =dn_min ₁ ;

⑥步骤⑤完成后返回步骤②，直至工作结束。⑥After step ⑤ is completed, return to step ② until the work is finished.

一种直流稳压输出的AC-DC串联谐振矩阵变换器控制装置，如图7所示，包括串联谐振回路、三相电源2、滤波器3、开关矩阵1、串联谐振回路、高频变压器与硅堆17、输出滤波电路18组成；控制系统由信号滤波器4、过零比较电路5、电网电压检测电路15、电网故障检测电路12、负载电压采集电路11、开关矩阵驱动电路16、过流检测电路13、过温检测电路14和以FPGA为控制核心的的控制器6～10；控制器内部由相位检测单元6、开关选择模块7、时序生成单元8、控制参数计算单元9和闭环控制单元10组成。An AC-DC series resonant matrix converter control device with DC stabilized voltage output, as shown in Figure 7, including a series resonant circuit, a three-phase power supply 2, a filter 3, a switch matrix 1, a series resonant circuit, a high-frequency transformer and Silicon stack 17, output filter circuit 18; control system consists of signal filter 4, zero-crossing comparison circuit 5, grid voltage detection circuit 15, grid fault detection circuit 12, load voltage acquisition circuit 11, switch matrix drive circuit 16, overcurrent Detection circuit 13, over-temperature detection circuit 14, and controllers 6-10 with FPGA as the control core; inside the controller are phase detection unit 6, switch selection module 7, timing generation unit 8, control parameter calculation unit 9 and closed-loop control Unit 10 is composed.

所述三相电源2通过滤波器3与开关矩阵1连接，开关矩阵1与串联谐振电路连接、串联谐振电路与变压器和整流硅堆17连接后经输出滤波电路18连接负载。三相电源2的中性线和三相电压通过EMI滤波器4分别与电网故障检测电路12、过零比较器电路5和电压采集模块15连接；过零比较器电路5与相位检测单元6的输入端连接，相位检测单元6的一个输出端和控制参数计算单元9连接，用于查表得到K所需的地址；负载电压采集电路11连接闭环控制单元10，闭环控制单元根据设定值与实际输出值比较和计算得到控制量并输入给控制参数计算单元9，控制参数计算单元9根据电网电压检测电路15得到的电网电压以及查表得到的K值计算出控制所需的时间控制量，并输送给时序生成单元8，时序生成单元8按照一定换流时序生成12路控制信号并输送给开关选择模块7，开关选择模块7根据相位检测单元6判断电网所处区间，以及正负交替的电流输出方向选择与时序生成单元8输出的12路信号相对应的开关编号，开关选择模块7输出的与实际开关位置相对应的控制信号经开关驱动电路16驱动开关矩阵1完成变换器主电路的控制。故障检测电路包括电网故障检测电路12检测到电网异常、过流检测电路13检测到电流异常以及过温检测电路14检测到温度过高后，将故障信号输送给开关选择模块7，开关选择模块7检测到有故障信号后立即锁存故障并关断所有开关完成保护动作的执行。The three-phase power supply 2 is connected to the switch matrix 1 through the filter 3, the switch matrix 1 is connected to the series resonant circuit, the series resonant circuit is connected to the transformer and the rectifier silicon stack 17, and then connected to the load through the output filter circuit 18. The neutral line and the three-phase voltage of the three-phase power supply 2 are respectively connected with the grid fault detection circuit 12, the zero-crossing comparator circuit 5 and the voltage acquisition module 15 through the EMI filter 4; the zero-crossing comparator circuit 5 and the phase detection unit 6 are connected The input terminal is connected, and an output terminal of the phase detection unit 6 is connected with the control parameter calculation unit 9, and is used for looking up a table to obtain the address required by K; the load voltage acquisition circuit 11 is connected with the closed-loop control unit 10, and the closed-loop control unit is based on the set value and The actual output value is compared and calculated to obtain the control quantity and input to the control parameter calculation unit 9, the control parameter calculation unit 9 calculates the time control quantity required for control according to the grid voltage obtained by the grid voltage detection circuit 15 and the K value obtained by looking up the table, And send it to the timing generation unit 8, the timing generation unit 8 generates 12 control signals according to a certain commutation timing and sends them to the switch selection module 7, the switch selection module 7 judges the interval of the power grid according to the phase detection unit 6, and the alternating positive and negative The current output direction selects the switch number corresponding to the 12 signals output by the timing generation unit 8, and the control signal corresponding to the actual switch position output by the switch selection module 7 drives the switch matrix 1 through the switch drive circuit 16 to complete the main circuit of the converter. control. The fault detection circuit includes that the power grid fault detection circuit 12 detects that the power grid is abnormal, the overcurrent detection circuit 13 detects the current abnormality, and the overtemperature detection circuit 14 detects that the temperature is too high, and then sends the fault signal to the switch selection module 7, and the switch selection module 7 After detecting a fault signal, immediately latch the fault and turn off all switches to complete the execution of the protection action.

过零比较器电路5的输入端与电网三相交流2相连，将输入各相电压通过与零线的过零比较转变为与各相电压极性一致的数字信号，此信号经数字滤波之后传输给相位检测单元6和开关状态控制单元7。The input terminal of the zero-crossing comparator circuit 5 is connected to the three-phase AC 2 of the power grid, and the input voltage of each phase is converted into a digital signal with the same polarity as the voltage of each phase through the zero-crossing comparison with the zero line, and the signal is transmitted after digital filtering To the phase detection unit 6 and the switch state control unit 7.

相位检测单元6对电网极性信号进行跟踪和同步，对极性信号宽度进行测量以识别电网是否有故障，根据极性信号的前后变化得到电网相序，同步计数器值与电网的相位对应，根据此值可以间接得到电网的相位，用于辅助处理同一电网极性下的不同区间，并得到不同时刻需要的电荷分析比例K。The phase detection unit 6 tracks and synchronizes the polarity signal of the power grid, measures the width of the polarity signal to identify whether there is a fault in the power grid, and obtains the phase sequence of the power grid according to the change of the polarity signal before and after, and the value of the synchronization counter corresponds to the phase of the power grid. This value can indirectly obtain the phase of the grid, which is used to assist in the processing of different intervals under the same grid polarity, and to obtain the charge analysis ratio K required at different times.

电压采集15模块得到电网各相整流后的实时电压，结合电网极性和需要的激励选择相得到实际的激励电压，并传给计算单元9，计算单元9根据实际的激励电压、等效到变压器初级的负载电压、电荷分配的比例k、谐振周期以及闭环控制给出的控制量计算得到当前状态下的控制所需的时间节点t₁～t₆，最终实现输出稳压电压的同时，电网侧具有高功率因数，低谐波的特点。The voltage acquisition module 15 obtains the real-time voltage after rectification of each phase of the power grid, combines the polarity of the power grid and the required excitation selection phase to obtain the actual excitation voltage, and transmits it to the calculation unit 9, and the calculation unit 9 is equivalent to the transformer according to the actual excitation voltage The primary load voltage, the ratio k of charge distribution, the resonance period and the control quantity given by the closed-loop control are calculated to obtain the time node t ₁ ~ t ₆ required for the control in the current state, and finally realize the output of the regulated voltage. At the same time, the power grid side It has the characteristics of high power factor and low harmonic.

时序生成单元8根据换流的时隙需求以及计算单元给出的各过程关键时间节点(t₁、t₂、t₃、t₄、t₅、t₆)并产生与6种类型开关相对应的时序信号(见附图4)，其中min₁，mid₁，max₁分别为最小相、中间相和最大相且为流通电流的开关信号，min₂，mid₂，max₂分别为最小相、中间相和最大相的辅助开关信号，实际基本不流过电流。The timing generation unit 8 generates the corresponding six types of switches according to the time slot requirements of the commutation and the key time nodes of each process (t ₁ , t ₂ , t ₃ , t ₄ , t ₅ , t ₆ ) given by the calculation unit. timing signal (see Figure 4), where min ₁ , mid ₁ , and max ₁ are the minimum phase, middle phase, and maximum phase, respectively, and are switching signals for current flow, and min ₂ , mid ₂ , and max ₂ are the minimum phase, mid phase, and max 2 respectively. The auxiliary switching signals of the intermediate phase and the maximum phase do not actually flow current.

开关状态控制单元7根据电网极性、相位检测单元6提供的供辅助信号和电流输出方向，选择与不同类型开关相对应的具体开关编号，选定的开关信号受时序生成单元8输出时序的控制，并传给触发驱动电路10加以执行，以电网处于第I区间的开关控制为例，开关信号逻辑选择表达式如下：The switch state control unit 7 selects specific switch numbers corresponding to different types of switches according to the grid polarity, the auxiliary signal provided by the phase detection unit 6 and the current output direction, and the selected switch signal is controlled by the output timing of the timing generation unit 8 , and passed to the trigger drive circuit 10 for execution, taking the switch control of the grid in the I interval as an example, the switch signal logic selection expression is as follows:

触发驱动电路16将触发驱动电路10传送的信号功率放大后，提供门极触发信号给矩阵变换器的各双向功率开关1。The trigger driving circuit 16 amplifies the power of the signal transmitted by the trigger driving circuit 10 , and then provides gate trigger signals to the bidirectional power switches 1 of the matrix converter.

故障保护电路包括三相输入检测电路12，过流保护电路13，过温保护电路14，输出接开关状态控制单元7，有故障时关闭所有开关实现故障保护。三相输入检测电路连接矩阵变换器的输入端，测量三相输入过压，欠压，缺相和不平衡故障。过流保护电路连接串联谐振单元，测量谐振电流，实现过流保护。过温保护电路连接双向功率开关底板和安装变压器与硅堆的油箱，实现过温检测和保护。The fault protection circuit includes a three-phase input detection circuit 12, an overcurrent protection circuit 13, and an overtemperature protection circuit 14. The output is connected to the switch state control unit 7, and all switches are closed to realize fault protection when there is a fault. The three-phase input detection circuit is connected to the input terminal of the matrix converter to measure the three-phase input overvoltage, undervoltage, phase loss and unbalanced faults. The over-current protection circuit is connected with the series resonant unit to measure the resonant current to realize over-current protection. The over-temperature protection circuit is connected to the bottom plate of the bidirectional power switch and the oil tank where the transformer and the silicon stack are installed to realize over-temperature detection and protection.

本发明控制装置中，过零比较器5电路安装在三相输入电源2与输入滤波器3之间，在三相信号进入过零比较器电路之前再经过一个EMI滤波器4，输入相电压波形好，干扰少，过零比较器电路5采用简单易行的常规电路，考虑到EMI滤波器等环节引起的电压相位滞后，相位检测单元6中通过同步修正实现相位的补偿。相位检测单元6，控制参数计算单元9，时序生成单元8和开关状态控制单元7等用现场可编程逻辑门阵列（FPGA）实现，如图6所示。In the control device of the present invention, the zero-crossing comparator 5 circuit is installed between the three-phase input power supply 2 and the input filter 3, and the three-phase signal passes through an EMI filter 4 before entering the zero-crossing comparator circuit, and the input phase voltage waveform Well, there is less interference. The zero-crossing comparator circuit 5 adopts a simple and easy conventional circuit. Considering the voltage phase lag caused by the EMI filter and other links, the phase detection unit 6 realizes phase compensation through synchronous correction. The phase detection unit 6, the control parameter calculation unit 9, the sequence generation unit 8 and the switch state control unit 7 are realized by field programmable logic gate array (FPGA), as shown in Fig. 6 .

Claims

1. a control method of AC-DC series resonant matrix converter, is characterized in that, comprises the following steps:

a. Collect the load voltage V ₀ and the three-phase input phase voltage u _a , u _b , u _c of the three-phase voltage source in real time;

b. According to the relative size relationship of the three-phase input phase voltage u _a , u _b , u _c collected in real time, each input phase voltage cycle is divided into 12 intervals, and the polarity and magnitude of the phase voltage in each interval are determined , and maintain a monotonous change, the 12 intervals are specifically:

Interval Ⅰ: u _a > u _c > u _b , U _P = u _a , U _M = u _c , U _N = u _b ;

Interval II: u _a > u _b > u _c , U _P = u _a , U _M = u _b , U _N = u _c ;

Interval III: u _a > u _b > u _c , U _P = u _c , U _M = u _b , U _N = u _a ;

Interval IV: u _b >u _a >u _c , U _P =u _c , U _M =u _a , U _N =u _b ;

Interval Ⅴ: u _b > u _a > u _c , U _P = u _b , U _M = u _a , U _N = u _c ;

Interval VI: u _b >u _c >u _a , U _P =u _b , U _M =u _c , U _N =u _a ;

Interval VII: u _b > u _c > u _a , U _P = u _a , U _M = u _b , U _N = u _c ;

Interval Ⅷ: u _c > u _b > u _a , U _P = u _a , U _M = u _b , U _N = u _c ;

Interval IX: u _c > u _b > u _a , U _P = u _c , U _M = u _b , U _N = u _a ;

Interval X: u _c > u _a > u _b , U _P = u _c , U _M = u _a , U _N = u _b ;

Interval Ⅺ: u _c > u _a > u _b , U _P = u _b , U _M = u _a , U _N = u _c ;

Interval Ⅻ: u _a > u _c > u _b , U _P = u _b , U _M = u _c , U _N = u _a ;

Among them, the amplitude of U _p is the largest, and the amplitude of U _M is the smallest; define the high line voltage U _j =| _UP -U _N |, the low line voltage U _k =| _UP -U _M |;

d. According to the peak value uc _max of the resonant capacitor voltage and the load voltage V ₀ , obtain the time required for each of the sequentially connected low-line voltage U _k , high-line voltage U _j and 0 voltage and the switching time of the three voltages point;

e. According to the interval of the phase voltage of the power grid at the working time and the current direction to be output, and according to the working process described in step c, assign the corresponding switching state combination of the power switch;

f. According to the switching time points of the three voltages obtained in step d, a general sequence control signal is generated to control the switching between the various working processes;

g. Complete the selection and switching of the three phases according to the control of step f, judge whether the work is over, if yes, exit, if not, return to step a.

2. the control method of a kind of AC-DC series resonant matrix converter according to claim 1, is characterized in that, the concrete method of step d is:

According to the working characteristics of the series resonant converter, a rectangular coordinate system is constructed with the resonant capacitor voltage as the horizontal axis and the product value of the resonant current i and the characteristic impedance Z as the vertical axis, and the characteristic impedance of the resonant circuit

\{\begin{matrix} {R R}_{22}^{22} - - {(({O o}_{22} - - {u u}_{11}))}^{22} + + {(({O o}_{11} - - {u u}_{11}))}^{22} = = {R R}_{11}^{22} \\ {R R}_{22}^{22} - - {(({u u}_{22} - - {O o}_{22}))}^{22} + + {(({u u}_{22} - - {O o}_{33}))}^{22} = = {R R}_{33}^{22} \\ {uc uc}_{max max} = = {R R}_{11} - - {O o}_{11} \\ {R R}_{33} = = {uc uc}_{max max} - - {O o}_{33} = = {R R}_{11} - - {O o}_{11} - - {O o}_{33} \end{matrix}

The first two are available:

\{\begin{matrix} {(({u u}_{11} - - {O o}_{22}))}^{22} - - {(({u u}_{11} - - {O o}_{11}))}^{22} = = {R R}_{22}^{22} - - {R R}_{11}^{22} \\ {(({u u}_{22} - - {O o}_{33}))}^{22} - - {(({u u}_{22} - - {O o}_{22}))}^{22} = = {(({R R}_{11} - - {O o}_{11} - - {O o}_{33}))}^{22} - - {R R}_{22}^{22} \end{matrix}

After simplification:

{u u}_{11} = = [[\frac{{R R}_{22}^{22} - - {R R}_{11}^{22}}{{O o}_{11} - - {O o}_{22}} + + (({O o}_{11} + + {O o}_{22}))]] / / 22

{u u}_{22} = = [[\frac{{(({R R}_{11} - - {O o}_{11} - - {O o}_{33}))}^{22} - - {R R}_{22}^{22}}{{O o}_{22} - - {O o}_{33}} + + (({O o}_{22} + + {O o}_{33}))]] / / 22

According to the charge distribution constraints:

K K = = \frac{{Q Q}_{11}}{{Q Q}_{22}} = = \frac{CΔ CΔ {uc uc}_{11}}{CΔ CΔ {uc uc}_{22}} = = \frac{{u u}_{11} - - ((- - {uc uc}_{max max}))}{{u u}_{22} - - {u u}_{11}}

Available:

{R R}_{22} = = \sqrt{\frac{K K (({O o}_{11} - - {O o}_{22})) (({uc uc}^{22}_{max max} + + {O o}_{22}^{22} - - 22 {uc uc}_{max max} {O o}_{33})) + + ((11 + + K K)) (({O o}_{22} - - {O o}_{33})) (({uc uc}^{22}_{max max} = = {O o}_{22}^{22} + + {22 uc uc}_{max max} {O o}_{11})) - - 22 {uc uc}_{max max} (({O o}_{11} - - {O o}_{22})) (({O o}_{22} - - {O o}_{33}))}{(({O o}_{22} - - {O o}_{33})) + + K K (({O o}_{11} - - {O o}_{33}))}}

The values of u ₁ and u ₂ can be obtained according to uc _max , O ₁ , O ₂ , O ₃ and K;

Assuming that the radian corresponding to the first working process trajectory is θ ₁ , the corresponding radian of the second working process trajectory is θ ₂ , and the corresponding radian of the third working process trajectory is θ ₃ , the corresponding expressions are respectively:

{θ θ}_{11} = = {cos cos}^{- - 11} ((\frac{{O o}_{11} - - {u u}_{11}}{{R R}_{11}}))

{θ θ}_{22} = = π π - - {cos cos}^{- - 11} (({θ θ}_{22__11})) - - {cos cos}^{- - 11} (({θ θ}_{22__22})) = = π π - - {cos cos}^{- - 11} ((\frac{{u u}_{22} - - {O o}_{22}}{{R R}_{22}})) - - {cos cos}^{- - 11} ((\frac{{O o}_{22} - - {u u}_{11}}{{R R}_{22}}))

{θ θ}_{33} = = {cos cos}^{- - 11} ((\frac{{u u}_{22} - - {O o}_{33}}{{R R}_{33}}))

According to θ=ωt, it can be obtained: the end time of the first working process t ₁ = θ ₁ /ω, the ending time of the second working process t ₂ =(θ ₁ +θ ₂ )/ω, the ending time of the third working process t ₃ = (θ ₁ +θ ₂ +θ ₃ )/ω; where ω is the resonant angular frequency,

Switching point times t ₁ , t ₂ and t ₃ can be obtained respectively.

3. A control device for an AC-DC series resonant matrix converter, comprising a three-phase power supply (2), a filter (3), an EMI filter (4), a silicon rectifier stack (17) and a filter circuit (18), It is characterized in that it also includes a switch matrix (1), a trigger drive circuit (16), a fault protection circuit, a zero-crossing comparator circuit (5), a phase detection unit (6), a voltage acquisition module (15), and a load voltage acquisition circuit (11), a closed-loop control unit (10), a control parameter calculation unit (9), a sequence generation unit (8) and a switch state control unit (7), the fault protection circuit includes a power grid fault detection circuit (12), an overcurrent detection circuit (13) and over-temperature detection circuit (14); the phase voltage of the three-phase power supply (2) is connected to the switch matrix (1) through the filter (3), and the neutral line of the three-phase power supply (2) and The phase voltages are respectively connected to the grid fault detection circuit (12), the zero-crossing comparator circuit (5) and the voltage acquisition module (15) through the EMI filter (4); the zero-crossing comparator circuit (5) is connected to the phase detection unit (6 ), one output terminal of the phase detection unit (6) and the output terminal of the voltage acquisition module (15) are connected to the input terminal of the control parameter calculation unit (9), and the output terminal of the control parameter calculation unit (9) is connected to The input terminal of the sequence generation unit (8) is connected, the output terminal of the sequence generation unit (8), the other output terminal of the phase detection unit (6), and the outputs of the overcurrent detection circuit (13) and the overtemperature detection circuit (14) terminal is connected to the input end of the switch state control unit (7), the output end of the switch state control unit (7) is connected to the input end of the trigger drive circuit (16), and the switch matrix (1) is connected to the output of the trigger drive circuit (16) terminal, resonant circuit excitation terminal and the input terminal of the overcurrent detection circuit (13), the input terminal of the overtemperature detection circuit (14) is inserted into the oil tank equipped with a high frequency transformer and a rectifier silicon stack (17), and the rectifier silicon stack (17) The other end of the filter circuit (18) is connected to the load voltage acquisition circuit (11), the output end of the load voltage acquisition circuit (11) is connected to the input end of the closed-loop controller (10), and the output end of the closed-loop controller (10) is connected to the control The input terminal of the parameter calculation unit (9); where,

The zero-crossing comparator circuit (5) transforms each phase voltage input by the three-phase power supply (2) into a digital signal with the same polarity as each phase voltage through zero-crossing comparison with the zero line, and the digital signal is transmitted to the A phase detection unit (6) and a switch state control unit (7);

The load voltage acquisition circuit (11) collects the actual output voltage to the load part and transmits it to the closed-loop control unit (10), and the closed-loop control unit (10) compares and calculates the control quantity according to the fixed value and the actual output value and inputs it to the control parameter calculation unit(9);

The phase detection unit (6) tracks and synchronizes the polarity signal of the power grid, measures the width of the polarity signal to identify whether there is a fault in the power grid, obtains the phase sequence of the power grid according to the change of the polarity signal, and the synchronization counter value corresponds to the phase of the power grid , according to this value, the phase of the power grid can be indirectly obtained, which is used to assist in the processing of different intervals under the same grid polarity, and to obtain the charge analysis ratio K required at different times. The data obtained by the phase detection unit (6) is transmitted to the control parameter calculation unit (9) and switch state control unit (7);

The control parameter calculation unit (9) obtains the current cycle in the current state and the time required to switch phases according to the received data transmitted by the phase detection unit (6) and the voltage acquisition module (15), and transmits the obtained time control amount To timing generation unit (8);

The timing generation unit (8) generates a timing signal for controlling the switch according to the received signal, and transmits the control signal to the switch state control unit (7);

The switch state control unit (7) judges the interval of the power grid according to the received timing signal and the data provided by the phase detection unit (6), as well as the selection of the positive and negative alternating current output direction and the 12 channels output by the timing generation unit (8). The switch number corresponding to the signal, the control signal corresponding to the actual switch position output by the switch selection module (7) drives the switch matrix (1) through the switch drive circuit (16) to complete the control of the main circuit of the converter.