CN103457549A - Tri-band radio frequency power amplifier and impedance matching method of matching network of tri-band radio frequency power amplifier - Google Patents

Abstract

The invention provides an impedance matching method of a matching network of a tri-band radio frequency power amplifier. Firstly, real part matching is carried out by a cross network, so that all real parts in three frequency bands can be accurately matched; simultaneously, an imaginary part can be generated for a certain frequency, but the imaginary part can be compensated in an imaginary part matching network. In the imaginary part matching, the characteristic of influence of a network connected in parallel with a one-fourth open circuit stub on a corresponding frequency point on admittance can be shielded by the one-fourth open circuit stub; an imaginary part generated by the cross network is combined to carry out matching on the imaginary parts in the three frequency bands one by one, so that the tri-band radio frequency power amplifier is matched with the required optical imaginary part. According to the invention, the three frequency bands are irrelevant and can be randomly selected as required; the design space of the tri-band power amplifier is expanded and design flexibility is improved. The problem that a conventional radio frequency power amplifier is difficult to work in three different frequency bands is solved; optical matching is ensured in each working frequency band; the relative bandwidth pressure of the working frequency bands is reduced; the impedance matching method is beneficial for reducing hardware cost of wireless communication and can be used for a multimode base station and a future wireless communication system.

Description

The impedance matching methods of three band radio frequencies power amplifiers and matching network thereof

Technical field

The present invention relates to the multiband RF power amplifier, relate in particular to the impedance matching methods of a kind of three band radio frequencies power amplifiers and matching network thereof, belong to the technology for radio frequency field.

Background technology

Now, the sharp increase of wireless communication data traffic carrying capacity, the demand of high speed data transfer constantly enlarges, but the frequency spectrum resource growing tension.In order to solve this contradiction, utilize fully frequency spectrum resource, mobile device industrial organization starts to introduce the technology of carrier aggregation.Carrier aggregation is by a plurality of carrier wave parallel transmission up-downgoing data, thereby the frequency spectrum pressure of a high speed data transfer is dispersed on a plurality of frequency ranges, has improved to a great extent dispatching flexibility and the availability of frequency spectrum.The standard that the carrier aggregation of take is basic modern wireless mobile communications, make the power amplifier of multiband obtain the extensive concern from academia and industrial quarters.

For the power amplifier that need to meet the communication band of a plurality of apart from each others simultaneously, be difficult to guarantee that power amplifier can both finely mate in whole bandwidth, and there is higher delivery efficiency.

Multiband power amplifier can finely address this problem, and it both can guarantee, in each working band, optimum Match is arranged, and had reduced again the relative bandwidth pressure of working frequency range, was conducive to again reduce the hardware cost of radio communication.

Multiband power amplifier need to be operated in a plurality of different frequency bands, and, in each frequency band, radio-frequency power amplifier all needs a plural load impedance to make performance reach optimum.How the load resistance of a determined value can be converted to required separately complex impedance by a matching network in a plurality of frequency bands becomes a key problem, also there is no at present especially effectively technological means.

Summary of the invention

One of purpose of the present invention, be to provide a kind of impedance matching methods of three band radio frequencies power amplifier matching networks, and its technical scheme is:

A kind of impedance matching methods of three band radio frequencies power amplifier matching networks, described radio-frequency power amplifier is comprised of power amplifier tube and input matching network thereof, output matching network, described input matching network makes power amplifier tube that suitable voltage standing wave ratio be arranged in each frequency band, described output matching network carries out power match to power amplifier, makes it in each frequency band, can obtain peak power output; The impedance of described input matching network, output matching network is complex impedance, all there is a real part and an imaginary part, the matching process of this complex impedance is: by input matching network, output matching network parallel connection, only mate the real part of input matching network complex impedance, do not mate imaginary part; The imaginary part of only mating the output matching network complex impedance, do not mate real part, and step is:

(1) with the real part of " ten " font net mate input matching network complex impedance, in matching process, can produce an imaginary part in a certain frequency (frequency band);

(2) utilize 1/4th open stub can shield and its parallel network characteristic on the impact of admittance on corresponding frequency, mate one by one the imaginary part of output matching network complex impedance at each frequency band, during coupling, the imaginary part that step (1) is produced compensates.

The detailed step of described step (1) is:

If when power tube is carried out to the load traction, in frequency f ₁, f ₂, f ₃the admittance that place needs is respectively G (f ₁)+jB (f ₁), G (f ₂)+jB (f ₂), G (f ₃)+jB (f ₃), the admittance of namely locating on the LL' plane in output matching network figure, the admittance of eyeing right on JJ ' plane is at f ₁, f ₂, f ₃the admittance at place is respectively G (f ₁), G (f ₂)+jQ, G (f ₃), quarter-wave transmission line is expressed as by form of admittance:

its ABCD transmission matrix is:

$\left[\begin{array}{cc}{A}_0& B_0\\ C_0& {\mathrm{<figure-callout\; id="0"\; label="D"\; filenames="HDA0000381410820000013.png"\; state="\{\{state\}\}">D</figure-callout>}}_0\end{array}\right]=\left[\begin{array}{cc}0& \&PlusMinus;j{Z}_T\\ \&PlusMinus;j\frac{1}{Z_T}& 0\end{array}\right]---\left(1\right)$

Substitute quarter-wave transmission line with " ten " font network, make the cross network at f ₁and f ₃the ABCD transmission matrix identical with quarter-wave transmission line inside arranged;

If the stub of quarter-wave transmission line 1 and stub 2 are opened a way, the ABCD transmission matrix of " ten " font network can obtain into:

$\left[\begin{array}{cc}A& B\\ C& D\end{array}\right]=\left[\begin{array}{cc}\cos {2\mathrm{\&theta;}}_s-\frac{1}{2}(\frac{{\tan \mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{{\tan \mathrm{\&theta;}}_{b2}}{Z_{b2}})Z_s\sin {2\mathrm{\&theta;}}_s& j{Z}_s[\sin {2\mathrm{\&theta;}}_s-(\frac{{\tan \mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{{\tan \mathrm{\&theta;}}_{b2}}{Z_{b2}})Z_s{\sin }^2{\mathrm{\&theta;}}_s]\\ j_{}\frac{1}{Z_s}[\sin {2\mathrm{\&theta;}}_s+(\frac{{\tan \mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{{\tan \mathrm{\&theta;}}_{b2}}{Z_{b2}})Z_s{\cos }^2{\mathrm{\&theta;}}_s]& \cos {2\mathrm{\&theta;}}_s-\frac{1}{2}(\frac{{\tan \mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{{\tan \mathrm{\&theta;}}_{b2}}{Z_{b2}})Z_s\sin 2{\mathrm{\&theta;}}_s\end{array}\right]---\left(2\right)$

Allow the transmission matrix of " ten " font network at f ₁and f ₃place equals the transmission matrix of quarter-wave transmission line;

Make A=D=0 obtain:

$\frac{1}{Z_s}\left(\frac{{\cos }^2\mathrm{\&theta;}-{\sin }^2\mathrm{\&theta;}}{\sin \mathrm{\&theta;}\cos \mathrm{\&theta;}}\right)=\frac{\tan {\mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{\tan {\mathrm{\&theta;}}_{b2}}{Z_{b2}}---\left(3\right)$

To in the B of (3) substitution transmission matrix and C, can obtain:

Z _T=Z _stanθ _s (4)

" ten " font network is at f ₁and f ₃there is characteristic impedance at place:

Z _T1=Z _stanθ _s (5)

Z _T3=Z _stan(mθ _s) (6)

Wherein, m=f ₃/ f ₁, the admittance of eyeing right on JJ ' plane is at f ₁, f ₃the admittance at place is respectively G (f ₁), G (f ₃), have:

$Z_{T1}^2=\frac{1}{G_{L}G\left(f_1\right)}---\left(7\right)$

$Z_{T3}^2=\frac{1}{G_{L}G\left(f_3\right)}---\left(8\right)$

Simultaneous (5), (6), (7), (8) formula can solve Z _sand θ _s;

Formula (3) is in frequency f ₁and f ₃place sets up, so have:

$\frac{{Z_{s_2}}^2-{Z_{T_2}}^2}{{Z_{s_2}}^2{Z}_{T_2}}=\frac{{\tan \mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{{\text{tan\&theta;}}_{b2}}{Z_{b2}}---\left(9\right)$

$\frac{{Z_{s_3}}^2-{Z_{T_3}}^2}{{Z_{s_3}}^2{Z}_{T_3}}=\frac{{\tan \mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{{\text{tan\&theta;}}_{b2}}{Z_{b2}}---\left(10\right)$

And for frequency f ₂, on JJ ' plane to the right be transmission matrix:

$\left[\begin{array}{cc}{A}_{{\mathrm{Jf}}_2}& B_{{\mathrm{Jf}}_2}\\ C_{{\mathrm{Jf}}_2}& C_{{\mathrm{Jf}}_2}\end{array}\right]=\left[\begin{array}{cc}A& B\\ C& D\end{array}\right]\left[\begin{array}{cc}1& 0\\ G_L& 1\end{array}\right]=\left[\begin{array}{cc}A+{\mathrm{BG}}_L& B\\ C+{\mathrm{DG}}_L& D\end{array}\right]---\left(11\right)$

So:

$Y_J\left(f_2\right)=\frac{C_{{\mathrm{Jf}}_2}}{A_{{\mathrm{Jf}}_2}}=\frac{C+{\mathrm{DG}}_L}{A+{\mathrm{BG}}_L}=P+\mathrm{jQ}---\left(12\right)$

Because Y _j(f ₂)=G (f ₂)+jQ.Make P=G (f ₂), simultaneous formula (9) and (10), can obtain θ _b1, Z _b1, θ _b2, Z _b2; And then can determine the value of Q;

The detailed step of described step (2) is:

In frequency f ₁the admittance of the B of place part is jB (f ₁), in frequency f ₂the admittance that goes out the B part is jB (f ₂)-jQ, in frequency f ₃admittance jB (the f of the B of place part ₃), in the situation that load short circuits, design microstrip line a makes it in frequency f ₃the admittance at place is jB (f ₃), design microstrip line b is in frequency f ₃place is the quarter-wave open stub, and K point place is for frequency f like this ₃for short circuit, and the network on M point the right is in frequency f ₃place is on not impact of admittance;

mean that microstrip line m is in frequency f _nunder characteristic impedance,

mean that microstrip line m is in frequency f _nunder electrical length, Y _a(f ₂) and Y _b(f ₂) can calculate according to the formula of impedance of microstrip line:

$Y_A\left(f_2\right)=\frac{j}{Z_{a{f}_2}}\left[\frac{Z_{a{f}_2}(B\left(f_2\right)-Q)-\tan {\mathrm{\&theta;}}_{a{f}_2}}{1+Z_{a{f}_2}(B\left(f_2\right)-Q)\tan {\mathrm{\&theta;}}_{a{f}_2}}\right]---\left(13\right)$

$Y_B\left(f_2\right)=\frac{Z_{b{f}_2}}{j\tan {\mathrm{\&theta;}}_{b{f}_2}}---\left(14\right)$

In the situation that load short circuits, design microstrip line c is in frequency f ₂the admittance at place is Y _a(f ₂)-Y _b(f ₂), design microstrip line d is in frequency f ₂place's quarter-wave open stub, N point place is for frequency f like this ₂for short circuit, the network on L point the right is in frequency f ₂place is on not impact of admittance, for frequency f ₁, according to the impedance transformation theory of microstrip line, have following formula to set up:

$Y_A\left(f_2\right)=\frac{j}{Z_{a{f}_1}}\left[\frac{Z_{a{f}_1}B\left(f_1\right)-\tan {\mathrm{\&theta;}}_{a{f}_1}}{1+Z_{a{f}_1}B\left(f_1\right)\tan {\mathrm{\&theta;}}_{a{f}_1}}\right]---\left(15\right)$

$Y_B\left(f_1\right)=\frac{Z_{b{f}_1}}{j\tan {\mathrm{\&theta;}}_{b{f}_1}}---\left(16\right)$

Y _C(f ₁)=Y _A(f ₁)-Y _B(f ₁) (17)

$Y_D\left(f_2\right)=\frac{j}{Z_{c{f}_1}}\left[\frac{Z_{c{f}_1}{Y}_C\left(f_1\right)-\tan {\mathrm{\&theta;}}_{c{f}_1}}{1+Z_{c{f}_1}{Y}_C\left(f_1\right)\tan {\mathrm{\&theta;}}_{c{f}_1}}\right]---\left(18\right)$

$Y_E\left(f_1\right)=\frac{Z_{{\mathrm{df}}_1}}{j\tan {\mathrm{\&theta;}}_{{\mathrm{df}}_1}}---\left(19\right)$

Y _F(f ₁)=Y _D(f ₁)-Y _E(f ₁) (20)

Design microstrip line e, make it in the situation that load open circuit, in frequency f ₁admittance be Y _e(f ₁).

Two of purpose of the present invention, be to provide a kind of three band radio frequencies power amplifiers, and its technical scheme is:

A kind of three band power amplifiers is characterized in that said method develops.

Beneficial effect of the present invention:

The present invention is mated respectively by real part and the imaginary part of the input impedance to required and load impedance, makes power amplifier in three frequency bands can reach highest-gain and peak power output.Solve the conventional radio frequency power amplifier and be difficult to be operated in the problem in three different frequency bands; guarantee has optimum Match in each working band; reduce the relative bandwidth pressure of working frequency range, be conducive to reduce the hardware cost of radio communication, can be used for Multi-Mode Base Station and future broadband wireless communication systems.

The accompanying drawing explanation

The detailed description of the most preferred embodiment by the reference accompanying drawing, above-mentioned and further feature of the present invention and advantage will become obvious, wherein:

Fig. 1 tri-band power amplifier architecture figure

Fig. 2 tri-frequency band matching network structure charts

Fig. 3 tri-frequency band real part coupling schematic diagrames

Fig. 4 tri-frequency band imaginary part coupling schematic diagrames

Embodiment

Hereinafter, with reference to the accompanying drawing of the embodiment of the present invention, describe technical scheme of the present invention in detail, make it more clear, complete, clear.Described herein is a part of embodiment of the present invention, rather than whole embodiment.

Fig. 1 illustrates the structure of this three band power amplifiers, and it is comprised of power amplifier tube and input matching network thereof, output matching network.

Shown in Fig. 2 is the output matching network schematic diagram of three band radio frequencies power amplifiers.The key property of this matching network, be in three frequency bands, all the load impedance of 50 ohm can be converted to needed plural load.Matching network can be divided into two parts, and A partly is responsible for the coupling of real part, and B partly is responsible for the coupling of imaginary part.Suppose power tube is carried out to the load traction, at f ₁, f ₂, f ₃the admittance that place needs is respectively G (f ₁)+jB (f ₁), G (f ₂)+jB (f ₂), G (f ₃)+jB (f ₃), the admittance of namely locating on the LL' plane.

In the A part, we are only mated the real part of impedance, but also can introduce imaginary part, and this can be revised during the imaginary part coupling.For simplified design, the way that we adopt is at f in the admittance that need to eye right on JJ ' plane due to us ₁, f ₂, f ₃the admittance at place is respectively G (f ₁), G (f ₂)+jQ, G (f ₃).

Quarter-wave transmission line has the effect of impedance transformation, by form of admittance, is expressed as:

its ABCD transmission matrix is:

$\left[\begin{array}{cc}{A}_0& B_0\\ C_0& {\mathrm{<figure-callout\; id="0"\; label="D"\; filenames="HDA0000381410820000013.png"\; state="\{\{state\}\}">D</figure-callout>}}_0\end{array}\right]=\left[\begin{array}{cc}0& \&PlusMinus;{\mathrm{jZ}}_T\\ \&PlusMinus;j\frac{1}{Z_T}& 0\end{array}\right]---\left(1\right)$

We substitute quarter-wave transmission line with " ten " font network, make the cross network at f ₁and f ₃the ABCD transmission matrix identical with quarter-wave transmission line inside arranged, as shown in figure (3).Stub 1 and stub 2, can be that to open a way can be also short circuit, for choosing of open circuit and short circuit, needs the consideration stub whether have physical realizability.

Suppose that stub 1 and stub 2 open a way, the ABCD transmission matrix of " ten " font network can obtain into:

$\left[\begin{array}{cc}A& B\\ C& D\end{array}\right]=\left[\begin{array}{cc}\cos {2\mathrm{\&theta;}}_s-\frac{1}{2}(\frac{\tan {\mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{\tan {\mathrm{\&theta;}}_{b2}}{Z_{b2}})Z_s\sin {2\mathrm{\&theta;}}_s& j{Z}_s[\sin {2\mathrm{\&theta;}}_s-(\frac{\tan {\mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{\tan {\mathrm{\&theta;}}_{b2}}{Z_{b2}}\left)Z_s\mathrm{si}{n}^2{\mathrm{\&theta;}}_s\right]\\ j\frac{1}{Z_s}[\sin {2\mathrm{\&theta;}}_s+(\frac{\tan {\mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{\tan {\mathrm{\&theta;}}_{b2}}{Z_{b2}}\left)Z_{s}c{\mathrm{os}}^2{\mathrm{\&theta;}}_s\right]& \cos {2\mathrm{\&theta;}}_s-\frac{1}{2}(\frac{\tan {\mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{\tan {\mathrm{\&theta;}}_{b2}}{Z_{b2}})Z_s\sin {2{\mathrm{\&theta;}}_s}_{}\end{array}\right]$

$\left(2\right)$

Allow the transmission matrix of " ten " font network at f ₁and f ₃place equals the transmission matrix of quarter-wave transmission line.Make A=D=0 obtain:

$\frac{1}{Z_s}\left(\frac{{\cos }^2\mathrm{\&theta;}-{\sin }^2\mathrm{\&theta;}}{\sin \mathrm{\&theta;}\cos \mathrm{\&theta;}}\right)=\frac{\tan {\mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{\tan {\mathrm{\&theta;}}_{b2}}{Z_{b2}}---\left(3\right)$

To in the B of (3) substitution transmission matrix and C, can obtain:

Z _T=Z _stanθ _s (4)

" ten " font network is at f ₁and f ₃there is characteristic impedance at place:

Z _T1=Z _stanθ _s (5)

Z _T3=Z _stan(mθ _s) (6)

Wherein, m=f ₃/ f ₁.The admittance that need to eye right on JJ ' plane due to us is at f ₁, f ₃the admittance at place is respectively G (f ₁), G (f ₃), have:

$Z_{T1}^2=\frac{1}{G_{L}G\left(f_1\right)}---\left(7\right)$

$Z_{T3}^2=\frac{1}{G_{L}G\left(f_3\right)}---\left(8\right)$

Simultaneous (5), (6), (7), (8) formula can solve Z _sand θ _s.

Formula (3) is in frequency f ₁and f ₃place sets up, so have:

$\frac{{Z_{s_2}}^2-{Z_{T_2}}^2}{{Z_{s_2}}^2{Z}_{T_2}}=\frac{\tan {\mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{\tan {\mathrm{\&theta;}}_{b2}}{Z_{b2}}---\left(9\right)$

$\frac{{Z_{s_3}}^2-{Z_{T_3}}^2}{{Z_{s_3}}^2{Z}_{T_3}}=\frac{\tan {\mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{\tan {\mathrm{\&theta;}}_{b2}}{Z_{b2}}---\left(10\right)$

And for frequency f ₂, on JJ ' plane to the right be transmission matrix:

$\left[\begin{array}{cc}{A}_{J{f}_2}& B_{J{f}_2}\\ C_{J{f}_2}& C_{J{f}_2}\end{array}\right]=\left[\begin{array}{cc}A& B\\ C& D\end{array}\right]\left[\begin{array}{cc}1& 0\\ G_L& 1\end{array}\right]=\left[\begin{array}{cc}A+{\mathrm{BG}}_L& B\\ C+{\mathrm{DG}}_L& D\end{array}\right]---\left(11\right)$

So:

$Y_J\left(f_2\right)=\frac{C_{J{f}_2}}{A_{{\mathrm{Jf}}_2}}=\frac{C+{\mathrm{DG}}_L}{A+{\mathrm{BG}}_L}=P+\mathrm{jQ}---\left(12\right)$

Because Y _j(f ₂)=G (f ₂)+jQ.Make P=G (f ₂), simultaneous formula (9) and (10), can obtain θ _b1, Z _b1, θ _b2, Z _b2.And then can determine the value of Q.

B is mated the imaginary part of impedance, will compensate " ten " font network at f simultaneously ₂the imaginary part that place produces.As shown in Figure 2, in frequency f ₁the admittance of the B of place part is jB (f ₁), in frequency f ₂the admittance that goes out the B part is jB (f ₂)-jQ, in frequency f ₃admittance jB (the f of the B of place part ₃).As shown in Figure 4, in the situation that load short circuits, design microstrip line a makes it in frequency f ₃the admittance at place is jB (f ₃), design microstrip line b is in frequency f ₃place is the quarter-wave open stub, and K point place is for frequency f like this ₃for short circuit, and the network on M point the right is in frequency f ₃place is on not impact of admittance.

mean that microstrip line m is in frequency f _nunder characteristic impedance, mean that microstrip line m is in frequency f _nunder electrical length.As shown in Figure 4, Y _a(f ₂) and Y _b(f ₂) can calculate according to the formula of impedance of microstrip line:

$Y_A\left(f_2\right)=\frac{j}{Z_{{\mathrm{af}}_2}}\left[\frac{Z_{{\mathrm{af}}_2}(B\left(f_2\right)-Q)-\tan {\mathrm{\&theta;}}_{{\mathrm{af}}_2}}{1+Z_{{\mathrm{af}}_2}(B\left(f_2\right)-Q)\tan {\mathrm{\&theta;}}_{{\mathrm{af}}_2}}\right]---\left(13\right)$

$Y_B\left(f_2\right)=\frac{Z_{{\mathrm{bf}}_2}}{j\tan {\mathrm{\&theta;}}_{{\mathrm{bf}}_2}}---\left(14\right)$

In the situation that load short circuits, design microstrip line c is in frequency f ₂the admittance at place is Y _a(f ₂)-Y _b(f ₂), design microstrip line d is in frequency f ₂place's quarter-wave open stub, N point place is for frequency f like this ₂for short circuit, the network on L point the right is in frequency f ₂place is on not impact of admittance.For frequency f ₁, according to the impedance transformation theory of microstrip line, have following formula to set up:

$Y_A\left(f_2\right)=\frac{j}{Z_{{\mathrm{af}}_1}}\left[\frac{Z_{{\mathrm{af}}_1}B\left(f_1\right)-\tan {\mathrm{\&theta;}}_{{\mathrm{af}}_1}}{1+Z_{{\mathrm{af}}_1}B\left(f_1\right)\tan {\mathrm{\&theta;}}_{{\mathrm{af}}_1}}\right]---\left(15\right)$

$Y_B\left(f_1\right)=\frac{Z_{{\mathrm{bf}}_1}}{j\tan {\mathrm{\&theta;}}_{{\mathrm{bf}}_1}}---\left(16\right)$

Y _C(f ₁)=Y _A(f ₁)-Y _B(f ₁) (17)

$Y_D\left(f_2\right)=\frac{j}{Z_{{\mathrm{cf}}_1}}\left[\frac{Z_{{\mathrm{cf}}_1}{Y}_C\left(f_1\right)-{\tan \mathrm{\&theta;}}_{{\mathrm{cf}}_1}}{1+Z_{{\mathrm{cf}}_1}{Y}_C\left(f_1\right){\tan \mathrm{\&theta;}}_{{\mathrm{cf}}_1}}\right]---\left(18\right)$

$Y_E\left(f_1\right)=\frac{Z_{{\mathrm{df}}_1}}{j\tan {\mathrm{\&theta;}}_{{\mathrm{df}}_1}}---\left(19\right)$

Y _F(f ₁)=Y _D(f ₁)-Y _E(f ₁) (20)

Design microstrip line e, make it in the situation that load open circuit, in frequency f ₁admittance be Y _e(f ₁).

The three frequency band matching networks that design like this, at f ₁, f ₂, f ₃place's admittance is respectively G (f ₁)+jB (f ₁), G (f ₂)+jB (f ₂), G (f ₃)+jB (f ₃), can be that power amplifier can obtain peak power output at three frequency bands.

Show particularly and described the present invention with reference to embodiment, for one of ordinary skill in the art, the thought according to the embodiment of the present invention, all will change in specific embodiments and applications, in sum, this description should not be construed as limitation of the present invention.

Claims (4)

1. the impedance matching methods of a band radio frequencies power amplifier matching network, described radio-frequency power amplifier is comprised of power amplifier tube and input matching network thereof, output matching network, described input matching network makes power amplifier tube that suitable voltage standing wave ratio be arranged in each frequency band, described output matching network carries out power match to power amplifier, makes it in each frequency band, can obtain peak power output; The impedance of described input matching network, output matching network is complex impedance, all there is a real part and an imaginary part, the matching process of this complex impedance is: by input matching network, output matching network parallel connection, only mate the real part of input matching network complex impedance, do not mate imaginary part; The imaginary part of only mating the output matching network complex impedance, do not mate real part, and step is:

(1) with the real part of " ten " font net mate input matching network complex impedance, in matching process, can produce an imaginary part in a certain frequency;

(2) utilize 1/4th open stub can shield and its parallel network characteristic on the impact of admittance on corresponding frequency, mate one by one the imaginary part of output matching network complex impedance at each frequency band, during coupling, the imaginary part that step (1) is produced compensates.

2. the impedance matching methods of multiband RF power amplifier matching network as claimed in claim 1, is characterized in that, the detailed step of described step (1) is:

If when power tube is carried out to the load traction, in frequency f ₁, f ₂, f ₃the admittance that place needs is respectively G (f ₁)+jB (f ₁), G (f ₂)+jB (f ₂), G (f ₃)+jB (f ₃), the admittance of namely locating on LL ' plane in output matching network figure, the admittance of eyeing right on JJ ' plane is at f ₁, f ₂, f ₃the admittance at place is respectively G (f ₁), G (f ₂)+jQ, G (f ₃), quarter-wave transmission line is expressed as by form of admittance:

its ABCD transmission matrix is:

$\left[\begin{array}{cc}{A}_0& B_0\\ C_0& D_0\end{array}\right]=\left[\begin{array}{cc}0& \&PlusMinus;{\mathrm{jZ}}_T\\ \&PlusMinus;j\frac{1}{Z_T}& 0\end{array}\right]---\left(1\right)$

Substitute quarter-wave transmission line with " ten " font network, make the cross network at f ₁and f ₃the ABCD transmission matrix identical with quarter-wave transmission line inside arranged;

If the stub of quarter-wave transmission line 1 and stub 2 are opened a way, the ABCD transmission matrix of " ten " font network can obtain into:

$\left[\begin{array}{cc}A& B\\ C& D\end{array}\right]=\left[\begin{array}{cc}\cos 2{\mathrm{\&theta;}}_s-\frac{1}{2}(\frac{\tan {\mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{\tan {\mathrm{\&theta;}}_{b2}}{Z_{b2}})Z_s\sin 2{\mathrm{\&theta;}}_s& j{Z}_s[\sin 2{\mathrm{\&theta;}}_s-(\frac{\tan {\mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{\tan {\mathrm{\&theta;}}_{b2}}{Z_{b2}})Z_s{\sin }^2{\mathrm{\&theta;}}_s]\\ j\frac{1}{Z_s}[\sin 2{\mathrm{\&theta;}}_s+(\frac{\tan {\mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{\tan {\mathrm{\&theta;}}_{b2}}{Z_{b2}})Z_s{\cos }^2{\mathrm{\&theta;}}_s]& \cos 2{\mathrm{\&theta;}}_s-\frac{1}{2}(\frac{\tan {\mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{\tan {\mathrm{\&theta;}}_{b2}}{Z_{b2}})Z_s\sin 2{\mathrm{\&theta;}}_s\end{array}\right]$

$\left(2\right)$

Allow the transmission matrix of " ten " font network at f ₁and f ₃place equals the transmission matrix of quarter-wave transmission line;

Make A=D=0 obtain:

$\frac{1}{Z_s}\left(\frac{{\cos }^2\mathrm{\&theta;}-{\sin }^2\mathrm{\&theta;}}{\sin \mathrm{\&theta;}\cos \mathrm{\&theta;}}\right)=\frac{\tan {\mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{\tan {\mathrm{\&theta;}}_{b2}}{Z_{b2}}---\left(3\right)$

To in the B of (3) substitution transmission matrix and C, can obtain:

Z _T＝Z _stanθ _s (4)

" ten " font network is at f ₁and f ₃there is characteristic impedance at place:

Z _T1＝Z _stanθ _s (5)

Z _T3＝Z _stan(mθ _s) (6)

Wherein, m=f ₃/ f ₁, the admittance of eyeing right on JJ ' plane is at f ₁, f ₃the admittance at place is respectively G (f ₁), G (f ₃), have:

$Z_{T1}^2=\frac{1}{G_{L}G\left(f_1\right)}---\left(7\right)$

$Z_{T3}^2=\frac{1}{G_{L}G\left(f_3\right)}---\left(8\right)$

Simultaneous (5), (6), (7), (8) formula can solve Z _sand θ _s;

Formula (3) is in frequency f ₁and f ₃place sets up, so have:

$\frac{{Z_{s_2}}^2-{Z_{T_2}}^2}{{Z_{s_2}}^2{Z}_{T_2}}=\frac{\tan {\mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{\tan {\mathrm{\&theta;}}_{b2}}{Z_{b2}}---\left(9\right)$

$\frac{{Z_{s_3}}^2-{Z_{T_3}}^2}{{Z_{s_3}}^2{Z}_{T_3}}=\frac{\tan {\mathrm{\&theta;}}_{b1}}{Z_{b1}}+\frac{\tan {\mathrm{\&theta;}}_{b2}}{Z_{b2}}---\left(10\right)$

And for frequency f ₂, on JJ ' plane to the right be transmission matrix:

$\left[\begin{array}{cc}{A}_{{\mathrm{Jf}}_2}& B_{{\mathrm{Jf}}_2}\\ C_{{\mathrm{Jf}}_2}& C_{{\mathrm{Jf}}_2}\end{array}\right]=\left[\begin{array}{cc}A& B\\ C& D\end{array}\right]\left[\begin{array}{cc}1& 0\\ G_L& 1\end{array}\right]=\left[\begin{array}{cc}A+B{G}_L& B\\ C+D{G}_L& D\end{array}\right]---\left(11\right)$

So:

$Y_J\left(f_2\right)=\frac{C_{{\mathrm{Jf}}_2}}{A_{{\mathrm{Jf}}_2}}=\frac{C+{\mathrm{DG}}_L}{A+{\mathrm{BG}}_L}=P+\mathrm{jQ}---\left(12\right)$

Because Y _j(f ₂)=G (f ₂)+jQ.Make P=G (f ₂), simultaneous formula (9) and (10), can obtain θ _b1, Z _b1, θ _b2, Z _b2; And then can determine the value of Q.

3. the impedance matching methods of multiband RF power amplifier matching network as claimed in claim 1, is characterized in that, the detailed step of described step (2) is:

In frequency f ₁the admittance of the B of place part is jB (f ₁), in frequency f ₂the admittance that goes out the B part is jB (f ₂)-jQ, in frequency f ₃admittance jB (the f of the B of place part ₃), in the situation that load short circuits, design microstrip line a makes it in frequency f ₃the admittance at place is jB (f ₃), design microstrip line b is in frequency f ₃place is the quarter-wave open stub, and K point place is for frequency f like this ₃for short circuit, and the network on M point the right is in frequency f ₃place is on not impact of admittance; mean that microstrip line m is in frequency f _nunder characteristic impedance,

mean that microstrip line m is in frequency f _nunder electrical length, Y _a(f ₂) and Y _b(f ₂) can calculate according to the formula of impedance of microstrip line:

$Y_A\left(f_2\right)=\frac{j}{Z_{{\mathrm{af}}_2}}\left[\frac{Z_{{\mathrm{af}}_2}(B\left(f_2\right)-Q)-\tan {\mathrm{\&theta;}}_{{\mathrm{af}}_2}}{1+Z_{{\mathrm{af}}_2}(B\left(f_2\right)-Q)\tan {\mathrm{\&theta;}}_{{\mathrm{af}}_2}}\right]---\left(13\right)$

$Y_B\left(f_2\right)=\frac{Z_{{\mathrm{bf}}_2}}{j\tan {\mathrm{\&theta;}}_{{\mathrm{bf}}_2}}---\left(14\right)$

In the situation that load short circuits, design microstrip line c is in frequency f ₂the admittance at place is Y _a(f ₂)-Y _b(f ₂), design microstrip line d is in frequency f ₂place's quarter-wave open stub, N point place is for frequency f like this ₂for short circuit, the network on L point the right is in frequency f ₂place is on not impact of admittance, for frequency f ₁, according to the impedance transformation theory of microstrip line, have following formula to set up:

$Y_A\left(f_2\right)=\frac{j}{Z_{{\mathrm{af}}_1}}\left[\frac{Z_{{\mathrm{af}}_1}B\left(f_1\right)-\tan {\mathrm{\&theta;}}_{{\mathrm{af}}_1}}{1+Z_{{\mathrm{af}}_1}B\left(f_1\right)\tan {\mathrm{\&theta;}}_{{\mathrm{af}}_1}}\right]---\left(15\right)$

$Y_B\left(f_1\right)=\frac{Z_{{\mathrm{bf}}_1}}{j\tan {\mathrm{\&theta;}}_{{\mathrm{bf}}_1}}---\left(16\right)$

Y _C(f ₁)＝Y _A(f ₁)-Y _B(f ₁) (17)

$Y_D\left(f_2\right)=\frac{j}{Z_{{\mathrm{cf}}_1}}\left[\frac{Z_{{\mathrm{cf}}_1}{Y}_C\left(f_1\right)-\tan {\mathrm{\&theta;}}_{{\mathrm{cf}}_1}}{1+Z_{{\mathrm{cf}}_1}{Y}_C\left(f_1\right)\tan {\mathrm{\&theta;}}_{{\mathrm{cf}}_1}}\right]---\left(18\right)$

$Y_E\left(f_1\right)=\frac{Z_{{\mathrm{df}}_1}}{j\tan {\mathrm{\&theta;}}_{{\mathrm{df}}_1}}---\left(19\right)$

Y _F(f ₁)＝Y _D(f ₁)-Y _E(f ₁) (20)

Design microstrip line e, make it in the situation that load open circuit, in frequency f ₁admittance be Y _e(f ₁).

4. a band radio frequencies power amplifier, is characterized in that, by the method for claim 1-3, develops.

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