CN102882670A - Synchronous processing method based on CMMB signals - Google Patents

Synchronous processing method based on CMMB signals Download PDF

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CN102882670A
CN102882670A CN2012103374125A CN201210337412A CN102882670A CN 102882670 A CN102882670 A CN 102882670A CN 2012103374125 A CN2012103374125 A CN 2012103374125A CN 201210337412 A CN201210337412 A CN 201210337412A CN 102882670 A CN102882670 A CN 102882670A
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sequence
frequency offset
cmmb
multiple
estimation
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CN2012103374125A
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雍芝奎
李多烨
甘露
黄磊
廖红舒
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电子科技大学
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Abstract

The invention provides a synchronous processing method based on new CMMB (China Mobile Multimedia Broadcasting) synchronous signals. The CMMB synchronous signals comprise first training sequences and second training sequences; the first training sequences comprise CAZAC (Constant Amplitude Zero Auto Correlation) sequences, and the second training sequences comprise PN (Pseudo-Noise) sequences; in the synchronous processing method, the CAZAC sequences are utilized to achieve coarse symbol timing offset estimation and decimal frequency offset coarse estimation; the PN sequences are utilized to achieve the estimation of strongest path time delay in multipath, and the multipath is taken as coarse symbol timing positioning; the CAZAC sequences and the PN sequences are utilized to achieve the integer frequency offset estimation; the fast Fourier transform (FFT) is performed on the PN sequences to achieve the channel response estimation so as to estimate a first path time delay and achieve the fine symbol timing position estimation; and a maximum likelihood (ML) criterion is utilized to process the PN sequences in the second training sequences to obtain the decimal frequency offset fine estimation. The synchronous processing method can effectively improve the synchronous accuracy and can achieve better synchronization performance in the mobile communication environment with low signal-to-noise ratio.

Description

—种基于CMMB信号的同步处理方法 - synchronization processing method CMMB signals based species

技术领域 FIELD

[0001] 本发明涉及无线通信同步技术,尤其涉及无线通信中的正交频分复用OFDM通信系统。 [0001] The present invention relates to synchronization of wireless communication technologies, and particularly to a wireless communication in the orthogonal frequency division multiplexing OFDM communication system.

技术背景 technical background

[0002] 正交频分复用OFDM是一种高速传输技术,其特点是将单个的高速数据流在一组低速正交子载波上并行传输,其具有频谱利用率高,抗频率选择性衰落,适合高速率数据传输且实现简单的特点。 [0002] Orthogonal Frequency Division Multiplexing (OFDM) technology is a high-speed transmission, which is characterized by a single high-speed data streams on a set of low-speed parallel transmission orthogonal subcarriers, which has a high frequency efficiency, is robust against frequency selective fading , suitable for high-speed data transmission, and simple features. 与单载波系统相比,它对时间和频偏很敏感,因此为了消除符号间干扰和载波间干扰,准确的时间同步和精确的频率同步是OFDM系统正常工作的前提。 Compared with single carrier systems, it is very sensitive to frequency offset and time, and therefore in order to eliminate inter-symbol interference and inter-carrier interference, accurate time synchronization and precise frequency synchronization is provided an OFDM system to work properly. 接收端的频偏按照子载波的间隔进行归一化,就出现了小数倍频偏和整数倍频偏。 Receiving end a frequency offset normalized subcarrier intervals, there have been multiple small integer frequency offset and frequency offset. 对OFDM系统而言,在AWGN信道要求频率偏移值不大于子载波间隔的4% ;对于衰落信道要求频率偏移值不大于子载波间隔的2〜3%,要求符号定时位置处在ISI-FREE区间内。 For an OFDM system, in an AWGN channel requires frequency offset value is not more than 4% of the subcarrier spacing; fading channel requirements for a frequency offset is not greater than 2 to 3% of the subcarrier spacing, symbol timing location in claim ISI- FREE within range.

[0003]目前OFDM系统的同步技术分为两大类:数据辅助(DA)的方法和非数据辅助(NDA)的方法。 [0003] Synchronization of OFDM systems currently divided into two categories: data-aided (DA) and the non-data-aided methods (NDA) method. 数据辅助(DA)的估计方法是利用同步序列(训练序列)的方法,运算复杂度较低,估计范围较大,但会增加系统开销。 Data-aided (DA) estimation method using synchronization sequence (training sequence) method, a low computational complexity, a larger estimation range, but increases system overhead. 非数据辅助(NDA)的方法是利用CMMB (中国移动多媒体广播)信号的本身特性和接收数据的统计特性来完成同步估计,其中包括盲估计、半盲估计,但是其运算复杂度较高,估计范围有限。 Non-data assisted (NDA) and the method of statistical properties per se characteristic data received by the CMMB (China Mobile Multimedia Broadcasting) to complete the synchronization signal estimate, including blind estimation, semi-blind estimation, but higher computational complexity is estimated limited range.

[0004] CMMB信号利用了PN序列构成的同步信号来完成定时和频差同步,估计范围较宽,运算复杂度低,但是其PAR (峰值平均功率比)较高,在低信噪比的情况下同步效果不佳,而且频偏估计性能不好。 [0004] CMMB signals using the PN sequence constituting the synchronizing signal to the timing and frequency difference complete synchronization, estimation range is wide, low computational complexity, but higher PAR (peak to average power ratio), in the case of low SNR synchronous poor results, and estimated frequency offset poor performance.

[0005] 在移动通信信道中,多径效应带来的频率选择性衰落和多普勒扩展带来的时间选择性衰落都对同步提出了一定的要求。 [0005] In a mobile communication channel, frequency selective fading caused by multipath and Doppler spread caused by time selective fading are made synchronous certain requirements. 由于小数倍频偏会导致ICI (载波间干扰),为了保证同步效果,我们需要保证在第一步完成小数倍频偏的估计,并保证一定的精度。 Due to the small multiple of the frequency offset can cause ICI (ICI), in order to ensure synchronization effect, we need to ensure the completion of a small multiple of the frequency offset estimated in the first step, and to ensure a certain degree of accuracy.

[0006] 现有的同步信号设计以及对应的同步算法主要有以下几种: [0006] The conventional design of a synchronization signal and the corresponding synchronization algorithm are the following:

[0007] I、SCA 算法 [0007] I, SCA algorithm

[0008] SCA算法是利用两个训练序列来完成符号定时的估计和频偏估计,其频偏估计性能较好,但是由于SCA算法的平台效应,在定时方面方差很大,定时估计结果易处于 [0008] SCA algorithm is accomplished using the two training sequences and estimating the frequency offset estimation symbol timing, offset estimation performance which is better, but the effect due to the internet SCA algorithm, a large variance in terms of timing, timing estimation results easily in

ISE-FREE区间以外。 ISE-FREE zone outside. 频偏估计范围为-γ4/,γ4/,其中Λ f为CMMB信号的子载波间隔。 Frequency offset estimation range -γ4 /, γ4 /, where Λ f is the subcarrier spacing CMMB signals.

[0009] 2、MorelIi 算法 [0009] 2, MorelIi algorithm

[0010] Morelli算法是利用在一个CMMB符号中传输J个相同的部分,频偏估计精度有 [0010] Morelli transmission algorithm using the same parts in the J CMMB a symbol, frequency offset estimation precision

一定程度的提高,但是其频偏估计范围有限,估计范围为-|δ/·,香4/。 Improved to some extent, but it is estimated frequency offset range is limited, estimated in the range of - | δ / ·, incense 4 /. 为了增大估计范 In order to increase the range of estimates

围,可以增大J值。 Wai, J value can be increased. 但是J的值不能取的太大,因为为了保证估计结果可靠,必须保证估计每一个相同的数据块的长度必须大于信道时延扩展,此外这种算法的定时方差偏大。 But can not take the value of J is too large, since in order to guarantee a reliable estimation result, must ensure that the same length of each of the estimated data block must be greater than the channel delay spread, moreover the timing variance of this algorithm is too large. [0011 ] 3、Ren 算法 [0011] 3, Ren algorithm

[0012] 利用一个CAZAC序列在时域完成定时偏差和频偏的同步,训练符号包含两个相同的CAZAC序列,第二部分采用PN序列加权。 [0012] using a CAZAC sequence in the time domain to complete synchronization training symbol timing offset and frequency offset of the CAZAC sequence comprising two identical, the second part is weighted sequence PN. 由于其利用PN序列的差分互相关来完成定时,所以其在慢衰落的ISI信道中的定时效果很好,此外,其PAR很低,但是在独立衰落信道下,由于差分相关性能的下降会降低估计性能;而且由于加权PN序列的利用导致同步信号结构非线性扭曲,导致小数倍频偏估计性能下降。 Because of its use of the differential cross correlation PN sequence completion timing, the timing of its good effect of ISI in the slow fading channel, in addition, its low PAR, but independent fading channels, since the differential will reduce the associated performance drop estimation performance; and since the weighted PN sequence results in the nonlinear distortion synchronization signal structure, resulting in a small multiple of the frequency offset estimation performance.

[0013] 4、Adegbenga B 算法 [0013] 4, Adegbenga B Algorithm

[0014] 利用一个训练符号,由两个相同的部分构成。 [0014] using a training symbol, composed of two identical parts. 整个同步过程在时域中完成。 The entire synchronization process is completed in the time domain. 利用自相关、带限的互相关并结合基于门限的检测来完成整个定时和频率的同步,运算量较低。 The autocorrelation, cross-correlation and bandlimited binding threshold based synchronization detection is completed, the entire computation timing and a lower frequency. 但是小数倍频偏的估计在低信噪比的情况下,估计效果较差。 However, a small multiple of the frequency offset estimates at low SNR, it estimates less effective. 在衰落信道中,其效果存在一定程度的降低。 In fading channels, there is a certain degree of reduction effect. 本文的算法就是基于这种算法的改进,CMMB信号的同步信号就是Adegbenga B训练符号的一个例子,该CMMB同步信号的结构见附图I。 Paper is based on modified algorithms of this algorithm, CMMB signal synchronized signal is an example Adegbenga B training symbol, the structure of the synchronization signal CMMB see Figure I.

发明内容 SUMMARY

[0015] 本发明所要解决的技术问题是,为保证CMMB信号在低信噪比的衰落信道下定时同步精度和频率同步精度,提供一种基于新CMMB同步信号的同步处理方法。 [0015] The present invention solves the technical problem, to ensure CMMB signals at low signal fading channel accuracy of timing synchronization and frequency synchronization accuracy, there is provided a synchronous processing method CMMB new synchronizing signal.

[0016] 本发明为解决上述技术所米用的技术方案是,一种基于CMMB信号的同步处理方法,利用CMMB同步信号对CMMB信号的小数倍频率偏移、整数倍频率偏移以及CMMB信号的起始时间进行估计从而完成同步处理,所述CMMB同步信号由第一训练序列与第二训练序列组成,所述第一训练序列包括长度为N的恒包络零自相关CAZAC序列以及长度为GI的循环前缀,所述第二训练序列包括长度为N的PN序列以及长度为GI的循环前缀; [0016] aspect of the present invention is to solve the above technology used is m, a synchronous processing method CMMB signals based on the sync signal using the CMMB CMMB signals of the smaller multiple of the frequency shift of an integer multiple of the frequency offset signal and CMMB the estimated time to complete the initial synchronization, the synchronization signal CMMB a first training sequence and the second training sequences, the first training sequence comprises a length N constant amplitude zero autocorrelation sequence and a length of CAZAC GI cyclic prefix, wherein the second training sequence comprises a PN sequence of length N and a cyclic prefix length of the GI;

[0017] 同步处理的具体步骤如下: [0017] DETAILED synchronization processing steps as follows:

[0018] I)利用第一训练序列中CAZAC序列完成粗定时偏移估计以及小数倍频偏粗估计 [0018] I) a first training sequence using CAZAC sequence is completed and a coarse timing offset estimation multiple small frequency offset estimation crude

P · P ·

GF 9 GF 9

[0019] 2)使用小数倍频偏粗估计'对接收序列进行小数倍频偏补偿,利用小数倍频偏补偿后的PN序列完成多径中最强路径时延S的估计,将最强路径时延別乍为粗定时位置; [0019] 2) coarse frequency offset estimation using a small multiple of 'small multiple of the received frequency offset compensation sequence, a PN sequence of a small multiple of the estimated frequency offset compensation is completed strongest path delay multipath S will Chad is not the strongest path delay coarse timing location;

[0020] 3)使用粗定时位置g以及小数倍频偏粗估计4对接收序列进行补偿,利用补偿后的CAZAC序列和PN序列完成整数倍频率偏移之的估计,将整数倍频率偏移4作为CMMB信号的整数倍频率偏移; [0020] 3) using the coarse timing offset positions g and small multiple of four pairs of rough estimation is compensated received sequence, the CAZAC sequence and the PN sequence is completed using the compensated estimate of an integer multiple of a frequency offset of an integer multiple of the frequency offset 4 is an integer multiple of a frequency offset CMMB signals;

[0021] 4)使用整数倍频率偏移毛对补偿后的接收序列再进行整数倍频率补偿,对整数倍频率补偿后的第二训练序列中PN序列进行快速傅里叶变换FFT完成信道响应估计,估计出第一条路径的时延f并完成精定时位置Θ的估计,0 = 0 + τ,将精定时位置Θ作为CMMB信号的起始时间; [0021] 4) the use of an integral multiple of the frequency offset of the received sequence hair compensated again multiplied by an integer frequency compensation, of the second training sequence after the integral multiple of the frequency compensation PN sequence to complete a fast Fourier transform FFT channel response estimate estimate the first path delay f and complete refined timing estimate position Θ, 0 = 0 + τ, the fine timing position Θ CMMB signals as start time;

[0022] 5)使用精定时位置Θ更新整数倍频率补偿后的接收序列,对精定时更新后的第二训练序列中PN序列利用最大似然ML准则获得小数倍频率偏移精估计B ,将小数倍频率偏移精估计έ作为CMMB信号的小数倍频率偏移; [0022] 5) The refined timing position Θ update the compensated received sequence integral multiple of the frequency of the second training sequence PN sequence refined timing update maximum likelihood ML criterion to obtain a small multiple of the frequency offset estimation precision B, the small multiple of the frequency offset estimation precision as CMMB έ small multiple of the frequency offset of the signal;

[0023] 11)利用确定的CMMB信号的小数倍频率偏移、整数倍频率偏移以及CMMB信号的起始时间完成同步处理。 [0023] 11) CMMB signals using a small multiple of the determined frequency offset, an integral multiple of the frequency offset and the start time CMMB signals synchronize.

[0024] 本发明为了减低同步信号的PAR,采用CAZAC序列和模值的PN序列来构成CMMB信号的同步信号来达到较小的PAR,同时为了保证频偏估计效果较好,在后端利用了ML准则在一定的范围内通过搜索获得较好的估计值。 [0024] In order to reduce the PAR of the present invention, the synchronization signal using CAZAC sequences and PN sequences modulus values ​​constitute CMMB signals synchronizing signal to achieve the PAR is small, and in order to ensure better frequency offset estimation, the rear end of the use of ML criteria to obtain better estimates by searching within a certain range.

[0025] 具体的,第一训练序列中CAZAC序列Stl表示为: [0025] Specifically, the first training sequence is a CAZAC sequence represented Stl:

[0026] Stl= [c (n), c (n) ]n=0, I, . . . , (N/2)_l ; [0026] Stl = [c (n), c (n)] n = 0, I,, (N / 2) _l...;

[0027]其中,构成 CAZAC 序列的2 个相同序列c (η)为c (n) =exp (j π η2/ (Ν/2)), exp ( ·)表示以自然对数e为底的指数函数,N为子载波的个数。 [0027] wherein the configuration of the CAZAC sequence two identical sequences c (η) as c (n) = exp (j π η2 / (Ν / 2)), exp (·) represents the natural logarithm of the base e index function, N is the number of subcarriers.

[0028] 具体的,第二训练序列中PN序列St2表示为: [0028] Specifically, the second training sequence is expressed as a PN sequence St2:

[0029] st2 = Ipn(O),-··, pn( N -1)], pn(i) g ·| (I + ϊ),-j=· (I - /), (-1 - /), (-1 τ /) [0029] st2 = Ipn (O), - ··, pn (N -1)], pn (i) g · | (I + ϊ), - j = · (I - /), (-1 - / ), (-1 τ /)

[0030]其中,i = 0,…,Nl。 [0030] where, i = 0, ..., Nl. [0031] 具体的,步骤I)的具体方法为: [0031] Specifically, step I) is a specific method:

[0032] 利用CAZAC序列相同的重复部分的自相关进行峰值平台搜索,将所有大于最大相关值的O. 95倍的定时位置构成的集合称为粗定时范围,取粗定时范围中所有值的平均作为粗定时估计值Θ opt ; Overlapping portions of the same autocorrelation [0032] using a CAZAC sequence a peak search platform, the set of all correlation values ​​greater than the maximum timing position O. 95 times the coarse timing is referred to as range, a coarse timing range taking the average of all the values as the coarse timing estimate Θ opt;

[0033] 利用粗定时估计值Θ opt对应位置的相关值的相位估计出粗小数倍频偏; [0033] The coarse timing estimation using the phase correlation values ​​corresponding to the position value Θ opt estimated coarse frequency offset times smaller;

[0034] 来完成粗符号定时和粗小数倍频偏估计& βΡ = migle、p(eopt)}jπ,其中,P ( ·)为相位函数,angle ( ·)为角度函数。 [0034] The coarse symbol timing to complete multiple small and coarse frequency offset estimation & βΡ = migle, p (eopt)} jπ, where, P (·) is the phase function, angle (·) as a function of angle.

[0035] 具体的,步骤2)中利用小数倍频偏补偿后的PN序列完成多径中最强路径时延§的估计,具体方法为:利用已知的PN同步序列与接收的小数倍频偏补偿后的PN序列完成互相关,搜索互相关结果的峰值出现的位置来获得最强路径时延 [0035] Specifically, step 2) using a small multiple of the PN sequence estimated frequency offset compensation is completed strongest path delay multipath § specific methods of: using known PN sequence synchronization with the received decimal PN sequence after the cross-correlation frequency offset compensation is completed, the position of the cross peak search of correlation results to obtain the strongest path delay

[0036] 具体的,步骤3)中利用补偿后的CAZAC序列和PN序列完成整数倍频率偏移估计,具体方法为: [0036] Specifically, in step 3) CAZAC sequence and the PN sequence by an integer multiple of the frequency offset compensation is completed estimation, the specific method is:

[0037] 利用接收并补偿后的CAZAC序列和PN序列的频域数据与已知的CAZAC序列和PN序列的频域数据进行滑动相关,从而获得整数倍频率偏私ε Frequency domain data and frequency domain data and the known PN sequence CAZAC sequence CAZAC sequence and the PN sequence is [0037] received and compensated using the sliding correlation, thereby obtaining an integer multiple of the frequency favoritism ε

[0038] 具体的,步骤4)中利用整数倍频率补偿后的第二训练序列中PN序列得到精定时位置Θ的具体方法是: [0038] Specifically, in step 4) the position of a specific method refined timing Θ second training sequence after compensation PN sequence utilized frequencies are integer multiples of:

[0039] 对整数倍频率补偿后的第二训练序列中PN序列进行快速傅里叶变换FFT来获得信道响应估计,再经快速傅里叶变换IFFT获得信道时域响应,利用利用信道的能量准则作为判决量估计多径中的第一路径时延f,完成精定时位置Θ的估计,β 二。 [0039] The second training sequence after the integral multiple of the frequency compensation PN sequence FFT fast Fourier transform to obtain a channel response estimate, then by fast Fourier transform IFFT to obtain a time domain channel response, with the energy channel utilization criteria as the judgment amount estimating multipath first path delay f, complete refined timing estimate position Θ, β II.

[0040] 本发明的有益效果是在低信噪比移动通信环境时,能有效提高同步精度,达到更好的同步性能。 [0040] Advantageous effects of the present invention is that at low SNR mobile communication environment, can improve synchronization accuracy, achieve better synchronization performance.

附图说明 BRIEF DESCRIPTION

[0041] 图I为CMMB的PN同步信号结构图; [0041] Figure I is a configuration diagram of a synchronization signal PN of CMMB;

[0042] 图2为本发明CMMB同步信号的结构图; [0042] FIG 2 is a configuration diagram of a synchronization signal CMMB invention;

[0043] 图3为本发明实施例流程图; [0043] FIG. 3 flowchart embodiment of the present invention;

[0044] 图4为本发明CMMB同步信号现有CMMB的PN同步信号的定时估计性能对比图。 FIG timing estimation performance comparison PN synchronization signal [0044] FIG. 4 CMMB CMMB synchronizing signal prior to the present invention.

[0045] 图5为本发明CMMB同步信号与现有CMMB的PN同步信号的频偏估计性能对比图。 [0045] FIG. 5 shows the frequency offset estimation performance comparison chart CMMB PN synchronization signal synchronized with the signal of the existing CMMB invention. 具体实施方式 Detailed ways

[0046] 本发明CMMB同步信号的结构如图2所示,由第一训练序列与第二训练序列组成,所述第一训练序列包括长度为N的恒包络零自相关CAZAC序列以及长度为GI的循环前缀CP,所述第二训练序列包括长度为N的PN序列以及长度为GI的循环前缀CP。 [0046] CMMB synchronization signal structure shown in Figure 2 of the present invention, the first training sequences and the second training sequences, the first training sequence comprises a length N constant amplitude zero autocorrelation sequence and a length of CAZAC GI of the cyclic prefix CP, the second training sequence comprises a PN sequence of length N and a cyclic prefix length of the GI CP. 相同灰度的部分代表其值也相同。 Part of the same gradation representative value are the same.

[0047] 实施例流程如附图3所示,具体实施例的方法包括以下步骤: [0047] Example embodiment of the process as shown in Figure 3, the specific embodiment comprises the method steps of:

[0048] I.利用第一训练序列中CAZAC序列部分的性质完成粗定时估计以及小数倍频差粗估计: [0048] I. Nature of the first training sequence using CAZAC sequence is completed the coarse timing estimation portion of the fractional frequency difference between the coarse and estimated:

[0049] 设接收到的一帧信号序列离散表示为r (η),n=l,2··· ,N。 [0049] provided to a received signal sequence is represented as a discrete r (η), n = l, 2 ···, N. [0050] 利用CAZAC序列相同的重复部分的自相关进行峰值平台搜索,将所有大于最大相关值的O. 95倍的定时位置构成的集合称为粗定时范围,取粗定时范围中所有值的平均作为粗定时估计值Θ opt : Overlapping portions of the same autocorrelation [0050] using a CAZAC sequence a peak search platform, the set of all correlation values ​​greater than the maximum timing position O. 95 times the coarse timing is referred to as range, a coarse timing range taking the average of all the values as the coarse timing estimate Θ opt:

[0051] Θ opt=mean (U) [0051] Θ opt = mean (U)

[0052] 其中,mean(·)表示求均值,U为峰值平台范围; [0052] wherein, mean (·) denotes averaging, U is a peak range of the platform;

[0053] U= {d labs (Μ (d)) ^ O. 95Q, de {1,2-,Q}};其中,Q 为接收序列的长度,abs ( ·) [0053] U = {d labs (Μ (d)) ^ O. 95Q, de {1,2-, Q}}; wherein, Q is the length of the received sequence, abs (·)

表示求幅度函数; It means sum amplitude function;

I N-1 I I N-1 I

[0054] C1 二argmax{M(d)) ,M(d)= IP(d) I /(R(d)), R{d) — (d + λ)| , [0054] C1 two argmax {M (d)), M (d) = IP (d) I / (R (d)), R {d) - (d + λ) |,

ά ^=O ά ^ = O

Njl-1 Njl-1

=Σ r、d + k、r'd + k + ;V/2) ;argmax(M丨J))表示返回M(d)为最大值时对应的d值; = Σ r, d + k, r'd + k +; V / 2); argmax (M Shu J)) which returns M (d) d is the value corresponding to the maximum;

[0055] 利用粗定时估计值Θ opt对应位置的相关值的相位估计出粗小数倍频偏έΡ : [0055] The use of coarse timing estimation value Θ opt phase correlation value corresponding to the coarse position estimate small frequency offset times έΡ:

[0056] iF = angle(P (θορί ))/π [0056] iF = angle (P (θορί)) / π

[0057] 其中,angle(·)为求角度函数,Ρ(·)为求相位函数。 [0057] wherein, angle (·) is a function of the angle demand, Ρ (·) is a function of calculating the phase.

[0058] 2.利用第二训练序列中PN序列部分的性质来完成多径中最强路径时延4的估计: [0058] 2. accomplished strongest multipath estimating delay path 4 by the nature of the second portion of the PN sequence training sequence:

[0059] 使用小数倍频偏粗估计办对接收序列r(n)进行小数倍频偏补偿,得到小数倍频偏补偿后的序列r_p(n): [0059] using a small multiple of the coarse frequency offset estimation do received sequence r (n) for multiple small frequency offset compensation, to obtain sequence r_p (n) after a small multiple of the frequency offset compensation:

[0060] rcomp (ii) = r{n) ■ ejlKCFn « = {1,2···, Q} [0060] rcomp (ii) = r {n) ■ ejlKCFn «= {1,2 ···, Q}

[0061] 利用已知的PN同步序列与接收的小数倍频偏补偿后的PN序列完成互相关,搜索互相关结果的峰值出现的位置来获得最强路径时延 [0061] using known PN synchronization sequence with the complete PN sequence after frequency offset compensation received multiple small cross-correlation, mutual position of the peak search of correlation results to obtain the strongest path delay

[0062] I)从峰值平台范围U中选取一个值k,即ke U,求得W(k,η)与E (k)。 [0062] I) U is selected from a range of internet peak value k, i.e. ke U, to obtain W (k, η) and E (k). W(k,n)表示在定时位置k时刻η接收到的进行了小数倍频偏补偿后的PN序列与已知的PN序列的相关性,E(k)表示在定时位置k时η接收到的进行了小数倍频偏补偿后的PN序列与已知的PN序列的相关性: W (k, n) η represents the timing position of received at time k be the correlation between the PN sequence and the known PN sequence after a small multiple frequency offset compensation, E (k) η represents the reception timing position at the k to the correlations between the PN sequence to the known PN sequence after a small multiple of the frequency offset compensation:

[0063] H) = /:■ ik + GI + N+a)xS,,(//) η = [0063] H) = /: ■ ik + GI + N + a) xS ,, (//) η =

[0064] 其中,N为PN序列长度,GI为循环前缀CP的长度,A2(〃)为已知的第二训练序列,X表示数据点乘;[0065] [0064] where, N is the length of the PN sequence, GI is the length of the CP, A2 (〃) is a known second training sequence, X represents the data point multiplication; [0065]

Figure CN102882670AD00081

[0066] 2)利用相关性W(k,n)与E(k)得到多径中最强路径时延古的估计作为粗同步位置 [0066] 2) using the correlation between W (k, n) and E (k) to give the strongest multi-path estimation path delay coarse synchronization position as ancient

[0067] 3.利用CAZAC和PN同步序列完成整数倍频率偏移乏估计: [0067] 3. The use of CAZAC sequences and PN synchronization complete lack of an integral multiple of the frequency offset estimation:

[0068] 将粗同步位置向前移动λ。 [0068] The coarse synchronization position is moved forward λ. 个位置: Position:

[0070] 此处λ。 [0070] Here λ. 的值为一个经验值,根据最大多径时延和循环前缀长度之间的关系确定。 Value is an empirical value, determined according to the relationship between the maximum multi-path delay and the cyclic prefix length. 一般取循环长度的1/4。 Generally the 1/4 cycle length.

[0071] 使用粗定时位置^以及小数倍频偏粗估计尽对接收序列进行补偿,获得小数倍频率补偿后的时域PN序列*\和CAZAC序列〜. [0071] using the coarse timing location and the time domain PN sequences ^ * \ ~ CAZAC sequence and a small multiple of the coarse frequency offset estimation compensation to make the received sequence, to obtain a small multiple of the frequency compensation.

[0073] 对时域CAZAC序列&和PN序列A2以及已知的CAZAC序列序列Sis进行FFT : [0073] & CAZAC sequence on the time domain and A2 and the known PN sequence CAZAC sequence Sis for FFT:

Figure CN102882670AD00082

[0076] 接收并补偿后的CAZAC序列&和PN序列P2的频域数据与已知的CAZAC序列Sii和 [0076] The frequency domain data & CAZAC sequence and the PN sequence P2, and the receiving compensation CAZAC sequence with known and Sii

PN序列&的频域数据进行滑动相关,从而获得整数倍频率偏移之: Frequency domain PN sequence & sliding correlation data to obtain a frequency offset of an integer multiple of:

Figure CN102882670AD00083

[0078] 其中,V(k)为已知的CAZAC序列和PN序列频域之间的关系 [0078] wherein, V (k) is the relationship between the CAZAC sequences and frequency domain PN sequence known

Figure CN102882670AD00084

[0080] 最终整数倍频偏估计结果为: [0080] The final results for the integer frequency offset estimation:

[0081] [0081]

Figure CN102882670AD00085

[0082] 得到粗的频率估计结果为矣=Sc^ep。 [0082] The coarse frequency estimation result obtained for the carry = Sc ^ ep.

[0083] 4.利用第二训练序列FFT完成信道响应估计,估计出第一条路径的时延并完成精定时。 [0083] 4. The complete FFT training sequence using the second channel response estimate, estimated first path delay and complete refined timing.

[0084] 使用整数倍频率偏移毛对补偿后的接收序列再进行整数倍频率补偿得到序列 [0084] The use of an integral multiple of the frequency shift of the received sequence hair compensated again multiplied by an integer frequency compensation to obtain sequence

Figure CN102882670AD00086

[0086] PN序列表示为:[0087] &舰=,:啊ί;ι:1,(" + Θ + Ν + 67),π = 0/1,···^-] [0086] PN sequence is represented as: [0087] & ship ,: ah = ί; ι: 1, ( "+ Θ + Ν + 67), π = 0/1, ··· ^ -]

[0088] 对整数倍频率补偿后的第二训练序列中PN序列进行快速傅里叶变换FFT来获得信道响应估计; [0088] The second training sequence after the integral multiple of the frequency compensation PN sequence FFT fast Fourier transform to obtain channel response estimates;

[0089] Y = ) · [0089] Y =) ·

ί ί

[0090] 信道频域估计向量为: [0090] The estimated channel frequency-domain vectors:

[0091] H = | = ["(0y:T气…,"⑷#攻,.·//(乂— Ι)#叫r] [0091] H = | = [. "(0y: T gas ...," ⑷ # attack, · // (qe - Ι) # called r]

■I ο " ? ■ I ο "?

[0092] 再经快速傅里叶变换IFFT,Y = FFTK,n.),获得信道时域响应: [0092] and then the Fast Fourier Transform IFFT, Y = FFTK, n), the time domain response to obtain a channel:

[0093] IFFT得到时域信道冲击响应估计为: [0093] IFFT time-domain channel impulse response estimate as:

^ NI -j2r— ^ NI -j2r-

[0094] = 一、 1 = 1,2···,# [0094] a = 1 = 1,2 ..., #

k=Q k = Q

[0095] 估计得到最强路径ft— =max丨/;, :/ = 0Λ··_ί;"的幅值,利用一定窗长信道能量作 [0095] the estimated strongest path ft- = max Shu / ;,: / = 0Λ ·· _ί; "amplitude, the use of certain window for a long channel energy

为判决准则确定第一条路径的时延。 Determining the delay of the first path to sentencing guidelines. 第一条路径的估计函数可以表示为: The first path estimation function may be expressed as:

[0096] τ = argmax{£A (1):1 = 0,1,···,GI-K+} [0096] τ = argmax {£ A (1): 1 = 0,1, ···, GI-K +}

- 2 . - 2 .

Ύ\ hk , if h, >σ·h Ύ \ hk, if h,> σ · h

[0097]式中 Α'„(/) = Ή 1 ' . ' m"X [0097] wherein Α ' "(/) = Ή 1'. 'M" X

ί U. otherwise ί U. otherwise

[0098] 式中Eh(I)表示从I处开始的窗长为K+的信道能量,信道时延扩展L〈K+。 [0098] the Eh in the formula (I) represents a window length I from the starting energy of K + channel, the channel delay spread L <K +. σ为选择第一条路径的门限。 σ threshold to select the first path.

[0099] 在获得第一条路径的估计位置f以后,我们可以获得精定时位置Θ为: [0099] After obtaining the first path estimated position f, we can obtain a refined timing Θ is position:

[0100] θ = θ + τ [0100] θ = θ + τ

O O

[0101] 5.利用ML准则获得小数倍频偏精估计: [0101] The ML criterion is obtained by using a small multiple of the frequency offset estimation precision:

[0102] 由上步得到的Θ更新接收数据。 [0102] obtained by the above step Θ received update data. 用来表示更新后的PN序列的数据: PN sequences used to represent the data of the updated:

[0103] 4腳=rcompnm (η + θ + Ν + GI),H=OX--NI [0103] 4 feet = rcompnm (η + θ + Ν + GI), H = OX - NI

[0104] 此时,接收端得到的更新后的PN序列为已知的发送端发送的PN序列为 [0104] In this case, the PN sequence updated at the receiving end are known PN sequence is transmitted by the transmitting end

[0105] 对精定时更新后的第二训练序列中PN序列利用最大似然ML准则获得小数倍频率 [0105] After the second training sequence to update the refined timing PN sequence maximum likelihood ML criterion to obtain a small multiple of the frequency

偏移精估计L获得小数倍频偏精估计的代价函数为: L offset estimation precision fine frequency offset estimate to obtain a small multiple of the cost function is:

Γ η w I Γ η w I

[0106] ε = argmax jr, W(S)BWff (f)(r^ ) : ε = ελ+ηι— jn = {-10,-9,---9,10} 1 上式 [0106] ε = argmax jr, W (S) BWff (f) (r ^): ε = ελ + ηι- jn = {-10, -9, --- 9,10} 1 formula

中:乓为之前步骤中得到的粗估计,B=S(ShS) In: pong crude estimate obtained in the previous step, B = S (ShS)

[0107] [0107]

V, (O) V, (-1)…,v(-67 + l) V, (O) V, (-1) ..., v (-67 + l)

λ·, (I) λ, (O)…s, (-G/ + 2) λ ·, (I) λ, (O) ... s, (-G / + 2)

S= " * 2 S = "* 2

Jh(NI) Sh(NI)…St2(N-GI) _ Jh (NI) Sh (NI) ... St2 (N-GI) _

[0108] ff(e)=diag{l,eJ2"E /N, e J川ε (ν-ι) ε /n}[0109] diag( ·)为对角矩阵函数,F表示小数倍频率搜索范围,可以根据粗频偏估计的精度适当调整F值的大小,一般情况下取为O. 005。 [0108] ff (e) = diag {l, eJ2 "E / N, e J Chuan ε (ν-ι) ε / n} [0109] diag (·) function is a diagonal matrix, F represents a small multiple of the frequency the search range can be appropriately adjusted according to the size of the F value of the accuracy of the coarse frequency offset estimation, generally taken to be O. 005.

[0110] 仿真实验以成都移动手持电视为例,考虑到CMMB信号的应用环境,信号的参数为:载波频率为fe=506MHz,仿真I帧CMMB,I帧划分为40个时隙,每个时隙的长度25ms。 [0110] In simulation Chengdu mobile handheld TV, for example, taking into account the application environment CMMB signals, signal parameters are: carrier frequency fe = 506MHz, CMMB simulation I frame, the I frame is divided into 40 time slots, each time gap length 25ms. CMMB同步信号时间长度为204. 8 μ S,每个CMMB符号的长度为460. 8us。 CMMB length of the synchronizing signal time 204. 8 μ S, the length of each symbol is CMMB 460. 8us. 信道最大时延扩展的长度为L=50,保护间隔GI=128,载波个数N=1024,频偏e=6. 4。 The length of the maximum delay spread of the channel is L = 50, a guard interval GI = 128, the number of carriers N = 1024, the frequency offset e = 6. 4. 仿真移动通信信道采用ITU-M. 1225 Vehicle Channel B信道(考虑速度v2=60km/h时)的条件,信号长度为P= (N+GI) *80=46080。 Simulation of a mobile communication channel using ITU-M. 1225 Vehicle Channel B channel (considering the speed of v2 = 60km / h when) condition, the length of the signal P = (N + GI) * 80 = 46080. 信道条件见下表: Channel condition table below:

[0111] [0111]

Figure CN102882670AD00101

[0112] 具体实施例的方法包括以下步骤: [0112] Specific embodiments of the method comprises the steps of:

[0113] a)从接收信号利用第一训练序列中CAZAC序列部分的性质完成粗定时估计以及小数倍频差粗估计。 Properties [0113] a) from the received signal using the CAZAC sequence in a first portion of the training sequence is completed, and the coarse timing estimation estimates the fractional frequency difference crude. 符号粗定时偏移估计表达式为: The crude symbol timing offset estimation expression is:

[0114] Θ opt=mean (U) [0114] Θ opt = mean (U)

[0115] 其中:mean(U)表示求U中所有值的均值,U为峰值平台范围。 [0115] wherein: mean (U) represents the mean of all values ​​required U, U is a peak range of the platform.

[0116] [0116]

Figure CN102882670AD00102

[0119] 小数倍频率偏差粗估计&利用符号定时位置Θ _的Ρ( Θ opt)获得,如下式所示。 [0119] The crude small multiple of the frequency offset estimate using & _ symbol timing [Theta] position of Ρ (Θ opt) is obtained, as shown in the following formula.

[0120] [0120]

Figure CN102882670AD00103

[0121] b)使用小数倍频偏粗估计4对接收序列r(n)进行小数倍频偏补偿,得到小数倍频偏补偿后的序列r_p (η)。 [0121] b) coarse frequency offset estimation using a small multiple of four pairs of the received sequence r (n) for multiple small frequency offset compensation, to obtain sequence r_p (η) after multiple small frequency offset compensation.

[0122] 利用小数倍频偏补偿后的PN序列完成多径中最强路径时延g的估计,具体方法为:利用已知的PN同步序列与接收的小数倍频偏补偿后的PN序列完成互相关,搜索互相关结果的峰值出现的位置来获得最强路径时延^利用第二训练序列中PN序列部分的性质来完成多径中最强路径时延的估计 After the PN sequence [0122] using a small multiple of the estimated frequency offset compensation is completed strongest path delay multipath g, a specific method is: after a small multiple of PN frequency offset compensation using the known synchronization sequence and the received PN cross correlation sequence is completed, the position of the cross correlation peak search result appears strongest path delay is obtained by using the properties ^ PN sequence portion of a second training sequence to complete the multi-path delay estimation strongest path

[0123] 将接收信号r (η)完成小数倍频差补偿,得到序列r_p(n)。 [0123] The received signal r (η) fractional frequency difference compensating completed, the sequence obtained r_p (n).

[0124] rcomp(n) = r(n)-e-J2— " = {1,2·.·,Ρ}[0125] 那么多径中最强路径的时延^可以按照下列步骤获得。 [0124] rcomp (n) = r (n) -e-J2- "= {1,2 ·. ·, Ρ} [0125] So the strongest multipath delay path ^ can be obtained according to the following steps.

[0126] 1)从范围U中选取一个值k,即ke U。 [0126] 1) selecting a value from the range U k, i.e. ke U.

[ [

Figure CN102882670AD00111

[0130] c)利用CAZAC和PN同步序列完成整数倍频率偏移毛估计。 [0130] c) using CAZAC sequences and PN synchronization completion of an integer multiple of the frequency offset estimation hair.

[0131] 将粗同步位置向前移动入。 [0131] The coarse synchronization position is moved forward into. =20个位置,SP古= ^-20。 = 20 positions, SP ancient ^ = -20. 按照类似于SCA的方法进行整数倍频差毛的估计。 Integer frequency difference between the estimated gross conduct of an analogous manner to the SCA.

Figure CN102882670AD00112

[0136] 得到整数倍频率偏差估计表示式为: [0136] to obtain an integer multiple of the frequency offset estimator is represented by the formula:

2 2

Figure CN102882670AD00113

[0138] 得到最终整数倍频率偏差估计结果为: [0138] to obtain a final integral multiple of the frequency offset estimation result:

[0139] [0139]

Figure CN102882670AD00114

[0140] d)利用FFT完成信道响应估计,估计出第一条路径的时延并完成精定时。 [0140] d) A FFT complete channel response estimate, estimated first path delay and complete refined timing.

[0141] 利用FFT获得信道时域冲击响应估计以后,需要适当的选择σ的值。 [0141] to obtain a time domain channel impulse response estimate using an FFT later, appropriate selection of the value of σ. σ为选择第一条路径的门限,σ不能太大,太大可能导致漏检第一路径。 selection threshold [sigma] is the first path, [sigma] is not too large, too much may result in missed detection of the first path. σ的值也不能太小,太小会增加噪声被误检为多径的第一条的可能性。 The value of σ is not too small, the possibility of noise is first multipath error detection increases too small. σ值的选取可以按照下式进行: Σ values ​​may be selected according to the following equation:

Figure CN102882670AD00115

[0143] σ i是在信噪比为SNR1时仿真获得的参考值,σ 2为在信噪比为SNR2的情况下获得的门限值。 [0143] σ i is a reference SNR value SNR1 is obtained by simulation, σ 2 is the threshold SNR is obtained in the case SNR2.

[0144] 通过仿真得到在SNR1=IOdB时σ 1=0. 05,可以取A 0 °S '考虑到实际移 [0144] σ 1 = 0. 05 obtained by simulation SNR1 = IOdB time, can take A 0 ° S 'taking into account the actual shift

V IO !() V IO! ()

动通信信道的特点,取窗长K+=54,按照说明书步骤4可以估计出精定时位置。 Dynamic characteristics of the communication channel, windowing length K + = 54, a refined timing position can be estimated according to the instructions in step 4.

[0145] e)利用ML准则获得小数倍频率偏差精估计。 [0145] e) using the ML criterion to obtain a small multiple of the frequency offset estimation precision.

[0146] 由上步得到的Θ更新PN序列的数据[0147] [0146] PN sequence data update Θ obtained by the above step [0147]

Figure CN102882670AD00121

[0148] 已知发送端发射的PN序列为\ ;将频差搜索范围的值设置为F=O. 005,那么利用ML准则可以获得小数倍频率偏差精估计的代价函数为: [0148] PN sequences are known to the transmitting side transmitted \; set the value of the search range of the frequency difference F = O 005, then the ML criterion can be obtained by using a small multiple of the frequency offset estimate fine cost function:

[0149] [0149]

Figure CN102882670AD00122

通过 by

仿真在低信噪比的情况下,两种同步信号的符号定时和频差估计性能,并将结果进行比较来衡量新同步信号以及同步机制的性能。 Simulation at low signal to noise ratio, symbol timing and frequency offset estimation performance of both the synchronization signal, and comparing the results to measure the performance of a new synchronization signal and the synchronization mechanism. 在比较过程中,将SCA算法也作为一个比较性能的参考。 During the comparison, the reference SCA algorithm is also a performance comparison.

[0150] 从附图4中,我们可以得到在低信噪比的情况下,新的同步信号能达到更好的定时。 [0150] From Figure 4, we can get at a low SNR, a new synchronizing signal to achieve better timing. 从附图5中,CMMB-PN曲线为以PN序列为同步信号的频差估计MSE曲线,new way fine曲线为新的同步信号的频差估计MSE曲线。 From the figures 5, CMMB-PN curve of PN sequence synchronization signal frequency difference estimation MSE curve, new way fine frequency difference curve as a new synchronization signal estimation MSE curve. SCA曲线为SCA算法的频差估计MSE曲线。 SCA SCA frequency difference curve estimation algorithm MSE curve. 我们可以发现在低信噪比的情况下,本发明的同步信号能达到更好的定时估计和频偏估计。 We found that in low SNR, a synchronization signal present invention can achieve better timing estimation and frequency offset estimation.

Claims (7)

1. 一种基于CMMB信号的同步处理方法,利用CMMB同步信号对CMMB信号的小数倍频率偏移、整数倍频率偏移以及CMMB信号的起始时间进行估计从而完成同步处理,其特征在于,所述CMMB同步信号由第一训练序列与第二训练序列组成,所述第一训练序列包括长度为N的恒包络零自相关CAZAC序列以及长度为GI的循环前缀,所述第二训练序列包括长度为N的PN序列以及长度为GI的循环前缀; 同步处理的具体步骤如下: 1)利用第一训练序列中CAZAC序列完成粗定时偏移估计以及小数倍频偏粗估计& 2)使用小数倍频偏粗估计办对接收序列进行小数倍频偏补偿,利用小数倍频偏补偿后的PN序列完成多径中最强路径时延#的估计,将最强路径时延识乍为粗定时位置; 3)使用粗定时位置《以及小数倍频偏粗估计4对接收序列进行补偿,利用补偿后的CAZAC序列和PN序列完成整数倍频率偏移毛 A synchronization processing method CMMB signals based on the sync signal using the CMMB CMMB signals of the smaller multiple of the frequency shift of an integer multiple of the frequency offset and the start time CMMB signal is estimated to complete the synchronization process, wherein, the CMMB training sequence from the first synchronization signal and the second training sequences, the first training sequence of length N comprises a constant Amplitude zero autocorrelation CAZAC sequence cyclic prefix and the length of the GI, the second training sequence comprising a PN sequence of length N and a cyclic prefix length of the GI; sync specific steps are as follows: 1) a first training sequence using CAZAC sequence is completed and a coarse timing offset estimation multiple small frequency offset estimation crude & 2) the crude multiple small frequency offset estimate the received sequence do small multiple frequency offset compensation, using the PN sequence after the completion of a small multiple of the frequency offset compensation strongest multipath # path delay estimation, the strongest path delay identification coarse timing location for the first glance; 3) using the coarse timing location "coarse frequency offset estimation, and small multiple of four pairs of compensating received sequence, CAZAC sequence and the PN sequence by an integer multiple of the frequency offset compensation is completed hair 估计,将整数倍频率偏移毛作为CMMB信号的整数倍频率偏移; 4)使用整数倍频率偏移毛对补偿后的接收序列再进行整数倍频率补偿,对整数倍频率补偿后的第二训练序列中PN序列进行快速傅里叶变换FFT完成信道响应估计,估计出第一条路径的时延f并完成精定时位置Θ的估计,(9 = 3+卩,将精定时位置Θ作为CMMB信号的起始时间; 5)使用精定时位置Θ更新整数倍频率补偿后的接收序列,对精定时更新后的第二训练序列中PN序列利用最大似然ML准则获得小数倍频率偏移精估计^,将小数倍频率偏移精估计S作为CMMB信号的小数倍频率偏移; 6)利用确定的CMMB信号的小数倍频率偏移、整数倍频率偏移以及CMMB信号的起始时间完成同步处理。 Estimating the frequency offset an integral multiple of an integer multiple of the frequency shift of the hair as CMMB signals; 4) are an integer multiple of the frequency offset hair compensated received sequence and then multiplied by an integer frequency compensation, after a second integer multiple of the frequency compensation PN sequence training sequence to complete a fast Fourier transform FFT channel response estimate, estimated first path delay f and complete position estimate refined timing Θ of (3 + 9 = Jie, the fine timing position Θ as CMMB start time signal; 5) using the refined timing Θ location update received sequence of an integer multiple of the frequency compensation for the second training sequence PN sequence refined timing update maximum likelihood ML criterion to obtain a small multiple of the frequency offset fine ^ estimation, a small multiple of the frequency offset estimation precision S CMMB signals as a small multiple of the frequency offset; 6) CMMB signals using the determined small multiple of the frequency offset, and the offset of an integer multiple of the frequency of the starting signal CMMB time to complete the synchronization process.
2.如权利要求I所述一种基于CMMB信号的同步处理方法,其特征在于,第一训练序列中CAZAC序列Stl表不为: I as claimed in claim 2 for synchronizing the signal processing method based CMMB, wherein the first training sequence is not a CAZAC sequence table Stl:
Figure CN102882670AC00021
其中,构成CAZAC序列的2个相同序列c (η)为c (n) =exp (j π η2/ (Ν/2)), exp ( ·)表示以自然对数e为底的指数函数,N为子载波的个数。 Wherein constituting the CAZAC sequence two identical sequences c (η) as c (n) = exp (j π η2 / (Ν / 2)), exp (·) represents the natural logarithm of the base e exponential function, N subcarrier number.
3.如权利要求I所述一种基于CMMB信号的同步处理方法,其特征在于,第二训练序列中PN序列st2表不为: I claim the A synchronization processing method based CMMB signals, wherein the second training sequence is not the PN sequence st2 table:
Figure CN102882670AC00022
其中,i=0,- ,N-I0 Where, i = 0, -, N-I0
4.如权利要求2所述一种基于CMMB信号的同步处理方法,其特征在于,步骤I)的具体方法为: 利用CAZAC序列相同的重复部分的自相关进行峰值平台搜索,将所有大于最大相关值的O. 95倍的定时位置构成的集合称为粗定时范围,取粗定时范围中所有值的平均作为粗定时估计值Θ opt ; 利用粗定时估计值Θ Qpt对应位置的相关值的相位估计出粗小数倍频偏4 ·» 来完成粗符号定时和粗小数倍频偏估计七fr = migle(P(eupt))丨π,其中,Ρ( ·)为相位函数,angle ( ·)为角度函数。 As claimed in claim 2 for synchronizing the signal processing method based CMMB, wherein step I) is a specific method: a peak search internet autocorrelation same repeating portions CAZAC sequence, all greater than the maximum relevant O. 95 times the set value of the timing position is referred to as a coarse timing range, taking the average of the range of a coarse timing Θ opt all values ​​as a coarse timing estimate; coarse timing phase estimation value [Theta] using correlation values ​​corresponding to the position estimate Qpt the crude 4-small offset times »accomplished coarse symbol timing and coarse frequency offset estimation seven times smaller fr = migle (P (eupt)) Shu π, wherein, Ρ (·) is the phase function, angle (·) as a function of angle.
5.如权利要求3所述一种基于CMMB信号的同步处理方法,其特征在于,步骤2)中利用小数倍频偏补偿后的PN序列完成多径中最强路径时延^的估计,具体方法为:利用已知的PN同步序列与接收的小数倍频偏补偿后的PN序列完成互相关,搜索互相关结果的峰值出现的位置来获得最强路径时延g。 5. The estimated as claimed in claim 3 A synchronization processing method based CMMB signals, wherein, in step 2) using a small multiple of the PN sequence complete frequency offset compensation strongest path delay multipath ^, and specific methods: cross-correlation using known PN sequence with the PN synchronization sequence is completed after a small multiple of the received frequency offset compensation, the mutual position of the peak search of correlation results to obtain the strongest path delay g.
6.如权利要求I所述一种基于CMMB信号的同步处理方法,其特征在于,步骤3)中利用补偿后的CAZAC序列和PN序列完成整数倍频率偏移毛的估计,具体方法为: 利用接收并补偿后的CAZAC序列和PN序列的频域数据与已知的CAZAC序列和PN序列的频域数据进行滑动相关,从而获得整数倍频率偏移S I claim 6. A synchronization processing method based CMMB signals, characterized in that, the CAZAC sequence and the PN sequence using the compensated completion of step 3) is an integral multiple of the frequency offset estimation hair, specific methods: using frequency domain data and frequency domain data and the known PN sequence CAZAC sequence CAZAC sequence and the received PN sequence and compensation sliding correlation, to thereby obtain an integral multiple of the frequency offset S
7.如权利要求3所述一种基于CMMB信号的同步处理方法,其特征在于,步骤4)中利用整数倍频率补偿后的第二训练序列中PN序列得到精定时位置Θ的具体方法是: 对整数倍频率补偿后的第二训练序列中PN序列进行快速傅里叶变换FFT来获得信道响应估计,再经快速傅里叶变换IFFT获得信道时域响应,利用利用信道的能量准则作为判决量估计多径中的第一路径时延f,完成精定时位置Θ的估计,= 3 7. The method of Claim synchronization processing based CMMB signals, wherein step 4) using a second training sequence after the integral multiple of the frequency compensation method of fine PN sequence specific timing position Θ is obtained: the second training sequence of an integer multiple of the frequency compensation PN sequence FFT fast Fourier transform to obtain a channel response estimate, then by fast Fourier transform IFFT channel time-domain response is obtained by using the channel energy criterion of judgments as estimating multipath delays in the first path F, complete position estimate refined timing of Θ =
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