CN101902129B - Current-type multi-resonance direct current (DC) converter - Google Patents

Abstract

The invention relates to direct current (DC) step-up power conversion technology and discloses a current-type multi-resonance DC converter. The current-type multi-resonance DC converter comprises a square-wave current source generator, a multi-resonance network and a rectification filtering output unit which are sequentially connected in series, and is characterized in that: the multi-resonance network comprises a transformer, a parallel resonant inductor, a parallel resonant capacitor and a serial resonant inductor, wherein the serial resonant inductor is connected with a primary side of the transformer; the rectification filtering output unit comprises a diode rectifying circuit and a filtering capacitor which is connected with an output end of the diode rectifying circuit in parallel.

Description

A kind of current-type multi-resonance direct current (DC) converter

Technical field

The present invention relates to the DC boosting power conversion technology, particularly a kind of current-type multi-resonance direct current (DC) converter.

Background technology

The environmental problem of bringing along with fossil fuel the exhausted day by day of serious day by day and traditional energy that become, the new forms of energy industry has obtained developing fast.Yet, comprising that many emerging energies of solar energy and fuel cell all have the direct current output characteristic of low-voltage and high-current, high efficiency boost power converter technique becomes the key of effectively utilizing these emerging energies.

The voltage-type booster converter has simple in structure, controls advantages such as easy, yet, the place one's entire reliance upon hypermutation ratio of transformer of its boost function.It is relatively poor to have the often former secondary coupling of the transformer of high step-up ratio, and leakage inductance, turn-to-turn capacitance are bigger, can in circuit, produce the voltage and current spine, and this has not only reduced the operating frequency of transformer, has also limited the operating efficiency of converter simultaneously.In addition, emerging energies such as solar energy and fuel cell receive the influence of current ripples bigger, and the input current of voltage-type booster converter is pulsed, and must introduce bigger input filter and eliminate current ripples, and this has increased the volume and the cost of converter greatly.

Compare with voltage source converter; Current type converter has outstanding advantages such as the high and current ripples of boost capability is little; And adopt the current type converter of parallel resonance technology can further improve voltage gain, thereby further reduce transformer voltage ratio and parasitic parameters such as corresponding leakage inductance and turn-to-turn capacitance.

The former limit switching tube of traditional current mode parallel resonance DC converter mostly is operated in no-voltage and opens (ZVS:Zero Voltage Switching) state; Yet application scenario in the low-voltage, high-current input; ZVS is not very important, realize zero-current switching (ZCS:Zero Current Switching) to switching loss to reduce but be most important.For current mode parallel resonance DC converter, transformer leakage inductance still exists, and can cause the overvoltage of switching tube equally.Some current-type multi-resonance direct current (DC) converters are realized the ZCS of former limit switching tube through the resonance of introducing transformer leakage inductance and parallel resonance electric capacity; But in order to realize the ZCS in the full-load range; Must control former limit leakage inductance according to full load conditions; This can cause converter when underloading, to have bigger circulation to flow through the inverse parallel diode of former limit switching tube, thus the operating efficiency when having a strong impact on underloading.

Too much circulation energy also can increase the on-state loss of converter in the resonant network, reduces the operating efficiency of converter.Thereby someone introduces transformer and comes the extra resonant energy of feedback to reduce circulation loss, but the transformer of introducing can increase the volume of converter, and influence quality factor, so this method and be not suitable for the occasion that high power density requirement and pressure regulation requirement are arranged.

In addition, still there is reverse-recovery problems in the rectifier diode of current mode parallel resonance DC converter, and the voltage stress of rectifier diode is still higher, is difficult to select withstand voltage low Ultrafast recovery diode with lower on-state pressure drop as rectifier diode.

Summary of the invention

The present invention is directed to the deficiency that above-mentioned existing current mode resonance DC converter exists, its purpose is to provide a kind of novel current-type multi-resonance direct current (DC) converter.This converter can provide higher voltage gain, in full-load range, all has less circulation, and switching tube and rectifier diode all are operated in the ZCS state, and underloading also has higher efficient when working.

In order to achieve the above object, the present invention adopts following technical scheme to be achieved.

A kind of current-type multi-resonance direct current (DC) converter; The square wave current source generator, multi resonant vibrating network, the rectifying and wave-filtering output unit that comprise series connection successively; It is characterized in that; Said multi resonant vibrating network comprises transformer, parallel resonant inductor, parallel resonance electric capacity and series resonance inductance, and said series resonance inductance is connected the former limit of transformer; Said rectifying and wave-filtering output unit comprises diode rectifier circuit and the filter capacitor that is connected in parallel on the diode rectifier circuit output.

Characteristics of the present invention and effect are explained as follows:

(1) said square wave current source generator adopts current mode half-bridge structure, current mode full bridge structure or current mode push-pull configuration to constitute.The square wave current source generator can produce square wave current, has less input current ripple, and ripple frequency is the twice of switching frequency, can be through this ripple of polarity free capacitor filtering of low-cost small size.

(2) multi resonant vibrating network comprises transformer, parallel resonant inductor, parallel resonance electric capacity and series resonance inductance, and the series resonance inductance is connected the former limit of transformer.When the conducting simultaneously of the switching tube of square wave current source generator; Series resonance inductance and parallel resonance electric capacity generation resonance; Make the electric current that flows into multi resonant vibrating network that soft switching-over take place, realized the ZCS of switching tube, reduced the switch tube voltage spike that the series resonance inductance causes simultaneously.The resonance that multi resonant vibrating ruton is crossed shunt inductance and shunt capacitance is that converter provides high voltage gain.In the multi resonant vibrating network; The high voltage gain that shunt inductance and shunt capacitance provide through resonance; Can further reduce transformer voltage ratio and parasitic parameters such as relevant with it leakage inductance and turn-to-turn capacitance; At this moment, the diode junction capacitance of the turn-to-turn capacitance of transformer and rectifying and wave-filtering output unit is used as parallel resonance electric capacity, can be by good restraining by the current spike that they cause.

(3) switching tube of said square wave current source generator open duty ratio greater than 0.5, form one section half the duty ratio overlap the time (switching tube open duty ratio, less than 0.7) greater than 0.5 less than the harmonic period of series resonance inductance and parallel resonance electric capacity.In the time that duty ratio overlaps; The input of multi resonant vibrating network is by short circuit; Series resonance inductance and parallel resonance electric capacity are realized the soft switching-over of multi resonant vibrating network input current through resonance; Multi resonant vibrating network input current after the switching-over flows through its inverse parallel diode after switching tube turn-offs, the switching tube of square wave current source generator can be realized zero-current switching, has reduced the switch tube voltage spike that the series resonance inductance causes simultaneously.

(4) the rectifying and wave-filtering output unit comprises diode rectifier circuit and the filter capacitor that is connected in parallel on the diode rectifier circuit output.Said diode rectifier circuit adopts full bridge rectifier, full-wave rectifying circuit or voltage doubling rectifing circuit.Adopt voltage doubling rectifing circuit can further improve voltage gain, thereby reduce the no-load voltage ratio of transformer and parasitic parameters such as relevant with it leakage inductance and turn-to-turn capacitance, raise the efficiency.

Because the rectifying and wave-filtering output unit adopts the rectification circuit that has filter capacitor; Thereby the maximum voltage of parallel resonance electric capacity can be clamped to output voltage by filter capacitor, reduced the on-state loss that circulation energy and circulation brought in parallel resonant inductor and the parallel resonance electric capacity thus.The electric current of rectifying and wave-filtering output unit rectifier diode of flowing through is multi resonant vibrating network input current and parallel resonant inductor difference between currents; In rectifier diode ON time section; Multi resonant vibrating network input current is approximately constant; The parallel resonant inductor electric current is linear to be increased up to equating that with multi resonant vibrating network input current rectifier diode can be realized zero-current switching thus, has solved the rectifier diode reverse-recovery problems.

During underloading; Because the rectifier diode ON time shortens; Parallel resonance electric capacity is also reduced by the time of clamper relatively; The time of parallel resonant inductor and parallel resonance capacitor resonance can be elongated, makes the initial energy storage of parallel resonance electric capacity and series resonance inductance resonance be able to reduce, thereby can limit the overshoot of converter multi resonant vibrating network input current when underloading is worked.

The present invention's " current-type multi-resonance direct current (DC) converter " is the efficient that has improved boost power converter on the prior art basis (the actual measurement converter can obtain about 95% peak efficiencies under 15 times of conditions of boosting); Greatly reduce the input current ripple, thereby can more efficient use comprise many new forms of energy of solar energy and fuel cell with low-voltage and high-current direct current output characteristic.

Description of drawings

Below in conjunction with accompanying drawing and embodiment the present invention is explained further details.

Fig. 1 is the circuit theory diagrams of current-type multi-resonance direct current (DC) converter of the present invention;

Fig. 2 (a) to (d) comprises the multi resonant vibrating network topological diagram that former secondary does not have centre tapped transformer;

Fig. 3 (a) to (d) comprises the multi resonant vibrating network topological diagram that secondary has centre tapped transformer;

Fig. 4 (a) to (d) comprises the multi resonant vibrating network topological diagram that there is centre tapped transformer on former limit;

Fig. 5 (a) is the rectifying and wave-filtering output unit topological diagram that adopts the full bridge rectifier structure;

Fig. 5 (b) is the rectifying and wave-filtering output unit topological diagram that adopts the full-wave rectifying circuit structure;

Fig. 5 (c) is the rectifying and wave-filtering output unit topological diagram that adopts the voltage doubling rectifing circuit structure;

Fig. 6 is the simplification circuit theory diagrams of current-type multi-resonance direct current (DC) converter;

Fig. 7 is the working mode figure of the simplification circuit of current-type multi-resonance direct current (DC) converter;

Fig. 8 is the simplified model figure of the ac equivalent circuit of current-type multi-resonance direct current (DC) converter;

Fig. 9 is switching tube duty ratio overlapping in the time on former limit, the simplified electrical circuit diagram of resonant network;

Figure 10 (a) to (f) is six working mode figures of current-type multi-resonance direct current (DC) converter;

Figure 11 is the working waveform figure of current-type multi-resonance direct current (DC) converter;

Figure 12 is under fully loaded condition of work, the driving v of two switching tubes in former limit _G1, v _G2, A, B point-to-point transmission voltage v _ABAnd multi resonant vibrating network input current i _PriOscillogram;

Figure 13 is under 20% loaded work piece condition, the driving v of two switching tubes in former limit _G1, v _G2, A, B point-to-point transmission voltage v _ABAnd multi resonant vibrating network input current v _PriOscillogram;

Figure 14 is under fully loaded condition of work, the parallel resonance capacitor C _pVoltage v _Cp, rectifier diode D ₂Voltage v _D2, and the current i that flows into the rectifying and wave-filtering output unit _RecOscillogram;

Figure 15 is under 20% loaded work piece condition, the parallel resonance capacitor C _pVoltage v _Cp, rectifier diode D ₂Voltage v _D2, and the current i that flows into the rectifying and wave-filtering output unit _RecOscillogram;

Figure 16 is under full load conditions, and input inductance L flows through _In1Current i _Lin1, the input current i of converter _InAnd output voltage v _oOscillogram;

Figure 17 is the efficiency curve diagram of current-type multi-resonance direct current (DC) converter, and wherein, abscissa is represented load factor, and ordinate is represented the operating efficiency of converter.

Embodiment

(1) further specifies circuit topological structure of the present invention.

With reference to Fig. 1, for the typical current type multi resonant of the present invention DC converter that shakes, be three grades of series systems, wherein the first order is the square wave current source generator, and the second level is multi resonant vibrating network, and the third level is the rectifying and wave-filtering output unit.

(1) square wave current source generator

Low-voltage dc power supply V _InBe connected between the positive-negative input end of square wave current source generator, between the positive-negative input end of square wave current source generator, be parallel with the low-voltage filter capacitor C respectively _In, first switching branches, second switch branch road.First switching branches is by first inductance L _In1Series connection forms with first switching tube, wherein diode D _Q1, capacitor C _Q1Be switching tube Q ₁Self inverse parallel diode and output capacitance.Second switch props up route second inductance L _In2With second switching tube Q ₂Series connection forms, wherein diode D _Q2, capacitor C _Q2Be switching tube Q ₂Self inverse parallel diode and output capacitance.First inductance L _In1With the first switching tube Q ₁The series connection contact, second inductance L _In2With second switch pipe Q ₂The series connection contact, draw positive-negative output end respectively as the square wave current source generator.Break-make through controlling first switching tube and second switch pipe forms square wave current.

The square wave current source generator can adopt current mode half-bridge structure, current mode full bridge structure or current mode push-pull configuration; The square wave current source generator can produce square wave current; Has less input current ripple; And ripple frequency is the twice of switching frequency, can be through this ripple of polarity free capacitor filtering of low-cost small size.

(2) multi resonant vibrating network

Multi resonant vibrating network comprises transformer T, parallel resonant inductor L _p, the parallel resonance capacitor C _pWith the series resonance inductance L _r, the series resonance inductance L _rBe connected the former limit of transformer, parallel resonant inductor L _p, the parallel resonance capacitor C _pBe connected the transformer secondary, shown in Fig. 2 (a).In conjunction with Fig. 2 (b) parallel resonant inductor L _p, the parallel resonance capacitor C _pAlso can be connected the former limit of transformer; In conjunction with Fig. 2 (c), 2 (d) parallel resonant inductor L _p, the parallel resonance capacitor C _pAlso can be connected to former limit of transformer and secondary.

As shown in Figure 3, transformer T also can be that secondary has centre tapped structure, parallel resonant inductor L _p, the parallel resonance capacitor C _pCan all be connected the former limit of transformer with reference to Fig. 3 (a), also can shown in Fig. 3 (b), all be connected the transformer secondary, perhaps be connected to former limit of transformer and secondary with reference to Fig. 3 (c), 3 (d).

Transformer T also can adopt former sideband as shown in Figure 4 that centre tapped structure is arranged, at this moment the series resonance inductance L _rBe connected with the former limit of transformer centre cap.Parallel resonant inductor L _p, the parallel resonance capacitor C _pCan shown in Fig. 4 (a), all be connected the transformer secondary.Shown in Fig. 4 (b), parallel resonant inductor L _P1And L _P2, the parallel resonance capacitor C _pAlso can all be connected the former limit of transformer.Parallel resonant inductor L _p, the parallel resonance capacitor C _pCan also be connected to former limit of transformer and secondary with reference to Fig. 4 (c), 4 (d).

The series resonance inductance L _rCan be external independent inductance, also can be the leakage inductance of transformer T; Parallel resonant inductor L _pCan be external independent inductance, also can be the magnetizing inductance of transformer; The parallel resonance capacitor C _pCan be the turn-to-turn capacitance of transformer, also can be external independent capacitance.As parallel resonant inductor L _pWith the parallel resonance capacitor C _pThe two one of or when all being placed on the former limit of transformer, the two ends of parallel resonance element are directly parallelly connected with the former limit of transformer two-port, the series resonance inductance L _rPort of an end and the former limit of transformer link to each other, the other end links to each other with a port of square wave current source generator.

(3) rectifying and wave-filtering output unit

The rectifying and wave-filtering output unit comprises diode rectifier circuit and the filter capacitor that is connected in parallel on the diode rectifier circuit output.The diode full-wave rectifying circuit can adopt the full bridge rectifier shown in Fig. 5 (a), can adopt with reference to the full-wave rectifying circuit shown in Fig. 5 (b).Shown in Fig. 5 (c), the diode full-wave rectifying circuit also can adopt voltage doubling rectifing circuit further to improve voltage gain, thereby reduces the no-load voltage ratio of transformer and parasitic parameters such as relevant with it leakage inductance and turn-to-turn capacitance, raises the efficiency.D among Fig. 5 (a) ₁-D ₄Be rectifier diode, C ₀Be filter capacitor, R _LIt is the load resistance of converter.D among Fig. 5 (b) ₁-D ₂Be rectifier diode, C ₀Be filter capacitor, R _LIt is the load resistance of converter.D among Fig. 5 (c) ₁-D ₂Be rectifier diode, C ₁, C ₂Be the multiplication of voltage filter capacitor, R _LIt is the load resistance of converter.

(2) further specify the shake Parameters design of DC converter of typical current type multi resonant shown in Figure 1.

(a) design of resonant network:

When derivation transducer gain G,, suppose that the electric current of inflow series resonant network is a square wave current, the simplified electrical circuit diagram of converter and oscillogram such as Fig. 6 and shown in Figure 7, wherein v because the switching tube duty ratio of the square wave current source generator overlapping time is less _G1And v _G2Be the drive signal of first, second switching tube, V _oBe the output voltage of converter, v _LpBe to flow into L _pElectric current, i _CSBe square wave current, v _RecBe the output voltage of multi resonant vibrating network, i _RecBe the electric current that flows into the rectifying and wave-filtering output unit, v _{Rec_1}, i _{Rec_1}And i _{CS_1}Be respectively v _Rec, i _RecAnd i _CSThrough the fundametal compoment after the Fourier expansion, θ is the conducting phase angle of rectifier diode, and φ, Ψ and α are respectively v _{Rec_1}, i _{Rec_1}And i _{CS_1}Phase angle, δ is v _{Rec_1}And i _{Rec_1}Phase difference, λ is v _{Rec_1}And i _{CS_1}Phase difference.

Because v _{Rec_1}Lag behind i _{Rec_1}, angle of retard is δ, the output of multi resonant vibrating network is capacitive load, thus can draw the simplified model of ac equivalent circuit, as shown in Figure 8, i wherein _{CS_1}Be square wave current i _CSFirst-harmonic after the Fourier expansion, R _eAnd C _eBe the output capacitive loading of multi resonant vibrating network equivalence, Z _InBe the input impedance of ac equivalent circuit, V _{Rec1_pk}The output voltage that is multi resonant vibrating network is through the first-harmonic peak value after the Fourier expansion, I _{CS1_pk}Be to flow into the square wave current of series resonant network through the first-harmonic peak value after the Fourier expansion.R _e, C _e, Z _InAnd I _{CS1_pk}Expression formula like (2), (3), (4) are shown in (5).

Wherein, F is that the switching frequency angular frequency is with respect to series resonant network resonance frequency angular frequency _pNormalized value, Z _pBe the impedance of series resonant network, Q _pBe the quality factor of converter output loading, Q _eBe the loaded quality factor of ac equivalent circuit, expression formula is respectively by formula (6), and (7), (8) are shown in (9).Formula (10) is the expression formula of the efficiency eta of converter.Wherein, N is the no-load voltage ratio of transformer, V _In, I _InBe the input voltage and the input current of converter, V _oBe the output voltage of converter, R _LIt is the load resistance of converter.

$\frac{{V_o}^2}{R_L}=\frac{{{\mathrm{<figure-callout\; id="1"\; label="V\; rec"\; filenames="HSA00000192755900081.png"\; state="\{\{state\}\}">V</figure-callout>}}_{\mathrm{rec}}}^2}{{2R}_e}---\left(1\right)$

$R_e=\frac{{{\mathrm{<figure-callout\; id="1"\; label="V\; rec"\; filenames="HSA00000192755900081.png"\; state="\{\{state\}\}">V</figure-callout>}}_{\mathrm{rec}}}^2}{{{2V}_o}^2}{R}_L---\left(2\right)$

tan(|δ|)＝ωC _eR _e (3)

$\left|Z_{\mathrm{in}}\right|=\frac{{\mathrm{<figure-callout\; id="1"\; label="V\; rec"\; filenames="HSA00000192755900081.png"\; state="\{\{state\}\}">V</figure-callout>}}_{\mathrm{rec}}}{I_{\mathrm{CS}}}=\frac{Q_{\mathrm{<figure-callout\; id="1"\; label="e\; Z\; p"\; filenames="HSA00000192755900081.png"\; state="\{\{state\}\}">e</figure-callout>}}{Z}_p{}_{}}{\sqrt{1+{[Q_e(\frac{1}{F}-F)-\tan \left(\right|\mathrm{\&delta;}\left|\right)]}^2}}---\left(4\right)$

${\mathrm{<figure-callout\; id="1"\; label="I\; CS"\; filenames="HSA00000192755900081.png"\; state="\{\{state\}\}">I</figure-callout>}}_{\mathrm{CS}}=\frac{{2I}_{\mathrm{in}}}{\mathrm{\&pi;N}}---\left(5\right)$

$F=\frac{\mathrm{\&omega;}}{{\mathrm{\&omega;}}_p}---\left(6\right)$

$Z_p=\sqrt{\frac{L_p}{C_p}}={\mathrm{\&omega;}}_p{L}_p=\frac{1}{{\mathrm{\&omega;}}_p{C}_p}---\left(7\right)$

$Q_L=\frac{R_L}{Z_p}---\left(8\right)$

$Q_e=\frac{R_e}{Z_p}---\left(9\right)$

$P_o=\frac{{V_o}^2}{R_L}=\mathrm{\&eta;}{P}_{\mathrm{in}}=\mathrm{\&eta;}{V}_{\mathrm{in}}{I}_{\mathrm{in}}---\left(10\right)$

By (5), (1), (4), (9) and (10) can draw the gain G of converter, shown in (11).The input phase angle λ of the resonant network of ac equivalent circuit is suc as formula shown in (12).

When the design resonant network, at first, according to formula (1), (2), the scope of operating frequency, resonance angular frequency ω are confirmed in (3) _pAnd the quality factor q of load _L,, promptly realize multi resonant vibrating network input current i so that when satisfying the pressure regulation requirement, realize the ZCS of square wave current source generator switching tube _PriSoft switching-over.Realize multi resonant vibrating network input current i _PriThe condition of soft switching-over be i _PriBe ahead of the voltage v at multi resonant vibrating network two ends _Rec, promptly λ is less than zero, and

Moment C _pThe energy of storage

Be greater than resonant inductance L _rThe required energy of electric current switching-over, shown in (14).

$G=\frac{V_o}{V_{\mathrm{in}}}=\mathrm{\&eta;}{R}_L\frac{I_{\mathrm{in}}}{V_o}=\mathrm{\&pi;\&eta;N}\sqrt{\frac{R_L{R}_e}{2}}\frac{{\mathrm{<figure-callout\; id="1"\; label="I\; CS"\; filenames="HSA00000192755900081.png"\; state="\{\{state\}\}">I</figure-callout>}}_{\mathrm{CS}}}{V_{\mathrm{rec}}}=\mathrm{\&pi;\&eta;N}\sqrt{\frac{R_{\mathrm{<figure-callout\; id="2"\; label="L\; R\; e"\; filenames="HSA00000192755900052.png,HSA00000192755900081.png"\; state="\{\{state\}\}">L</figure-callout>}}{R}_e{}_{}}{2}}\frac{1}{\left|Z_{\mathrm{in}}\right|}=\mathrm{\&pi;\&eta;N}\sqrt{\frac{Q_L}{{2\mathrm{<figure-callout\; id="1"\; label="Q\; e"\; filenames="HSA00000192755900081.png"\; state="\{\{state\}\}">Q</figure-callout>}}_e}}\sqrt{1+{\left[Q_e(\frac{1}{F}-F)\tan \left(\right|\mathrm{\&delta;}\left|\right)\right]}^2}---\left(11\right)$

$\mathrm{\&lambda;}={\tan }^{-1}(Q_e(\frac{1}{F}-F)-\tan \left(\right|\mathrm{\&delta;}\left|\right))---\left(12\right)$

λ＜0 (13)

$\frac{1}{2}{C}_p{v_{\mathrm{Cp}}}^2\left(\frac{\mathrm{\&alpha;}}{\mathrm{\&omega;}}\right)\&GreaterEqual;L_r{\left(\frac{I_{\mathrm{in}}}{2}\right)}^2---\left(14\right)$

Secondly, calculate minimum voltage gain G _Min, select transformer voltage ratio N.According to fixing ω _pAnd Q _L, through type (7) calculates L _pAnd C _PValue.Calculate L according to formula (14) then _rValue.Utilize resonant network to absorb the parasitic parameter of transformer.

(b) calculating of duty ratio:

According to L _r, C _PAnd the value of N, by Fig. 9 and formula (15), (16), (17), (18), (19), (20), (21), (22), (23) can obtain the restrictive condition of duty ratio.Wherein, D _MinAnd D _MaxBe maximum duty cycle and minimum duty cycle, Δ T _1,2With Δ T _1,4Be respectively t among Figure 11 ₁To t ₂Time period and t ₁To t ₄Time period, C ' _pBe C _pEquivalence is to the capacitance on the former limit of transformer, ω _rBe L _rWith C ' _pThe series resonance angular frequency, Z _rBe L _rWith C ' _pThe impedance of the series resonance network of forming.

$D_{\min }=\frac{{\mathrm{\&Delta;<figure-callout\; id="1"\; label="T"\; filenames="HSA00000192755900081.png"\; state="\{\{state\}\}">T</figure-callout>}}_{\mathrm{1,2}}}{T}+0.5---\left(15\right)$

$D_{\max }=\frac{{\mathrm{\&Delta;<figure-callout\; id="1"\; label="T"\; filenames="HSA00000192755900081.png"\; state="\{\{state\}\}">T</figure-callout>}}_{1,4}}{T}+0.5---\left(16\right)$

ΔT _1，2＝t ₂-t ₁ (17)

ΔT _1，4＝t ₄-t ₁?(18)

C′ _p＝N ²C _p (19)

${\mathrm{\&omega;}}_r=\frac{1}{\sqrt{L_r{C}_p^{\&prime;}}}---\left(20\right)$

$Z_r={\mathrm{\&omega;}}_r{L}_r=\frac{1}{{\mathrm{\&omega;}}_r{C}_p^{\&prime;}}---\left(21\right)$

${\mathrm{\&Delta;<figure-callout\; id="1"\; label="T"\; filenames="HSA00000192755900081.png"\; state="\{\{state\}\}">\; <figure-callout\; id="2"\; label="T"\; filenames="HSA00000192755900052.png,HSA00000192755900081.png"\; state="\{\{state\}\}">T</figure-callout>\; </figure-callout>}}_{\mathrm{1,2}}=\frac{1}{{\mathrm{\&omega;}}_r}{\sin }^{-1}\left(\frac{-4{\mathrm{NZ}}_r{I}_{\mathrm{in}}{v}_{\mathrm{Cp}}\left(\frac{\mathrm{\&alpha;}}{\mathrm{\&omega;}}\right)}{N^2{Z_r}^2{I_{\mathrm{in}}}^2+{{4v}_{\mathrm{Cp}}}^2\left(\frac{\mathrm{\&alpha;}}{\mathrm{\&omega;}}\right)}\right)---\left(22\right)$

${\mathrm{\&Delta;<figure-callout\; id="1"\; label="T"\; filenames="HSA00000192755900081.png"\; state="\{\{state\}\}">\; <figure-callout\; id="4"\; label="T"\; filenames="HSA00000192755900081.png"\; state="\{\{state\}\}">T</figure-callout>\; </figure-callout>}}_{\mathrm{1,4}}=\frac{\mathrm{\&pi;}}{{\mathrm{\&omega;}}_r}---\left(23\right)$

(c) calculating of input inductance:

When switching tube conducting simultaneously, input current i _InLinear increase, i in all the other times _InLinearity reduces.Therefore the frequency of input current ripple is the twice of switching frequency, and the amplitude of input current ripple can be calculated by formula (24), wherein, and Δ i _InBe the peak-to-peak value of input current ripple, Δ T _OvIt is the time of former limit switching tube conducting simultaneously.

${\mathrm{\&Delta;i}}_{\mathrm{in}}=2\frac{V_{\mathrm{in}}}{L}{\mathrm{\&Delta;T}}_{\mathrm{ov}}---\left(24\right)$

Input current ripple value according to selected can calculate input inductance L by formula (24) _In1And L _In2Value L.

(3) further combine Figure 10, Figure 11, the shake operation principle of DC converter of typical current type multi resonant shown in Figure 1 is described.

Wherein, Figure 10 (a) to (f) is six working mode figures of current-type multi-resonance direct current (DC) converter in preceding half period, six working mode figures and the preceding half period symmetry of this converter in the half period of back, and Figure 11 is corresponding circuit working oscillogram.

At t ₀Constantly, Q ₁Conducting, Q ₂Turn-off.Input power supply V _InGive input inductance L _In1Charging, input inductance L _In2Energy stored is through rectifier diode D ₂And D ₃Pass to load.The parallel resonance capacitor C _pVoltage v _CpBe clamped at negative output voltage V _o, parallel resonant inductor L _pLinear increase of current reversal.Flow through rectifier diode D ₂And D ₃Electric current equal transformer secondary current i _SecDeduct the L that flows through _pCurrent i _Lp

Pattern 1 (t ₀-t ₁): at t ₀Constantly, i _LpEqual i _Sec, flow through D ₂And D ₃Electric current drop to zero.After this Q ₁Continue conducting, Q ₂Keep turn-offing, all rectifier diodes turn-off, V _InContinue to give L _In1Charging, the series resonance inductance L _rBe transfused to inductance L _In2Absorb, flow through L _In2Electric current flow into by L _pAnd C _pThe resonant network that constitutes, during pattern 1, C _pDischarge, its energy storage reduces.At t ₁Constantly, Q ₂Open-minded, pattern 1 finishes.

Pattern 2 (t ₁-t ₂): from t ₁Constantly begin Q ₁, Q ₂Conducting simultaneously, 2 of A, B are by short circuit, and all rectifier diodes keep turn-offing the series resonance inductance L _rParticipate in L _pAnd C _pResonance, capacitor C _pEnergy stored makes primary current i _PriBegin switching-over, flow through Q this moment ₂Current i _Q2Increase, flow through Q ₁Electric current t _Q1Reduce.At t ₂Constantly, i _Q1Be reduced to zero, pattern 2 finishes.Design suitable L _rValue is vital, and this can guarantee overlapping in the switching tube conducting of former limit i in the time _Q1Be reduced to zero and have only very little overshoot, make Q ₁Under the prerequisite of minimum circulation, realize zero-current switching.During pattern 2, V _InGive L simultaneously _In1And L _In2Charging, C _pVoltage continue to reduce.

Mode 3 (t ₂-t ₃): mode 3 and pattern 2 are similar, just i _Q1In the opposite direction, L _rContinue to participate in L _pAnd C _pResonance, work as i _PriResonance is during to peak value, C _pVoltage be zero, flow through L _pElectric current also reach peak value.At t ₃Moment Q ₁Turn-off, mode 3 finishes.During mode 3, C _pVoltage v _CpBegin after the zero passage to increase, but v _CpValue still less than v _o

Pattern 4 (t ₃-t ₄): pattern 4 is similar with mode 3, just t ₃Moment Q ₁Have no progeny in the pass, i _Q1Flow through Q ₁The inverse parallel diode.At t ₄Constantly, t _Q1Arrive zero through reversed peak resonance, Q ₁Inverse parallel diode D _Q1Turn-off, pattern 4 finishes.Through rational control Q ₁Turn-off time can reduce the inverse parallel diode current flow time, thereby reduce inverse parallel diode on-state loss.

Pattern 5 (t ₄-t ₅): from t ₄Constantly begin D _Q1Turn-off Q ₂Continue conducting, L _rBy big inductance L _In1Absorb, flow through L _In1Electric current flow into by L _pAnd C _pThe resonant network that constitutes, V _InGive L _In2Charging.During this period, C _pVoltage continues to increase, up to t ₅Constantly, v _CpEqual v _o, pattern 5 finishes.

Pattern 6 (t ₅-t ₆): t ₅Constantly, v _CpEqual v _o, rectifier diode D ₁And D ₄Open-minded, L _In1Energy stored is through diode D ₁And D ₄Pass to load, V _InGive L _In2Charging, V _oGive L _pCharging.Flow through D ₁And D ₄Electric current equal the current i that the transformer secondary flows out _SecWith flow through L _pCurrent i _LpDifference, at t ₆Constantly, i _LpEqual i _Sec, flow through D ₁And D ₄Electric current drop to zero, pattern 6 finishes.

Shown in figure 12, switching tube is operated in the ZCS state, and switching tube has less due to voltage spikes and current over pulse.v _ABThe voltage spine mainly be by switching tube Q ₁And Q ₂Inverse parallel diode D _Q1And D _Q2Reverse recovery cause.

Shown in figure 13; Under the underloading condition; The rectifier diode ON time shortens, and parallel resonance electric capacity is also reduced by the time of clamper relatively, and the time of parallel resonant inductor and parallel resonance capacitor resonance can be elongated; Make the initial energy storage of parallel resonance electric capacity and series resonance inductance resonance be able to reduce, thus when underloading is worked i _PriLess overshoot is arranged, and the light-load efficiency of converter does not have too big influence.

Shown in Figure 14 and 15, under full load and underloading condition, rectifier diode all is operated under the ZCS condition.Under the underloading condition, the ON time of rectifier diode and v _CpThe clamped time can obviously reduce, make the initial energy storage of parallel resonance electric capacity and series resonance inductance resonance be able to reduce, thereby can limit converter i when underloading is worked _PriOvershoot.

Shown in Figure 16 is the waveform when not having input filter capacitor, visible by figure, i _InLess ripple is arranged, and its frequency is the twice of switching frequency.

Visible by Figure 17, the light-load efficiency of converter is not serious to be reduced.

In sum, converter of the present invention can provide higher voltage gain, in full-load range, all has less circulation, and switching tube and rectifier diode all are operated in the ZCS state, and underloading also has higher efficient when working.

Claims (3)

1. current-type multi-resonance direct current (DC) converter; The square wave current source generator, multi resonant vibrating network, the rectifying and wave-filtering output unit that comprise series connection successively; Said multi resonant vibrating network comprises transformer, parallel resonant inductor, parallel resonance electric capacity and series resonance inductance; Parallel resonant inductor, parallel resonance electric capacity are all parallelly connected with the transformer secondary, and said series resonance inductance is connected the former limit of transformer; Said rectifying and wave-filtering output unit comprises diode rectifier circuit and the filter capacitor that is connected in parallel on the diode rectifier circuit output; It is characterized in that: the square wave current source generator comprises DC power supply, low-voltage filter electric capacity, first switching branches and second switch branch road; Low-voltage filter electric capacity, first switching branches and second switch branch road all are connected in parallel on the DC power supply two ends; First switching branches is formed by first inductance and the series connection of first switching tube; Second switch props up route second inductance and second switching tube series connection forms; The series connection contact of first inductance and first switching tube, the series connection contact of second inductance and second switch pipe is drawn the positive-negative output end as the square wave current source generator respectively; Break-make through controlling first switching tube and second switch pipe forms square wave current.

2. current-type multi-resonance direct current (DC) converter according to claim 1 is characterized in that, the switching tube of said square wave current source generator open duty ratio greater than 0.5, less than 0.7.

3. current-type multi-resonance direct current (DC) converter according to claim 1 and 2 is characterized in that, said diode rectifier circuit adopts full bridge rectifier, full-wave rectifying circuit or voltage doubling rectifing circuit.

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