CA1177759A - Circuit arrangements for modifying dynamic range - Google Patents

Abstract

ABSTRACT
Circuit arrangements for cross-coupling series connected bi-linear compressors and expanders are disclosed including the coupling of signal path signal components in one circuit to a signal path in another circuit, the coupling of control circuit signal components in one circuit to a control circuit in another circuit and cross-coupling by way of employ-ing a common control circuit for series connected devices.

Description

i 1~7759 11 1¦ Inventor: RAY M. DOLBY
21 Title: IMPROVEMENTS IN CIRCUIT ARRANGEMENTS
31 FOR MODI~YING DYNAMIC RANGE

4¦ The present invention is concerned in general with 51 circuit arrangements which alter the dynamic range of audio and 61 other signals, namely compressors which compress the dynamic 71 range and expanders which expand the dynamic range. Generally, 81 it relates to improvements in compressors and expanders that 91 comprise series connected circuits and more particularly to 10¦ the cross-coupling of such series circuits.
11¦ Compressors and complementary expanders are often used 12¦ together (a compander system) to effect noise reduction; the 13¦ signal is compressed before transmission or recording and 14¦ expanded after reception or playback from the transmission 15 ¦ channel. However compressors may be used alone to reduce the 16 ¦ dynamic range, e.g. to suit the capacity of a transmission 17 ~ channel, without subsequent expansion when the compressed signal 18 is adequate for the end purpose. In addition, compressors alone 19 are used in certain products, especially audio products which are intended only to transmit or record compressed broadcasts or 21 pre-recorded signals. Expanders alone are used in certain 22 . products, especially audio products which are intended only to 23 I receive or play back already compressed broadcasts or pre-recorded 24 signals. In certain products, particularly audio recording and 25 I play back products, a single device is often configured for 26 switchable mode operation as a compressor to record signals and 27 as an expander to play back compressed broadcasts or pre-recorded 28 ~ signals.
29 The amount of compression or expansion may be expressed in dB. For example, 10 dB of compression means that an input 31 dynamic range of N dB is compressed to an output range of (N-10) 32 dB. In a noise reduction system 10 dB of compression followed 1 177~59 1 b 10 dB o~ complementary expansion is said to provide 10 ~B oE
21 noise reduction.
31 The present invention relates in particular to 41 improvements in circuit arrangements for modifying the dynamic 51 range of an input signal, which circuit arrangements comprise 6~ series circuits, each with a bi-linear characteristic ~where 71 "linear" in this context denotes constant gain) composed of:
81 1) a low level linear portion up to a threshold, 91 2) an intermediate level non-linear (changing gain) 10¦ portion, above the threshold and up to a finishing 11¦ point, providing a predetermined maximum compression 12¦ ratio or expansion ratio, and 13¦ 3) a high level linear portion having a gain 14 ¦ different from the gain of the low level portion.
15 ¦ The characteristic is denoted a bi-linear characteristic 16 ~ because there are two portions of substantially constant gain.
17 In practice, the threshold and finishing point are 18l not always well defined "points". The two transition regions 19 where the intermediate level portion merges into the low level and high level linear portions can each vary in shape 21 from a smooth curve to a sharp curve, depending on the control 22 characteristics of the compressor and expander.
23 It is also pointed out that circuit arrangements with 24 bi-linear characteristics are distinguished from two other known classes of circuit arrangement, namely:
26 (a) a logarithmic or non-linear circuit arrangement 27 with either a fixed or changing slope and with no linear portion:
28 the gain changes over the whole dynamic range.
29 (b) circuit arrangements with a characteristic having two or more portions of which only one portion is linear 31~ ("uni-linear").
32 ~

-2-al7~7~s 1 A circuit arrangement with a bi-linear charaeteristic 2 has particular advantages and is widely used. The threshold can

3 be set above the input noise level or transmission channel noise

4 level in order to exclude the possibility of eontrol of the circuit by noise. The high level portion of substantially 6 constant gain avoids non-linear treatment of high level signals 7 which would otherwise introduce distortion. Moreover, in the 8 case of an audio signal, for which the circuit must be syllabic, 9 the high level portion provides a region within whieh to deal with the overshoots whieh oeeur with a syllabie eireuit when the 11 signal level inereases abruptly. The overshoots are suppressed 12 by elipping diodes or similar means. Only bi-linear eharaeter-13 istics are capable of providing this combination of advantages.
14 Known circuits employing a single stage with a bi-linear charaeteristie in use today in eonsumer audio products 16 provide 10 dB of compression and expansion, which is adequate 17 for many purposes. However, this leaves some noise audible to 181 some listeners and, for highest fidelity, more compression and 19 expansion is desirable, say 20 dB. A new circuit now also in use for eonsumer audio products is described in Belgian-PS
21 889,428, Belgian-PS 889,427, Belgian-PS 889,426, Audio, May, 22 , 1981, pp. 20~26 and in paper J-6 and preprint presented at 23 November, 1981 Convention, Audio Engineering Soeiety, New York, 24 New York.
Prior to the above-mentioned circuits, circuits were 26 known and commercially available which provided 20 dB of compres-27 sion or expansion, and even more, but these were usually constant 28¦ slope logarithmic circuit arrangements in which there is a 29~ constantly changing gain over the whole dynamic range or nearly 30¦ the whole dynamic range. Such circuits suffer from higher 31¦ distortion and signal tracking problems at very low and very 321 high signal levels than the bi-linear circuits in which the , _3_ 1 177~9 l change of gain is restricted to an intermediate portion of the 21 characteristic and overshoot problems are more severe than with 31 bi-linear characteristic arrangements. Known constant slope 41 companders employ compression ratios in the range 1.5:1, 2:1 and 51 3 1~ but 2:1 is most common.
61 ~ccording to the staggering aspect of bi-linear 71 circuits described in Belgian-PS 889,428, a first circuit, which 81 has a bi-linear input-output characteristic, is followed by one 91 or more further circuits which also have bi-linear character-lO¦ istics at any given frequency within a frequency range common to ll¦ the circuits. The thresholds and dynamic regions of the circuits 12¦ are set to different values so as to stagger the intermediate 13¦ level portions of the characteristics of the circuits to produce 14¦ a change of gain over a wider range of intermediate input levels 15¦ than for any of the circuits individually, and to produce an 16 ¦ increased difference between the gains at low and high input 17~ levels, but with a maximum compression or expansion ratio which ;3 is substantially no greater than the maximum compression ratio 19 of any single circuit, by virtue of the staggering.
In the case of audio circuits, if the circuits have 21 overshoot suppression (limiting) elements, then it is also 22 possible to stagger their thresholds along with the stagger of 23 I the syllabic thresholds. The overshoots of the lower level 24 circuits, or stages, are correspondingly reduced, with minimal overall overshoot of the several stages. This is in contrast 26 to conventional logarithmic compressors in which large over-27 shoots are inherently produced.
28 Each of the circuits may introduce an alteration of 29 the spectral content of the signal -- for example, a low level treble boost in the case of a compressor. Thus each succeeding 31 stage may be actuated by a signal of progressively changing 32 spectral content. In the case of complex signals, this has the Il -4-1.

11777~9 1 virtue of spectrally spreading out the chances for error in the 21 decoding function. In the case of a tape recorder with an 31 uneven frequency response characteristic, for example, the 41 spectral shifting tendency reduces the overall dynamic and frequency response errors of the decoded result.
6 The ability to stagger bi-linear stages provides the 7 designer with an additional way in which to optimize an overall 8 circuit. In so doing, the shapes of the compression character-9 istics of individual stages can be designed with staggering specifically in mind. The transient characteristics of the 11 circuits are also taken into account and the opportunity is 12 preferably taken to stagger the overshoot suppression thresholds 13 in audio compressors and expanders so as to result in minimal 14 overall overshoot.
A well known type of circuit, called "sliding band", 16 which can be used for each of the first and second circuits, 17 creates the specified desirable characteristic for the case of 18 high frequency audio compression or expansion by applying high 19 frequency boost (for compression) or cut (for expansion) by way of a high pass filter with a variable lower corner frequency.
21 As the signal level in the high frequency band increases, the 22 I filter corner frequency slides upwardly so as to narrow the 23 boosted or cut band and exclude the useful signal from the boost 24 or cut. Examples of such circuits are to be found in US-PS Re 28,426, US-PS 3,757,254, US-PS 4,072,914, US-PS 3,934,190 and 26 ~ V~S~ Pc~t~nt /~o,~ql 1,~`7 J ~
27 Accordingly, each of the first and second circuits can 28 be such a "sliding band" circuit. In principle the quiescent 29 corner frequencies of the two sliding band circuits can be different and use can be made of this to provide a degree of 31 compression or expansion which is higher in one part of the 32~

1 1777~9 1 ~ treated frequency band than in another. However, in a modifica- ¦
2 ¦ tion the corner frequencies are made substantially identical.
3 ¦ This leads to the advantage of sharper discriminations between 4 ¦ the frequency region where boost or cut is being applied and the

5 ¦ reqion where it is not applied and accordingly a sharper discrimi-

6 1 nation between the region where noise reduction is no longer

7 ¦ taking place, because of the appearance of a significant useful

8 ¦ signal, and the region where noise reduction remains effective.
91 On the other hand, circuits are also well known in which the frequency spectrum is split into a plurality of bands 11¦ by corresponding band-pass filters and the compression or expan-12¦ sion is effected in each band by a gain control device (whether 13¦ an automatically responsive, diode type of limiting device or a 14¦ controlled limiting device) in the case of a compressor, with 15¦ some form of reciprocal or complementary circuitry for an 16¦ expander. Examples of such circuits are to be found in US-PS
17¦ 3,846,719. These split band or multi-band circuits have the 181 advantage of independent action in the various frequency bands 19 and, if this property is required, such circuits may be employed as the first, second, or more stages in the series arrangements.
21 It is known to construct bi-linear compressors and 22 expanders, of both sliding band and split band type, by the use 23 of only a single signal path. However, it is generally preferred 24 to construct such devices by providing a main signal circuit which is linear with respect to dynamic range, with a combining 26 circuit in the main circuit, and a further circuit which derives 27 ! its input from the input or output of the further circuit and 28~ has its output coupled to the combining circuit. The further 291 circuit includes a limiter (self-acting or controlled) and the 30~ limited further circuit signal boosts the main circuit signal in 31¦ the combining circuit for the case of compression but bucks the 32~ main circuit signal for the case of expansion. The limited ~ 17775g 1 further path signal is smaller than the main path signal in the 21 upper part of the input dynamic range. The main and further 3 ~ circuits are preferably and most conveniently separately 41 identifiable signal paths.
Such known compressors and expanders are particularly 6¦ advantageous because they enable the desired kind of transfer 7~ characteristic to be established in a precise way without 81 problems of high-level distortion. The low level portion of 91 substantially constant gain is established by giving the further 10¦ path a threshold above the noise level; below this threshold the 11¦ further path is linear. The intermediate level portion is 12 ¦ created by the region over which the further path limiting 13 ¦ action becomes partially effective and the high level portion of 14¦ substantially constant gain arises after the limiter has become 15 ¦ fully effective so that the further path signal ceases to 16 ~ increase and becomes negligible compared to the main path 17~ signal. At the highest part of the input dynamic range, the 18 output of the circuit arrangement is effectively only the signal 19 passed by the linear main path, i.e. linear with respect to dynamic range. In dual path audio circuits the provision of 21 overshoot suppression is particularly convenient.
22 Examples of these known circuits are to be found in 23 US-PS 3,846,719, US-PS 3,9n3,485 and US-PS Re 28,426. There are 24 also known analogous circuits which achieve like results but 25 wherein the further path has characteristics inverse to limiter 26 characteristics and the further path output bucks the main path 27 signal for compression and boosts the main path signal for 28 expansion (US-PS 3,828, 2ao and US-PS 3,875,537).
29 Any of these known bi-linear circuits may accordingly be employed as the first and second circuits of the series 311 circuit arrangements embodying the invention.

1~777~9 1 ¦ As mentioned previously, it is not essential to create 2 ¦ the desired form of bi-linear characteristic by such "dual path"
3 ¦ techniaues. Alternatives exist, operating with single paths, as 4 1 described in US-P~ 3,757,254, US-P5 3,967,219, US-PS 4,072,914, 5 ¦ US-PS 3,909,733 and ~F_Ke~e~PatentA ~ , for 6~ example. Although these alternative circuits usually are not 71 capable of producing such good results as dual path circuits, or 81 may be less convenient and thereby less economical, they can

9 ¦ produce generally equivalent results. Accordingly, these known

10¦ circuits can also be used as one or more of the circuits of a

11¦ series circuit arrangement embodying the invention. If desired,

12¦ one of the first and second circuits can be a dual path circuit

13¦ and the other a single path circuit.

14¦ In the above described staggered series bi-linear

15¦ circuit arrangements, the series processors operate independently

16 ¦ of each other. In accordance with the teachings of the present

17¦ invention, means for cross-coupling staggered series bi-linear

18¦ circuits are provided, including the coupling of signal path

19 ¦ signal components in one circuit to a signal path in another

20 ¦ circuitl the coupling of control circuit signal components in

21 ¦ one circuit to a control circuit in another circuit and cross-

22 ¦ coupling by way of employing a common control circuit for series

23 ¦ connected devices. Cross-coupling may a) assist in immunizing

24 ¦ the system to control by undesired signals, b) help reduce noise

25 1 modulation effects, c) suppress spurious responses, d) allow

26 ¦ relaxation of individual circuit requirements, e) increase

27 compression or expansion without side effects, f) reduce overall

28 circuit complexity and cost, etc.

29 ¦ A characteristic of staggered series bi-linear circuits

30¦ is that they employ different operating thresholds and, usually,

31 different overshoot thresholds (at least in the case of audio

32 devices~. Consequently, the c rcuits respond differently.

l 1~775~
Cross-coupling of AC signals between (or among) the series circuits provides the system designer with an additional design parameter that may be useful in optimizing the operation of the system.
For example, in the case of dual path bi-linear cir-cuits, the signal output of the noise reduction path of a higher threshold level circuit ean be fed to the lower threshold level circuit and injected, with suitable frequeney response and phase modifications, into the fixed and variable filter signal eireuits, thereby to ereate aetive filter aetions that enhance the sliding band aetion and subsequent noise reduetion effeet.
Similarly, a eross-eoupling ean be effeeted between the eontrol signals in series bi-linear circuits to improve the response to rapidly changing signal amplitudes. In one embodi-ment, a control signal component from the low level eircuit (which has the first and largest response to incoming trans-ients, even though it is preferably the second device, because its threshold is reached first when an input signal rises in level) is fed via a coupling network to the eontrol cireuit of the high level proeessor. In this way, the high level proees-sor is provided with a timely warning of impending signal amp-litude ehanges. Sueh control signal cross-coupling is particu-larly useful in minimizing overshoots and distortion in series bi-linear audio cireuits.
In addition, eircuit complexity and cost can be re-dueed by the use of a single control circuit for two or more stages of series bi-linear circuits, in whieh ease the respee-tive inputs and outputs of the single eontrol eireuit are eross-eoupled between (or among) the series stages.
In summary the present invention provides a eircuit for modifying the dynamie range of an input signal comprising:

_g_ ~7'77S9 first circuit means having a characterlstic in which only a portion thereof has a changing gain between low and high input levels, at least one second circuit means which also has such a characteristic within a frequency range common to the circuit means, in series with the first circuit means, the changing gain portions of the characteristics of the circuit means being staggered within the frequency range common to the circuit means such as to provide a change of gain over a wider range of input levels than for any one of the circuit means individu-ally, and an increased difference between the gains at lowand high input levels, but with a maximum overall compression or expansion ratio which is substantially no greater than that of any single circuit means, by virtue of the staggering, and at least one coupling circuit means coupling signal components from one of the circuit means to another of the circuit means for modifying the action of the said other circuit means in response to coupled signal components from said one circui-t means.
The invention will be described in more detail, by way of example, with reference to the accompanying drawings, in which:
Figure 1 is an exemplary set of curves showing comple-mentary bi-linear compression and expansion characteristics.

-9a-1177~9 l Figure 2 i5 a block diagram showing series bi-linear 2 ¦ devices in general terms.
3 ~ Figure 3 is a schematic circuit diagram of a prior art 41 sliding band compressor.
51 Figure 4 is a schematic circuit diagram of a prior art 6 ¦ sliding band expander.
71 Figure 5 is a schematic circuit diagram of a modifi-81 cation to Figures 3 and 4.
9¦ Figure 6 is a block diagram of a dual path bi-linear lO¦ sliding band compressor such as that described in connection ll¦ with Figure 3 or Figure 3 with the modification of Figure 5.
12 ¦ Figures 7 and 8 are block diagrams showing a prior 13¦ art fixed band compressor and expander.
14¦ Figure 9 is a block diagram showing the present lS¦ invention in general terms.
16 ¦ Figure 10 is a block diagram showing the present 17¦ invention embodied in a dual stage bi-linear compressor and 18¦ expander.
19¦ Figure 11 is a more detailed block diagram of the 20 ¦ embodiment of Figure 10.
21 ¦ Figure 12 is a schematic diagram showing an exemplary 22 ¦ cross coupling network for use in the embodiment of Figure 11.
23 ¦ Figure 13 is a block diagram showing a further 24 embodiment of the invention.
25 ¦ Figure 14 is a schematic diagram showing an exemplary 26 ¦ cross-coupling network for use in the embodiment of Figure 13.
27 ¦ Figure 15 is a block diagram showing the invention 2~ ¦ embodied in an arrangement for providing a common control 29 circuit to series connected sliding band bi-linear devices.
Figure 16 is a block diagram showing the invention 311¦ embodied in an arrangement for providing a common control 32 circuit to series connected fixed band bi-linear devices.

1,' 1 Exemplary bi-linear complementary compression and 2 expansion transfer characteristics (at a particular frequency) 3 are shown in Figure 1, indicating (for the compression charac~
4 teristic) the low level portion of substantially constant gain, the threshold, the portion where dynamic action occurs, 6 the finishing point, and the high level portion of substantially 7 constant gain.
8 Figure 2 shows series bi-linear devices in general 9 terms: a first bi-linear compressor 2 receives the input infor-mation and applies its output to a second bi-linear compressor 4 11 connected in series, which has its output applied to a noisy 12 information carrying channel N. A pair of series connected 13 bi-linear expanders 6 and 8 receive the input from channel N at 14 expander 6 and provide a noise reduction system output at the output of expander 8. The areas of dynamic action of the series 16 devices are separated or staggered with respect to each other 17 within the freauency range that is common to the devices.
18 Although the figure shows two devices on each side of the 19 information channel N, two or more can be employed: the inven-tion contemplates the cross-coupling of two or more series 21 bi-linear compressors or expanders as explained further herein-22 after. When configured as a complementary noise reduction 23 system, like numbers of series bi-linear compressors and expanders 24 are provided.
25 ¦ The order of stages having particular characteristics 26 ¦ in the compressor is reversed in the expander. For example, 27 ¦ the last staqe of the expander is complementary to the first 28 ~ stage of the compressor in all respects--steady state and time 29 dependent dynamic response (frequency, phase and transient 30 I response under all signal level and dynamic conditions).
31 ¦ As mentioned earlier, it is usually preferable for 32 the hiqh level stage to be first in a compressor series and the I
l l 117775g 1 low-level stage to be last. However, a reversed arrangement 2 ¦ is also possible. In the reversed case the control amplifier 3 ¦ of the first stage needs a high gain in order to achieve the 4 ¦ required low threshold. This low threshold then applies even in the presence of high level signals, which in the case of sliding 61 band systems ~nown in the prior art usually leads to poor noise 71 modulation performance of the overall system. In this reversed 81 arrangement each stage must provide sufficient control amplifier 91 gain to achieve the threshold required of that stage. Moreover, 10¦ each threshold is essentially fixed and independent of the opera- ¦
11¦ tion of the other staaes. This is a consequence of the fact that 12 the signal gain of each earlier stage has fallen substantially to 13 unity when the threshold is reached for the corresponding suc-14 ceeding stage.
In contrast to the reversed situation, in the preferred 16 arrangement (in which the high level stage is first in the 17 compressor chain, and the low-level stage is last), there is a 18 useful interaction between the stage gains and the thresholds.
19 The thresholds of the downstream stages are partly determined by the signal gains of the preceding stages. Thus in a 2 stage 21 system with 10 dB of low level gain per stage, the control ampli- ¦
22 fier gain requirement of the second stage is reduced by 10 dB, by 23 virtue of the low-level siqnal gain of the first stage. When a 24 high level sianal appears, the 10 ds gain of the first stage is eliminated and the threshold of the low level stage is effectively 26 raised by 10 dB. With sliding band companders this improves the 27 noise modulation performance of the noise reduction action.
28 In the preferred arrangement the qains of all preceding 29 stages are fully effective up to the threshold of any particular succeeding stage. Thus, in contrast with the reversed order 31 system described above, the preferred arrangement takes best 11777~9 1 advantage of the prevailing signal gains of the individual 2 ¦ staqes. Namely:
3 ¦ 1. Under very low level (sub-threshold) signal 4 ¦ conditions the control amplifier gain requirement 5 ¦ of each staqe is reduced by an amount equal to the 61 cumulative signal gains of all preceding stages.
7¦ 2. A signal dependent variable threshold effect is 81 achieved, whereby with sliding band stages, noise 91 modulation effects are reduced. The effective 10¦ thresholds of the low level stages are progressively 11¦ raised with increasing signal level at a particular 12¦ frequency. At high signal levels (on the hiqh level 13¦ linear portion of the transfer characteristic) the 14¦ effective threshold of the lowest level stage is 15¦ raised by a level equal to all the low-level (sub-16¦ threshold) stage gains up to that point.

1~¦ One known embodiment of series bi-linear processors 19 employs series sliding band devices: the compressors 2 and 4 20 ¦ and the expanders 6 and 8 of Figure 2 are sliding band devices 21 ¦ as set forth in US-PS Re 28,426 with modifications thereof as 22 described in Belgian-PS 889,428. Such modifications include 23 staggering the syllabic and overshoot thresholds and changing 24 the filter corner frequencies.
Details of the basic circuit are set forth in Figures 26 3, 4 and 5 which are the same as Figure 4, 5 and 10 respectively 27 of US-PS Re 28,426 and further details of said circuits, their 28 operation and theory are set forth therein. The following 29 ~ description of Figures 3, 4 and 5 is taken in large part from 30 ¦ US-PS Re 28,426.
31 ¦ The circuit of Figure 3 is specifically designed for 32 incorporation in the record channel of a consumer tape recorder, j -13-l ¦ two such circuits being required for a stereo recorder. The 21 input signal is applied at terminal 10 to an emitter follower 3 ¦ stage 12 which provides a low impedance signal. This signal is 4¦ applied firstly through a main, straight-through path constituted 51 by a resistor 14 to an output terminal 16 and secondly through a 61 further path the last element of which is a resistor 18 also 71 connected to the terminal 16 The resistors 14 and 18 add the 8¦ outputs of the main and further paths to provide the reauired 91 compression law.
lO¦ The further path consists of a fixed filter 20, a ll¦ variable cut-off filter 22 including a FET 24 (these constituting ¦
12¦ the filter/limiter), and an amplifier 26 the output of which is 13 ¦ coupled to a double diode limiter or clipper 28 and to the 14¦ resistor 18. The non-linear limiter suppresses overshoots of 15 ¦ the output signal with abruptly increasing input signals. The 16¦ amplifier 26 increases the signal in the further path to a level 17¦ such that the knee in the characteristic of the limiter or 18¦ overshoot suppressor 28, comprising silicon diodes, is effective 19 at the appropriate signal level under transient conditions.
20 ¦ The effective threshold of the overshoot suppressor is somewhat 21 ¦ above that of the syllabic filter/limiter. The resistors 14 and 22 ¦ 18 are so proportioned that the required compensating degree of 23 ¦ attenuation is then provided for the signal in the further path. I
241 The output of the amplifier 26 is also coupled to an ~ -25 ¦ amplifier 30 the output of which is rectified by a germanium 26 ¦ diode 31 and integrated by a smoothing filter to provide the 27 ¦ control voltaqe for the FET 24.
28 Two simple RC filters are used, though equivalent 29 LC or LCR filters could be used. The fixed filter 20 provides a cut-off freguency of 1700 Hz (now 1500 Hz), below which dimishing 31 ¦I compression take place. The filter 22 comprises a series 32 1l ///

11777~9 1 ¦ capacitor 34 and shunt resistor 36 followed by a series resistor 2 ~ 38 and the FET 24, with its source-drain path connected as a 3 shunt resistorO Under quiescent conditions with zero signal on 41 the gate of the FET 24, the FET is pinched off and presents 51 substantially infinite impedance; the presence of the resistor 61 38 can then be ignored. The cut-off frequency of the filter 22 71 is thus 800 Hz (now 750 ~z), which it will be noted is substan-81 tially below the cut-off frequency of the fixed filter 20.
91 When the signal on the gate increases sufficiently for 10¦ the resistance of the FET to fall to less than say 1 K, the 11¦ resistor 38 effectively shunts the resistor 36 and the cut-off 12 ¦ frequency rises, markedly narrowing the pass band of the filter.
13¦ The rise in cut-off fre~uency is of course a progressive action.
14¦ The use of a FET is convenient because, within a 15¦ suitable restricted range of signal amplitudes, such a device 16¦ acts substantially as a linear resistor (for either polarity 17¦ signal), the value of which is determined by the control voltage 18¦ on the gate. - ¦
19~ The resistor 36 and FET are returned to an adjustable 20 ¦ tap 46 in a potential divider which includes a temperature 21 ¦ compensating germanium diode 48. The tap 46 enables the compres-22¦ sion threshold of the filter 22 to be adjusted.
23 ¦ The amplifier 26 comprises complementary transistors 24 ¦ giving high input impedance and low output impedance. Since 25 the amplifier drives the diode limiter 28, a finite output 26 impedance is required and is provided by a coupling resistor 27 1 50. The diodes 28 are, as already noted, silicon diodes 28 ~ and have a sharp knee around 1/2 volt.
29 The signal on the limiter and hence on the resistor 18 can be shorted to ground by a switch 53 when it is required to 31 ~ switch the compressor out of action.
32 ~ ///

l The amplifier 30 is an NPN resistor with an emitter 21 time constant network 52 giving increased gain at high fre-31 quencie~. Strong hiah frequencies (e.g. a cymbal crash) 4 will therefore lead to rapid narrowing of the band in which compression takes place, so as to avoid signal distortion.
6 The amplifier is coupled to the smoothing filter 32 7 through the rectifying diode 31. The filter comprises a series 8 resistor 54 and shunt capacitor 56. The resistor 54 is shunted 91 by a silicon diode 58 which allows rapid charging of the lO¦ capacitor 56 for fast attack, coupled with good smoothing under ll¦ steady-state conditions. The voltage on capacitor 56 is applied 12¦ directly to the gate of the FET 24.
13 A complete circuit diagram of the complementary 14 expander is provided in Figure 4, but a full description is not re~uired as substantially as the circuit is identical to Figure 16 3, component values, are therefore not for the most part shown 17 in Figure 4.
18 The differences between Figures 3 and 4 are as follows: ¦
19 In Figure 4, the further path derives its input from the output terminal 16a, the amplifier 26a is inverting, 21 and the signals combined by the resistors 14 and 18 are applied 22 to the input (base) of the emitter follower 12, the output 23 (emitter) of which is coupled to the terminal 16a. To ensure 24 low driving impedance, the input terminal 1Oa is coupled to the resistor 14 through an emitter follower 60. Suitable measures 26 must be taken to prevent bias getting in the expander.
27 The amplifier 26a is rendered inverting by taking the 28 output from the emitter, instead of the collector, of the second 29 (PNP) transistor. This alteration involves shifting the 10 K
resistor 62 (Figure 3) from the collector to the emitter (Figure 31 3), which automatically gives a suitable output impedance for 1177~59 l drivinq the limiter. The resistor 50 is therefore omitted in 2 Figure 4.
3 It should be noted that it is important in aligning 4 a complete noise reduction system to have equal signal levels on the emitters of the transistors 12 in both compressor 6 and expander. Metering terminals M are shown connected to 7 these emitters.
8 Figure 5 shows a preferred circuit, for replacing the 9 circuit between points A, B and C in Figures 3 and 4. When the lO ¦ FET 24 is pinched off, the second RC network 22 is inoperative, ll¦ and the first RC network 20 then determines the response of the 12 ¦ further path. The improved circuit combines the phase advantages 13¦ of having only a single RC section under quiescent conditions 14 with the 12 dB per octave attenuation characteristics of a 15¦ two-section RC filter under signal conditions.
,61 In the practical circuit, using MPF 104 FETIs, the 39 K ¦
17¦ resistor 36a is necessary in order to provide a finite source 18¦ impedance to work into the FET. In this way the compression 19¦ ratio at all frequencies and levels is held to a maximum of 20 ¦ about 2. The 39 K resistor 36a serves the same compression 21 ¦ ratio limiting function in the improved circuit as the resistor 22 ¦ 36 in the circuit of Figure 6 or Figure 7. In addition, this 231 resistor provides a low frequency path for the signal.
241 Certain details of the circuit of Figures 3, 4 and 5 25 ¦ have evolved over the years and more modern forms of the circuit 26 have been published and are well known in the art. Reference to 27 I the specific circuit in US-PS Re 28,426 is made for convenience 28 ¦ in presentation.
29 In Belgian-PS 889,428 modifications to the cir-cuit just described are disclosed, particularly for the purpose 31 I of operating two such circuits in series. These modifications l ll 117775~

1 ¦ include changing the frequencies of filters 20 and 22, the 2 1 changin~ of the overshoot suppression levels, and changin~ the 3 ¦ syllabic threshold of one of the circuits by modifying the 41 control amplifier 30. This is done by altering its pre-emphasis 51 characteristics which are controlled by emitter time constant 61 network 52. Increasing the capacitor value in the emitter 71 network of control amplifier 30 increases the amplifier gain at 8 any given frequency, thus causing the sliding band filter to 9 respond to lower signal levels. As explained above and and in US-P5 Re 28,426, as the control voltage (from amplifier 30, ll rectifier 31 and smoothing filter 32) increases, the cut off 12 fre~uency of the variable RC Filter 22 rises. Thus, with larger 13 values of capacitance in network 52 (thus lowering the control 14 amplifier turnover frequency), the variable filter responds by moving up in frequency from its quiescent value. The threshold 16 of the overshoot suppressor is lowered by the application of 17 suitable DC biases (in the forward direction) to the diodes 28.
18 Alternativelyv the gain of amplifier 26 (Figure 3) can be 19 increased to the required level or the amplifier 26 gain can be increased to a high level and attenuation used to adjust the 21 signal level to the diodes.
22 Figure 7 shows a block diagram of a fixed band dual-path 23 bi-linear compressor and expander configuration. The fundamental 24 aspects of this system is disclosed in US-PS 3,846,719, US-PS
3,903,485 and in Journal of the Au io Engineering Society, Vol.
26 15, No. 4, October, 1967, pp. 383-388.
27 In the known embodiment of Figure 7, the further path 28 networks 250 provide four bands. Bands 1, 3 and 4 have conven-29 tional 12 dB/octave input filters: an 80 Hz low pass filter 252 at the input of band 1, a 3 kHz high pass filter 2S4 at the 31 input of band 3 and a 9 kHz high pass filter 256 at the input of 1 1~759 1 ¦ band 4. Each is followed by an emitter follower isolation stage 2 ¦ 258. Band 2 has a frequency response which is complementary to 31 that of bands 1 and 3. Such a response is derived by adding (in 41 adder 260) the outputs of the emitter followers 258 in bands 1 5¦ and 3 and subtracting that sum from the overall input signal (in 61 subtractor 262). The output of emitter follower 258 in each 71 band and the output of subtractor 262 are applied to respective 8¦ except that limiters 264' in bands 1 and 2 have time constants 91 twice those in bands 3 and 4. The outputs of bands 1-4 are 10¦ combined with the main path signal in combiner 265. The com-11¦ pressor output is applied to a noisy channel for transmission to 12 ¦ the complementary expander in which the output of the identical 13¦ further path networks are subtracted from the input signal to 14¦ provide the complementary expansion characteristic.
15¦ Figure 8 shows further details of the limiters 264 16¦ and 264'. Each includes an FET attenuator 270 that operates 17¦ in response to a control signal. The attenuator output is 18¦ amplified by signal amplifier 272, the gain of which is set 19¦ to provide the desired low level signal gain. The outputs 20¦ of all the bands are combined with the main siqnal in such a 21 ¦ way as to produce a low level output from the compressor which 22 ¦ is uniformly 10 dB higher than the input signal up to about 23 ¦ 5 kHz, above which the increase in level rises s~oothly to 24 ¦ I 5 dB at 15 k~z.
25 ¦ The EET attenuator is controlled by a control signal 26¦ sub-circuit that provides a compression threshold of 40 dB below 27~ peak operating level. The control sub-circuit includes control 28 signal amplifier 276 followed by a phase splitter 278 which 29 I drives a full wave rectifier 280. The resulting DC is applied 30 ~ to a smoothing network 282 the output of which is the control 31 ~ signal. Network 282 includes an RC pre-integrator, an emitter 32 j follower and a final RC integrator that operate in conjunction 1~77759 l with diodes such that both the pre- and final integrators have 2 non-linear characteristics produced by the diodes. Fast, 3 large changes in signal amplitude are passed quickly, whereas 4 small changes are transferred slowly. This dynamic smoothing action produces optimum results with respect to modulation 6 effects, low fre~uency distortion, and distortion components 7 generated by the control sianal. The circuit achieves both fast 8 recovery and low signal distortion~
9 Figure 9 shows generally the possible cross-coupling configurations between two series bi-linear devices. If more ll than two devices are operated in series, the possible cross-12 coupling configurations increases. Thus, for example, the first 13 device may be cross-coupled to the third device, and so on.
14 Referring to Figure 9, "n" possible cross-couplings from com-pressor 2 to compressor 4 are shown, having respective transfer l6 functions f1(s), f2(s) to fn(s). Also, "n" possible cross-l7 couplings from compressor 4 to compressor 2 are shown, having l8 respective transfer functions g1(s), g2(s) to gn(s). In 19 the complementary expander arrangement of expanders 6 and 8, the cross-couplings are reversed such that the g1(s), g2(s) and 21 gn(s) cross-couplings are from expander 6 to expander 8 and 22 the f1(s), g2(s) and gn(s) cross-couplings are from 23 expander 8 to expander 6. Thus, in general, there may be one or 24 more cross-couplings, either forward or backward [f(s) or g(s)]
and the cross-couplina(s) may be in only one direction (e.g., 26 either the f(s) or g(s) directions are omitted) or, alterna-27 tively, may be in both directions via a single coupling means.
28 The transfer functions f1(s)~ f2(5), g1(s)~
29 etc., may be implemented by various active or passive devices which may include freauency and/or level dependent elements.
3l The input and output connections of the cross-coupling paths may ~777t~

1 include suitable points for deriving from or coupling to any of 2 the following: input signals, output signals, main path signals 3 ¦ (in a dual-path bi-linear device), further path signals (in a 41 dual-path bi-linear device) f the AC input signals to the control 51 circuit(s), or the DC control circuit signals themselves (the 61 latter two in bi-linear circuits where compression or expansion 71 is effected by a controllable element responding to a control 8~ signal sub-circuit).
91 In Figure 10, an exemplary cross-coupling arrangement 10¦ is shown between the further paths of dual path bi-linear 11¦ compressors and expanders. The series devices are arranged 12¦ such that the syllabic threshold of the first compressor circuit 13 is at a higher level than the second compressor circuit. For 14 complementarity, the order is reversed in the series expander circuits. Blocks M1 and N2 denote the further path circuitry.
16 In the arrangement of ~igure 10 the output of the high level 17 stage 280 further path from N1 is applied through coupling 18 circuitry having a transfer function f(s) into the circuitry of 19 the further path circuit N2 of low level stage 282. The transfer function f(s) may have suitable frequency and phase 21 characteristics so as to enhance the limiting action of the low 22 level compressor stage and conse~uently, the noise reduction 23 effect. If the bi-linear devices are sliding band devices the 24 signal from the high level stage is injected, for example, in the filter circuitry of the low level stage in order to enhance 26 the sliding band action. In the complementary expander the 27 output from N1 in the high level expander stage is applied 28 through the same transfer characteristic f(s) to the circuitry 29 of the low level stage further path circuit N2.
///

l ¦ A specific embodiment of the general arrangement of 2 ~ Figure 10 is shown in Fiqures 11 and 12. ~igure 11 shows the 3 ¦ series compressor circuits 280 and ~82, indicating the input and 4 ¦ output connection points for the cross-coupling that includes 51 the transfer function f(s). Figure 12 shows the details of the 61 transfer function f(s) network and its connection to the filter 71 circuitry of compressor 282. For ~onvenience, the filter 8¦ circuitry of compressor 282 is taken to be as described in 91 connection with Figure 5.
l0¦ The cross-coupling network of Figure 12 includes a ll¦ high frequency boost network 284 that provides a 10 dB boost at 12¦ high frequencies and having a corner frec~uency equal to the 13 quiescent filter corner frequency of circuit 280. The network 14 284 output is split into two paths and injected via adjustable gain means into the fixed filter 20 and the variable filter 22.
16 One end of the 3.3 K resistor in fixed filter 20 is lifted from 17 ground and the signal derived via potentiometer 286 and amplifier l8 288 is then applied. The end of the 39 K resistor 36a that was 19 formerly connected at the junction of the 0.033 capacitor and the 3.3 X resistor is lifted from that junction and the signal 21 derived via potentiometer 290 and amplifier 292 is applied 22 thereto. Amplifier 288 has a gain of about 3/4 and amplifier 23 292 has a gain of unity.
2~ In operation, at low levels, the high level circuit 280 would not yet operate. Under this condition voltage V2 26 eauals voltage V4, because voltage V3 contains a low level 27 high frequency boost (resulting from the below threshold gain 28 of high level circuit 280) which is matched by the boost 29 network 284. The signal levels applied to the fixed filter 30l 20 and variable filter 22 may be adjusted to obtain the best 31¦ results. If about 3/4 of the network 284 output sîgnal is fed 32~ to the 3.3 K resistor, its effective resistance becomes about 13 ~17775~

1 K. When the high level circuit reaches its threshold, both 2 filters (20 and 22~ in the low level circuit 282 would then have 3 a sliding band action that would improve the overall noise 4 modulation performance without exacerbating mid-band modulation effects, i.e. excessive band sliding is minimized.
6 Figure 13 shows generally an arrangement for cross-7 coupling control signal components from one series device to 8 another. In the example of Figure 13, a control signal component ¦
9 from smoothing network 32 of the low level compressor 232 is applied via transfer function f(s) block 294 to the smoothing 11 network 32 of the high level stage 22.
12 In operation, the low level circuit 282 has the first 13 and largest response to incoming transients and thus, the 14 cross-coupled signal provides the high level processor with a timely warning of impending signal amplitude changes.
16 One suitable cross-coupling transfer function f(s) is 17 provided by the network shown in Figure 14, which includes an 18 emitter follower 296 which drives series diodes 298, 300 in 19 parallel with a resistor 302. A 560 K resistor is connected between the anode of diode 300 and ground. The emitter follower 21 has a 220 K resistor in its base input that is driven from the 22 input of the first smoothing stage of the low level circuit.
23 Thus, the input signal is essentially a pulse from rectifier 31 24 that is not delayed by time constants in the low level smoothing stage. The pulse is AC coupled by the emitter follower stage, 26 which can provide adequate current dumping, into the output 27 filter section of the high level stage smoothing network, thus 28 providing a fast additional control signal component.
29 In Figure 15, a further embodiment of the invention for cross-coupling series sliding band dual-path devices is 31 shown. Elements common to the embodiment of Figure 11 retain 32~1 /// , ~ -23-11777~9 l the same reference numerals. According to the embodiment of 2 Figure 15, a single control circuit (blocks 30, 31, 32) is fed 3 by adder resistors 304 and 306 from the outputs of the further 4 ¦ paths in each of the series compressors 280 and 282. The 51 control circuit output is applied to the variable filter 22 of ~ j each series compressor through respective level setting means 71 308 and 310 (if reguired). The values of adder resistors 304 81 and 306 may be selected to weight the controlling effect of one 9¦ compressor with respect to another. The arrangement provides an lO¦ economy in circuit components, while providing performance in ll¦ large part similar to configurations employing individual 12 ¦ control circuits for each series device. A single control 13 circuit may also be provided for the case of three or more 14 series connected compressors or expanders, in which case each further path output is fed through a summing resistor to the 16 control circuit input, the output of which is applied through 17 respective level setting means (if required) to the respective 18 variable filters.
19 A single control circuit arrangement is also applicable 20 ¦ to fixed band series connected devices as shown in the embodiment 2l¦ of Figure 16, in which fixed band compressors having a single 22 ¦ further path are shown. The arrangement is also applicable to 23 ¦ series fixed band compressors and expanders each having a 24 I plurality of ~urther paths, in which case the common control 25 j circuit iSf of course, connected between or among the further 26 ¦ paths operating in the same frequency band.
27 ¦ Referrinq to Figure 16, the input signal is split in 28 ¦ compressor 326 into a main path which is applied to a combining 29 network 316 and to a further path that includes a fixed filter 30 ¦ 312 and a voltage controlled amplifier (VCA) 314. The VCA may 31 ¦ be an FET attenuator followed by an amplifier such as described 32 ¦ in connection with Figure 8 (blocks 270 and 272). In the second ~ 1~7'775~
l ¦ compressor 328, the threshold of VCA 314' is staggered with 2 ¦ respect to VCA 314 in the first compressor 326, as explained in 3 ¦ Belgian-PS 889, 428. A common control circuit (blocks 276, 278, 4 ¦ 280 and 282, as described in connection with Figure 8), is fed 5 ¦ by summina resistors 318 and 320 from the further path output in 6 the manner of the Figure 15 embodiment. The control circuit 7 output is applied to the V~A's 314 and 314' through level 81 adjusting means 322 and 324. The arrangement is also applicable 9 ~ to three or more series fixed band devices in the same way as lO¦ mentioned in connection with the embodiment of Figure 15.
ll¦ The embodiments of Figures 15 and 16 are both based on 12 ¦ the observation that in such series connected compressors and 13 ¦ expanders, the dynamic action occurs primarily in one stage at a 14 ¦ time. For example, starting at low input signal levels, the l5¦ lowest level stage generates most of the combined control 16¦ signal. As the input signal level rises the lowest level stage 17 ¦ phases out of dynamic action and the next higher threshold level l3 stage becomes active and contributes most of the combined 19 ¦ control signal.

21 ~

30 ~

~1 -25-

Claims (14)

THE EMBODIMENTS OF THE INVENTION IN WHICH AN EXCLUSIVE
PROPERTY OR PRIVILEGE IS CLAIMED ARE DEFINED AS FOLLOWS:

1. A circuit for modifying the dynamic range of an in-put signal, comprising first circuit means with a bi-linear characteristic composed of a low level portion of substantially constant gain up to a threshold, an intermediate level portion, above the threshold, of changing gain providing a maximum com-pression ratio or expansion ratio, and a high level portion of substantially constant gain different from the gain of the low level portion, at least one second circuit means, which also has a bi-linear characteristic within a frequency range common to the circuit means, in series with the first circuit means thereby to provide an overall characteristic which is also bi-linear, the intermediate level portions of the char-acteristics of the circuit means being staggered within the frequency range common to the circuit means such as to provide a change of gain over a wider range of intermediate input levels than for any of the circuit means individually, and an increa-sed difference between the gains at low and high input levels, but with an overall maximum compression or expansion ratio which is substantially no greater than that of any single cir-cuit means, by virtue of the staggering, and at least one coupling circuit means coupling signal components from one of the circuit means to another of the circuit means for modify-ing the action of the said other circuit means in response to coupled signal components from said one circuit means.

2. The circuit arrangement of claim 1 wherein said coup-ling circuit means includes means for processing coupled sig-nal components.

3. The circuit arrangement of claim 2 wherein said means for processing includes frequency or level dependent elements.

4. The circuit arrangement of claim 1 or 2 wherein said at least one coupling circuit means comprises a common control circuit for controlling the dynamic action of said circuit means.

5. The circuit arrangement of claim 1 or 2 wherein said at least one coupling circuit means comprises a common control circuit for controlling the dynamic action of such circuit means, said common control circuit deriving a control signal from signal components from each of said circuit means.

6. The circuit arrangement of claims 1 or 2 wherein each of the circuit means between or among which the coupling circuit means couples or couple signal components comprise a variable gain or variable pass filter circuit and a control circuit which derives a control signal for the said variable circuit, the coupling circuit means coupled from the control circuit of the said one circuit means to the control circuit in the said other circuit means for modifying the control sig-nal of said other circuit means.

7. The circuit arrangement of claim 1 or 2 wherein each of the circuit means between or among which the coupling cir-cuit means couples or couple signal components comprises a main signal path having a linear characteristic with respect to dynamic range, a combining circuit in the main signal path, and a further path having a non-linear characteristic with respect to dynamic range and having its input coupled to the input or output of the main signal path and its output coupled to the combining circuit which combines the signal from the further path additively or subtractively with the signal in the main path, said at least one coupling circuit means coupled from the output of the further path of the said one circuit means to the further path of the said other circuit means.

8. A circuit for modifying the dynamic range of an in-put signal, comprising first circuit means having a character-istic in which only a portion thereof has a changing gain bet-ween low and high input levels, at least one second circuit means which also has such a characteristic within a frequency range common to the circuit means, in series with the first circuit means, the changing gain portions of the characteristics of the circuit means being staggered within the frequency range common to the circuit means such as to provide a change of gain over a wider range of input levels than for any one of the circuit means individually, and an increased difference between the gains at low and high input levels, but with a maximum overall compression or expansion ratio which is substan-tially no greater than that of any single circuit means, by virtue of the staggering, and at least one coupling circuit means coupling signal components from one of the circuit means to another of the circuit means for modifying the action of the said other circuit means in response to coupled signal components from said one circuit means.

9. The circuit arrangement of claim 8 wherein said coup-ling circuit means includes means for processing coupled signal components.

10. The circuit arrangement of claim 9 wherein said means for processing includes frequency or level dependent elements.

11. The circuit arrangement of claim 8 or 9 wherein said at least one coupling circuit means comprises a common control circuit for controlling the dynamic action of said circuit means.

12. The circuit arrangement of claim 8 or 9 wherein said at least one coupling circuit means comprises a common control circuit for controlling the dynamic action of said circuit means, said common control circuit deriving a control signal from signal components from each of said circuit means.

13. The circuit arrangement of claims 8 or 9 wherein each of the circuit means between or among which the coupling circuit means couples or couple signal components comprise a variable gain or variable pass filter circuit and a control circuit which derives a control signal for the said variable circuit, the coupling circuit means coupled from the control circuit of the said one circuit means to the control circuit in the said other circuit means for modifying the control signal of said other circuit means.

14. The circuit arrangement of claim 8 or 9 wherein each of the circuit means between or among which the coupling cir-cuit means couples or couple signal componnets comprise a main signal path having a linear characteristic with respect to dynamic range, a combining circuit in the main signal path, and a further path having a non-linear characteristic with respect to dynamic range and having its input coupled to the input or output of the main signal path and its output coupled to the combining circuit which combines the signal from the further path additively or subtractively with the signal in the main path, said at least one coupling circuit means coupled from the output of the further path of the said one circuit means to the further path of the said other circuit means.