AU2002239060B1

AU2002239060B1 - Electronic watthour meter and power-associated quantity calculating circuit

Info

Publication number: AU2002239060B1
Application number: AU2002239060A
Authority: AU
Inventors: Keishu Kondo; Atsufumi Kuroda; Takashi Shindoi
Original assignee: Mitsubishi Electric Corp
Current assignee: Mitsubishi Electric Corp
Priority date: 2002-03-25
Filing date: 2002-03-25
Publication date: 2003-10-08
Anticipated expiration: 2022-03-25
Also published as: CN1292259C; CN1493002A; WO2003081264A1; JPWO2003081264A1; JP4127676B2

Description

09236

SPECIFICATION

ELECTRONIC ELECTRICITY METER AND ELECTRIC-POWER-RELATED QUANTITY ARITHMETIC CIRCUIT TECHNICAL FIELD The present invention relates to an electronic electricity meter and an electric-power-related quantity arithmetic circuit for measuring electric energies. More particularly, the invention is concerned with an electronic electricity meter and an electric-power-related quantity arithmetic circuit which can arithmetically determine or compute quantities relating to electric power, i.e., electric-power-related quantities, such as effective power, effective electric energy, reactive power, reactive electric energy, apparent power, distortion power, effective current value, effective voltage value, phase difference between current and voltage, values of these quantities of individual harmonic components and so forth.

BACKGROUND TECHNIQUES Related Art 1 The conventional electronic electricitymeter known heretofore includes first and second successive or sequential comparison type A/D converters as a means for converting analog quantities of voltage and current measured by a voltage sensor (instrument potential transformer or PT in short) andacurrent sensor (instrument current transformer or CT in short) into digital values, respectively, and is so arranged as to determine an electric power W by arithmetically processing these digital outputs (digitized voltage and current) bymeans of a multiplier, as is disclosed, for example, in Japanese Patent No. 3080207.

Ingeneral, ineachofthe first andsecondsequential comparison type A/D converters, the analog input signal is quantized into an output digital value which increases discretely or incrementally with a constant resolution.

Consequently, for ensuring, so to say, an absolute accuracy for the analog-to-digital conversion of the input signal of a low level, there is required a sequential comparison type A/D converter of a high resolution.

On the other hand, as a method of enhancing the accuracy of the analog-to-digital conversion, a method of increasing a sampling frequency of the first and second sequential comparison type A/D converters so-called oversampling method) is known. By way of example, when the sampling frequency is increased to a frequency 128 times as high as the sampling frequency determined on the basis of the Nyquist theorem, quantization noise is dispersed widely over a broad band, as a result of which the levels of the individual frequency components spectra become lower, whereby the noise levels of the signal frequency components are improved correspondingly. This is equivalent to increasing of the resolution of the first and second sequential comparison type A/D converters by a few or several bits.

However, in order to realize the electronic electricitymeterofhighaccuracy onthebasis oftheteachings disclosed in the related art 1 mentioned above, there are demanded the first and second sequential comparison type A/D converters each capable of ensuring high resolution and a multiplier having multi-bit inputs, which will naturally lead to complication of the circuit configuration, incurring high manufacturing cost as well. Thus, great obstacle has been encountered in the attempt to realize the electronic electricitymeterinamonolithicintegratedcircuitstructure on a mass production basis.

Related Art 2 As an example of the means for solving the problems mentioned above, there is also disclosed in Japanese Patent No. 3080207mentionedaboveamethodofintegratingthe current and the voltage with the respective integrators to output the digital values by way of comparators, wherein values obtained through D/A conversion of the digital outputs with a time lag are fed back to the input side of the above-mentioned integrators, respectively.

To this end, the electronic electricity meter includes first and second delta-sigma-type A/D converters for quantizing, respectively, the current and the voltage mentioned above with an oversampling frequency, first and second moving average processing means for determining moving averages of the quantized current and voltage, respectively, with digital filter means, a multiplication means for determiningthe electric power valuebymultiplying the current and the voltage resulting from the moving average processing, and an integrating means for integrating the electric power value determined through the multiplication.

As can be appreciated from the above, it is possible to reduce remarkably the quantization noise -in the low frequency band according to the teaching of the related art 2. To say in another way, with the teachings of the related art 2, the effective output bits of the A/D converter are virtually increased when compared with those of the sequential type A/D converter, and thus the electronic electricity meter of high accuracy can be implemented in a relatively simple circuit configuration. In particular, the circuit configuration can easily be simplified when the electronic electricity meter is to be implemented in a monolithic integrated circuit structure.

However, in theconventional electronic electricity meter, the electric energy is arithmetically determined or computed by multiplying directly the current and the voltage at every sampling time point (with the sampling frequency of theA/Dconverterbeing fixed) because it is required to measure (ordisplay) the electric energy on a real time basis. Further, the electric energy is arithmetically determined or computed by acquiring the input signals (current and voltage signals) each containing high frequency components which are not lower than the Nyquist frequency through the medium of the low-pass filters. The electric energy determined in this way will contain both the fundamental wave component and the harmonic components which are synthesized admixedly, and thus it is impossible to carry out the measurement only for the fundamental wave component or only for the harmonic component(s). Equally, neither the reactive power of the fundamental wave component nor that of the harmonic component can be measured.

On the other hand, in many of electric power instruments/equipments/machines developed in the recent years, thyristors and inverters are employed, as a result of which the current as handled contains harmonic component(s) in many applications. Such being the circumstances, in the electric power instruments/equipments/machines mentioned above, there arerequiredmultifariousmeasurements ofvarious electric-power-related quantities inclusive of those of the harmonic components.

In this conjunction, measurement of the harmonic component(s) with the conventional electric energy meter is considered to be theoretically possible by resorting to the Fourier transformprocedurewell known inthe art for detecting the harmonic component(s). However, since the sampling frequencyoftheA/Dconverterdoesnotcoincidewith a positive integer multiple (multiple of a natural number) of the frequency of the power supply source, power source frequency, the values obtained through the Fourier transform of the current and the voltage will cover the adjacent harmonic component value of K-th harmonic component will be admixed in (K+1)-th harmonic component as a result of the Fourier transform).

Such being the circumstances, it is necessary to arithmetically correct the harmonic components with the power source frequency in order to derive the harmonic component(s) in view of the fact that the sampling frequency of the A/D converter is fixed, which means that not only the detection of thepowersource frequencybut also thearithmeticoperation for the correction of the harmonic component is required. For these reasons, much expensive processor (Central Processing Unit or CPU) is required for the electronic electricity meter for which real-time-based operation is demanded among others.

In that case, since there are demanded the A/D converter of high sampling frequency and hardware structure similar to that of the expensive high-accuracy measuring instrument or meter incorporating the processor of high operation speed which are intrinsically developed for the applications other than the electronic electricity meter, it is practically impossible to adopt such A/D converter and measuring instrument or meter in the electronic electricity meter designed for the use by consumers in general.

Certainly, it is conceivable to adopt such arrangement to cause the sampling frequency of the A/D converter to follow up a positive integer multiple of the power source frequency with a view to rendering unnecessary the correcting or modifying operation mentioned above. To this end, however, it is necessary to control the sampling frequency by the CPU. In that case, when the sequential A/D converter mentioned previously in conjunction with the related art 1 is employed, resolution of the CPU output clock on the order ofseveralMHzisdemandedforthesampling frequencyofseveral Khz, by way of example. Accordingly, when the degree of the harmonic component to be measured increases, it becomes impossible to control the sampling frequency by the CPU with sufficiently high accuracy.

Further, even in the case where the oversampling-basedmeasurementisadoptedinconjunctionwith the sequential A/D converter of the related art 1 described previously (or when the delta-sigma-typeA/Dconverter of the related art 2 is employed), it is equally impossible to control the sampling frequency with sufficiently high accuracyby the CPU because the resolution of the output clock frequency of the CPU lies in the range of several Mhz to several ten MHz whereas the oversampling frequency is in the range of several hundred kHz to several MHz several kHz).

Incidentally, the effective power and the reactive power of the fundamental wave component can certainly be determinedarithmeticallybyrotatingthecurrentbyresorting to the Hilbert transform, as is disclosed in Patent Application (PCT 2002JP00045, not laid open yet) filedprecedently by the inventors of the present application. However, the range of frequency components which can be rotated for 90 degrees by one and the same Hilbert transformer (with the Hilbert transformercapableofrotatingthe fundamentalwavecomponent for 90 degrees, angle of rotation for the harmonic component will deviate from 90 degrees), it is difficult to determine the reactive power(s) of other component(s) than the fundamentalwave component by means of one andthe sameHilbert transformer.

In the foregoing, description has been made as to the structures of the conventional electronic electricity meters and electric-power-related quantity arithmetic circuits known heretofore. As is apparent from the above, in order to realize high accuracy of measurement with the electronic electricity meter of the related art 1, there are demanded the first and second sequential comparison type A/D converters each of high resolution and the multiplier having the multi-bit inputs, incurring complexity in the circuit configuration, giving rise to a problem that the cost is increased.

Further, intheconventionalelectronicelectricity meter, the current and the voltage are straightforwardly multiplied for computing the electric energy. Consequently, what is determined is the electric energy of the admixedly synthesized fundamental wave and harmonic components. In other words, it is impossible to measure the electric energy onlyforthefundamentalwavecomponentoronlyfortheharmonic component, presenting a problem remaining to be solved.

7 Besides, neither the reactive power of the fundamental wave component nor that of the harmonic component can be measured, to another disadvantage.

Furthermore, in the case where the harmonic component is to be measured by applying the Fourier transform well known in the art, arithmetic operation for correcting or modifying the harmonic component is demanded because the sampling frequency of the A/D converter is not represented by the positive integer multiple of the power source frequency and thus the value resulting from the Fourier transform also extends to the harmonic component of the adjacent degree, as pointed out hereinbefore. For these reasons, there is required the much expensive processor which can not be incorporated in the electronic electricity meter designed for the general purpose, presenting a problem.

Additionally, in the case where the oversampling is adopted in association with the sequential A/D converter for the measurement or where the delta-sigma-type A/D converter is made use of, it is impossible to control accurately the sampling frequency with the CPU output clock of the resolution on the order of several MHz to several ten MHz because the oversampling frequency is very high on the order of several hundred kHz to several MHz, incurring a problem that it is impossible to render unnecessary the correcting or modifying operation by making the sampling frequency of the A/D converter follow up a positive integer multiple of the power source frequency.

8 Furthermore, in the case where the effective power and the reactive power of the fundamental wave are arithmetically determined, there arises such problem that because the range of the frequency which can be rotated for 90 degrees by one and the same Hilbert transformer is narrow, the reactive power of other components than the fundamental wave component can not be measured by using one and the same Hilbert transformer.

SUMMARY OF INVENTION According to a first aspect of the present invention, there is provided an electronic electricity meter having a microprocessor (CPU) which comprises an A/D converter for converting measurement signals indicative of a current and a voltage of a power line into digital values to be fetched and electric energy arithmetic means for thereby arithmetically determining an electric energy of said power line on the basis of said digital values, wherein said electronic electricity meter comprises: power source frequency detecting means for detecting a power source frequency of said power line; sampling frequency control means provided separately from a clock circuit of said microprocessor (CPU) for thereby controlling a sampling frequency of said A/D converter such that said sampling frequency is set to be equal to a positive integer multiple of said power source frequency; and said electronic energy arithmetic means comprises a harmonic arithmetic unit for arithmetically determining electric energy of a harmonic component.

9 Preferably, said sampling frequency control means controls said sampling frequency so that said sampling frequency is set to be equal to 2 m of the power source frequency, and that said electric energy arithmetic means comprises fast Fourier transform means and is so arranged as to arithmetically determine the electric energy of the harmonic component through fast Fourier transform.

Preferably, said A/D converter is implemented in the form of a delta-sigma-type A/D converter for oversampling.

Preferably, said sampling frequency control means comprises: zero-cross point detecting means for detecting a zero-cross point rising or alternatively upon falling of said power source frequency; and power source phase difference detecting means for detecting a lag or lead of said power source frequency as a phase difference of the power source on the basis of an A/D conversion value of said power source frequency sampled at a time point after lapse of one period from a preceding zero-cross point upon rising or alternatively falling of said power source frequency, wherein said sampling frequency control means is so arranged that the sampling frequency of said A/D converter is controlled on the basis of said power source phase difference.

Preferably, said sampling frequency control means comprises: absolute phase detecting means for detecting absolute phase of the voltage of said power line; and 10 voltage phase difference detecting means for detecting as a voltage phase difference a lag or lead of said absolute phase at a time point after lapse of one period from the preceding absolute phase of said voltage, wherein said sampling frequency control means is so arranged that the sampling frequency of said A/D converter is controlled on the basis of said voltage phase difference.

Preferably, said voltage phase difference detecting means is so arranged that an absolute phase of 0 degree, degrees, 180 degrees or 270 degrees is set as a reference.

Preferably, said sampling frequency control means comprises: sampling frequency correcting means for correcting the sampling frequency of said A/D converter; and that said sampling frequency correcting means comprises: control quantity arithmetic means for arithmetically determining a control quantity for said sampling frequency; D/A conversion means for converting said control quantity for said sampling frequency into an analog voltage to be outputted through D/A conversion; and offset means for offsetting the output voltage of said D/A conversion means, wherein said offset means is so arranged as to offset said output voltage of said D/A conversion means by a predetermined voltage toward a prescribed frequency of said power line.

11 Preferably, said offset means comprises offset voltage regulating means for adjusting said predetermined voltage from outside.

Preferably, said electric energy arithmetic means comprises: A/D conversion data packing means for packing measurement signals of the voltage and the current over one period of said power source frequency; an electric power arithmetic unit for arithmetically determining a first power value through Fourier transform of data packed by said A/D conversion data packing means; packing time detecting means for detecting a packing time taken for packing said voltage or current of one period; electric power output means for holding said first power value for every packing time and outputting said first power value as a second power value every time a sampling command is issued at a predetermined arithmetic operation period; and electric energy pulse output means for integrating said second power value to thereby output an electric energy pulse every time a quadrature value of said second power value attains a predetermined value.

Preferably, said electric energy arithmetic means comprises: A/D conversion data packing means for packing measurement signals of the voltage and the current over one period of said power source frequency; an electric power arithmetic unit for arithmetically determining electric-power-related values and phase differences thereof for harmonic components, respectively, 12 on a per degree basis through Fourier transform of data packed by said A/D conversion data packing means; and phase correcting means for correcting said current or voltage of one period through rotational operation so that the phase differences of said electric-power-related values represent true phase differences of electric-powerrelated values on a measured system side.

According to a second aspect of the present invention, there is provided an electric-power-related quantity arithmetic circuit, comprising: an A/D converter for converting measurement signals indicative of a current and a voltage of a power line into digital values to be fetched; A/D conversion data packing means for packing digital values for one period; Fourier transform means for performing Fourier transform of data packed by said A/D conversion data packing means; electric-power-related quantity arithmetic means for arithmetically determining electric-power-related quantities on the basis of result of transforation performed by said Fourier transform means; sampling frequency correcting means for arithmetically determining a correcting quantity for a sampling frequency of said A/D converter on the basis of a frequency of said current or said voltage; and a voltage-controlled oscillator for outputting a sampling frequency which changes in dependence on said correcting quantity to said A/D converter.

13 BRIEF DESCRIPTION OF THE DRAWINGS Figure 1 is a view for illustrating a theoretical concept underlying arithmetic determination or computation of electric-power-related quantities according to a first embodiment of the present invention in terms of a voltageand-current vector diagram, by way of example.

Figure 2 is a block diagram showing a circuit arrangement of a major portion of an electronic electricity meter according to the first embodiment of the present invention.

Figure 3 is a view for illustrating behaviours of voltage of a fundamental wave when frequency of a power source changes in the electronic electricity meter according to the first embodiment of the present invention.

Figure 4 is a view for illustrating a zero-cross point and a sampling start point in the electronic electricity meter according to the first embodiment of the present invention.

Figure 5 is a block diagram showing a circuit arrangement of a major portion of an electronic electricity meter according to a second embodiment of the present invention.

Figure 6 is a view for illustrating a correcting processing for causing a sampling frequency of an A/D converter to follow up a positive integer multiple of a power source frequency in the electronic electricity meter according to the second embodiment of the present invention.

Figure 7 is a view for illustrating a sampling start point in an electronic electricity meter according to a third embodiment of the present invention.

Figure 8 is a block diagram showing a circuit arrangement of a major portion of an electronic electricity meter according to a fourth embodiment of the present invention.

Figure9isaviewforillustrating, bywayofexample, operation for offsetting a VCO control voltage in the electronic electricity meter according to the fourth embodiment of the present invention.

Figure 10 is a block diagram showing a voltage adder circuit capable of adjusting a VCO control voltage in the electronic electricity meter according to the fourth embodiment of the present invention.

Figure 11 is a block diagram showing a circuit arrangement of a major portion of an electronic electricity meter accordingtoa fifth embodiment ofthepresent invention.

Figure l2isatimingchart forillustratingelectric energy pulse output operation in the electronic electricity meter according to the fifth embodiment of the present invention.

Figure 13 is a block diagram showing a circuit arrangement of a major portion of an electronic electricity meter accordingto a sixthembodimentofthepresent invention.

BEST MODES FOR CARRYING OUT THE INVENTION Embodiment 1 In the following, a first embodiment of the present invention will be elucidated.

With the present invention, it is contemplated to arithmetically determine or compute electric energies and others throughFouriertransform (preferablythroughFFT: Fast Fourier Transform) of data over one period, differing from the arithmetic operation adopted in the conventional electronic electricity meter known heretofore.

In the first place, description will generally and briefly be made on the theoretical concept underlying the arithmetic operations adopted in the electronic electricity meter according to the first embodiment of the present invention.

When the sampling frequency of an analog-to-digital or A/D converter is given by a positive integer multiple (multiple of a natural number) of the frequency of a power source power source frequency), the electric-power-related quantities mentioned below can be measured through the Fourier transform.

It is presumed that the sampling frequency of the A/D converter is L times as high as the power source frequency (where L represents a positive integer or natural number) and that the order or degree of a harmonic is represented by n.

Accordingly, when n 0, this represents the DC component, while n 1 represents a fundamental wave (primary harmonic component). On the presumption, the maximum harmonic degree H which can be determined through the Fourier transform is represented by the following expression H floor (1) In the above expression "floor represents a function for rounding off the decimal fraction. Thus, maximum harmonic degree H is given by a natural number.

Through the Fourier transform of data of a point L as obtained by the A/D conversion, complex number values indicating the amplitudes and the phase information, respectively, can be obtained over the range of the harmonic components from the DC component to the harmonic component of degree H. Representing by "Vn cmp" and "In cmp", respectively, the complex number values of the harmonic component of degree n obtained through the Fourier transform of the A/D conversion of the voltage and the current data at the point L, then these complex number values "Vn_cmp" and "In cmp" can be represented by the undermentioned expressions and respectively: Vn _cmp Vn _re +j -Vn _im (2) In_ cmp In _re j. In _im (3) In the above expressions and j represents the imaginary unit and" "represents the multiplication sign.

Further, "Vn re", "Vn im", "In re" and "In im" are given by real numbers, respectively. In the following description, it is presumed that the values obtained after the Fourier transform are normalized in terms of the effective value.

At this juncture, the effective power Wn of the harmonic component of degree n can be determined in accordance with the undermentioned expression Wn=Vn _reIn _re+Vn _imIn _im (4) Further, the electric power W containing all the harmonic components can be determined in accordance with the following expression

H

W Wn 11=0 In this conjunction, the reactive power Varn of the harmonic component of degree n can be determined in accordance with the following expression Varn Vn im In re Vn re In im (6) In that case, when the reactive power is of positive sign (plus), the current lags relative to the voltage, whereas the current leads relative to the voltage when the reactive power is negative (minus). Further, the reactive power Var containing all the harmonic components can be determined in accordance with the following expression

H

Var Vain (7) n=0 On the other hand, the effective voltage value (effective value of the voltage) Vrmsn of the harmonic component of degree n can be determined in accordance with the following expression Vrmsn Vn re Vn re Vn im Vn _im (8) The effective voltage value (effective value of the voltage) Vrms containing all the harmonic components can be determined in accordance with the following expression

H

Vrms Z Vrmsn (9) 11=0 The effective current value Irmsn of the harmonic component of degree n can be determined in accordance with the following expression Irmsn jh7 re- In re In_ imn In_ im The effective current value Irms containing all the harmonic components can be determined in accordance with the following expression (11):

H

Irms hrmsn (11) n=o The apparent power VAn of the harmonic component of degree n can be determined in accordance with the following expression (12): VAn Vrmsn Irmsn (12) Further, the apparent power VA containing all the harmonic components can be determined in accordance with the following expression (13): VA Vrms- Irns (13) In the case where only the fundamental wave components are contained in the power source voltage and current, the relation among the effective power W, the reactive power and the apparent power VA satisfies the condition given by the undermentioned expression (14): VA W -W +Var -Var (14) However, when the harmonic component exists, the above-mentioned expression (14) does not hpld valid, but a distortion power component D exists, which can be determined in accordance with the following expression D= VA-VA-W-W-Var-Var The distortion power component D assumes a real number value of positive (plus) sign and does not assume negative or minus value.

The complex number values Vn_cmp and In_cmp of the voltage and the current of the harmonic component of degree n obtained through the Fourier transform can be represented as illustrated in the vector diagram of Fig. 1, in which the real axis is taken along the abscissa with the imaginary axis being taken along the ordinate.

In Fig. 1, 0 represents the absolute phase of the complex number value Vn_cmp, and 4 represents the absolute phase of the complex number value In_cmp. The reference for the absolute phase is taken in the positive direction along the abscissa, wherein the counterclockwise direction indicates the lead (positive polarity) while the clockwise direction indicates the lag (negative polarity).

Representing by Phase-VnIn the phase differences 0) between the complex number values Vn_cmp and In_cmp with reference to the complex number value Vn_cmp, then the phase difference Phase_VnIn can be determined in accordance with the undermentioned expression (i6): var 0 Phase VnIn arccos (Wn/VAn) else Phase _VnIn arccos (Wn/VAn) The phase difference Phase_VnIn is positive or plus when the current leads relative to the voltage while the former is negative or minus when the current lags relative to the voltage. Further, the range of the phase difference PhaseVnIn is within ±180 degrees.

In the foregoing, description has been made concerning only the single-phase two-wire system. It should however be understood that the analyses mentioned above can equally apply to the single-phase three-wire system, three-phase three-wire system and the three-phase four-wire system. Since the effective power, the reactive power and the apparent power are sums of the respective values determined for the individual phases, respectively, the distortion power component can also be determined on the basis of the sum of the electric powers by making use of the expression (16).

In addition, the phase difference between the phase voltages can be determined as well. More specifically, representing the n-degree harmonic component voltage voltage of harmonic component of degree n) of the phase A by Vancmp, then this voltage is given by the undermentioned expression (17): Van cn Van_ re j Van _im (17) Further, the n-degree harmonic component voltage voltage of the harmonic component of degree n) Van_cmp of the phaseBisrepresentedbythe following expression (18): Vbn cmp Vbn re j. Vbn _im (18) On the conditions mentioned above, the phase difference Phase_VanVbn with reference to the phase A can be determined in accordance with the undermentioned expression (19): Vabn Van re -Vbn _re Van imi. Vbn im Vabn' Van im Vbn re Van _re Vbn _im Varmsn VVan re Van re Van im Van _im Vbrmsn 4Vbn re -Vbn re Vbn im Vbn im if Vabn'>=0 (19) Vabn Phase Van Vbn -arccos aV Varmsn Vbrmsn else Vabn Phase VanVbn arccos Vbns Varmsn Vbrmsn In the above expression Vabn represents the pseudo-effective power between the phases A and B with the Vabn' representing the pseudo reactive power between the phases A and B.

In the similar manner, the phase differences can be computed for the voltages and the currents of different phases.

As is apparent from the above, so far as the sampling frequency of the A/D converter is a positive integer multiple of the power source frequency, it is possible to obtain the electric-power-related quantities on a harmonic-by-harmonic basis (or a per harmonic basis) for all the electric-power-related quantities including the harmonic components or the electric-power-related quantities including the concerned harmonic components by making use of the result of the arithmetic operation through Fourier transform.

In particular, it should be pointed out that the reactive power can be determined for all the harmonic components. Further, by selecting the sampling frequency of the A/D converter to be equal to the power source frequency multiplied by 2 N (N represents a natural number or positive integer number), the fast Fourier transform or FFT in abbreviation can be made use of in the arithmetic operations involved in the Fourier transform. Accordingly, in the ordinarycase, it ispreferredtoselect the sampling frequency for the A/D conversion equal to a positive integer multiple (multiple of a natural number) of the power source frequency and 2

N

Next, description will be directed to operation of the electronic electricity meter according to the first embodiment of the present invention.

In the description which follows, it is presumed thatintheVCO (voltage-controlledoscillator)controlmethod according to the first embodiment of the present invention, theVCOcontrolislockedupon zero-crossingofthepowersource voltage.

Figure 2 is a block diagram showing, by way of concrete example, a circuit arrangement of the electronic electricity meter according to the first embodiment of the present invention.

Referring to Fig. 2, an A/D converter 1 serves for converting a voltage V and a current I detected by respective sensors (not shown) into digital values, respectively. The A/Dconverter 1 hasanoutputterminaltowhichanA/Dconversion data packing means 2, a Fourier transform means 3 and an electric-power-related quantity arithmetic means 4 are connected serially in this order.

The A/D conversion data packing means 2 serves for packing measurement signals of the voltage and the current over one period of the power source frequency. Furthermore, the A/D conversion data packing means 2 has a function for detecting the power source frequency on the power supply line and includes a means for detecting a zero-cross point upon rising or falling of the power source frequency.

The Fourier transform means 3 performs Fourier transform while the electric-power-related quantity arithmetic means 4 (electric energy arithmetic means) serves also as a harmonic component arithmetic unit for computing the electric-power-related quantities inclusive of the harmonic components.

The data packed by the A/D conversion data packing means 2 is inputted to a voltage-controlled oscillator (hereinafter referred to as VCO in abbreviation) 6 for controlling the A/D converter 1 by way of a sampling frequency correcting means The voltage V and the current I inputted to the A/D converter 1 are derived from the output of the relevant sensors and thus are not physically the voltage and the current of the power supply source itself. In other words, the levels or values of the voltage V and the current I mentioned above are matched to the input performance of the A/D converter 1.

Accordingly, there may be provided on the input side of the A/Dconverterlananti-aliasingfilter, anoperationamplifier for amplification and others (not shown) as the case may be.

The A/D conversion data packing means 2 is designed to pack the A/D conversion data over one period of the power source at every point L.

On the other hand, the Fourier transform means 3 performs the Fourier transform at everypoint L (everyperiod) for the A/D conversion data (point L) of one period inputted from the A/D conversion data packing means 2.

The electric-power-related quantity arithmetic means 4 is designed to compute the electric-power-related quantity or quantities on the basis of the complex number values of the voltage and the current undergone the Fourier transform processing.

The sampling frequency correcting means 5 controls the voltage output applied to the VCO 6 on the basis of the data inputted from the A/D conversion data packing means 2 such that the A/D conversion sampling frequency is locked to a multiple L of the power source frequency, where L represents a natural number.

The VCO 6 converts the voltage output of the sampling frequency correcting means 5 to a clock output. The clock output applied to the A/D converter 1 from the VCO 6 represents the sampling frequency in the case where the A/D converter 1 is implemented as the sequential type A/D converter while it represents an oversampling frequency when the A/D converter 1 is implemented as the delta-sigma-type A/D converter. In any case, when the A/D converter 1 is imparted with the function for dividing the clock signal, the output of the VCO 6 is the clock output which has not undergone the frequency dividing operation.

Next, referring to Figs. 3 and 4, description will made of the operation of the electronic electricity meter according to the first embodiment of the invention shown in Fig. 2.

In the firstplace, referring to Fig. 3, description will be directed to the sampling frequency correcting means Figure 3 shows a vector space which is defined by a real axis (abscissa) and an imaginary axis (ordinate) similarly to the vector diagram shown in Fig. 1.

As canbe seen inFig. 3, thephaseof the fundamental wave lags as the power source frequency becomes low while it leads as the latter becomes high. For this reason, the A/D conversion sampling frequency outputted from the VCO 6 is so controlled that the sampling frequency increases as the phase of the fundamental wave leads while the sampling frequency decreases as the phase of the fundamental wave lags. As one of the simplest methods for carrying out such control, there can be mentioned a feedback control.

In this conjunction, let's represent the feedback factorby E them-thVCOcontrolvoltage theVCOcontrol voltage at them-th cycle) byVcntrl-m and the phase difference between the m-th voltage fundamental wave and the (m+l)-th voltage fundamental wave by I (of positive or plus polarity when the phase leads). Then, the VCO control voltage Vcntrl m+latthe (m+l)-th cycle is givenby the undermentioned expression (20) Vcntrl_,,,+ Vcntrl_,,, At this juncture, it is presumed that the clock frequency increases as the VCO control voltage value becomes higher. The feedback factor E (E 0) can be determined on the basis of the VCO 6 designed properly for control and the follow-up speed thereof.

When the phase difference is employed intactly as the error quantity for effectuating the feedback control in accordance with the above-mentioned expression trigonometric functional arithmetic for determining the phases is required, incurring increase of the operational overhead. Accordingly, the quantity which bears one-to-one correspondence to the phase difference T and which exhibits a monotone increasing or decreasing relation thereto is used as the error quantity. By making use of such error quantity Error, the above-mentioned expression (20) can be rewritten as follows: Vcnl _,Vcn Vcntrl 6 Error (21) The error quantity Error mentioned above assumes a positive value (value of plus sign) when the power source frequency is low when the phase lags) while assuming a negative value (value of minus sign) as the power source frequency increases when the phase leads).

Accordingly, the value whose polarity is inverted for the monotone increasing and which remains unchanged for the monotone decreasing can be used as the error quantity Error for the phase difference.

The frequency follow-up processing described below is executed with a view to decreasing the operational overhead by substituting a quantity other than the phase for the error quantity Error.

The VCO control voltage Vcntrl (voltage output of the sampling frequency correcting means 5) is ordinarily generated by a digital-to-analog or D/A converter.

This is for the reason that the follow-up speed becomes low because of necessity of using a low-pass filter when e.g. a PWM (Pulse Width Modulator) output is used.

Incidentally, a low-pass filter may be inserted in precedence to the D/A conversion with a view to preventing oscillation of theVCOcontrolvoltageVcntrl. Inthat case, the follow-up speed will become low.

Although the following description will be made by taking as an example the most basic feedback control, it should be appreciated that any other appropriate technique can be resorted to so far as the VCO control voltage Vcntrl can be determined on the basis of the error quantity Error.

Figure 4 is a view for graphically illustrating the power source frequency, in which time is taken along the abscissa with the power source amplitude being taken along the ordinate.

The sampling frequency of the A/D converter 1 is so controlled that the leading edge of one of the voltages sampled at the points L (A/D conversion values) crosses zero in the case where the sampling frequency of the A/D converter 1 is selected equal to the power source frequency multiplied by k (equal to a positive integer multiple of the power source frequency and 2 N thereof).

In that case, when the power source frequency is locked perfectly unless the power source frequency undergo variation), the A/D conversion value of the voltage becomes zerouponeverysampling. However, theA/Dconversion value assumes a positive polarity (plus sign) as the power source frequency increases while assuming a negative polarity (minus sign) as the power source frequency decreases (see Fig. 4).

Since the error quantity Error bears one-to-one correspondence to the phase difference and exhibits the monotone increasing within the range of ±90 degrees, the A/D conversion value at the sampling point selected upon every sampling can remain unchanged except that the polarity (sign) is inverted.

However, for effectuating the lock at the frequency equal to the sampling frequency multiplied by a natural number or divided by a natural number, the VCO control voltage Vcntrl has to be limited to a range of values in which the sampling frequency multiplied by a factor in a range of 1/2 to 2 becomes effective.

In the electronic electricity meter according to the first embodiment of the present invention, the sampling frequency is set equal to a positive integer multiple of the power source frequency and 2 Nthereof. Accordingly, the lock can be effectuated even at a frequency which is equal to the sampling frequency multiplied by 1/2. Further, limitation shouldbeimposedtothe feedbackquantity" E Error"appearing in the expression (21) mentioned previously.

In the case of the example now under consideration, the sampling frequency is locked to the rising edge of the power source voltage at the zero-cross point. It should however be appreciated that the sampling frequency may be locked to the power source voltage at the falling edge thereof at the zero-cross point. In this case, the A/D conversion value itself can be used as the error quantity Error.

As is apparent from the above, by virtue of such arrangement that the sampling frequency of the A/D converter 1 is caused to follow up a positive integer multiple of the power source frequency, it is possible to obtain the harmonic component(s) directlyfromtheresultsoftheFouriertransform performed by the Fourier transform means 3. Thus, not only the electric energy of the fundamental wave component but also that of the harmonic component(s) can be measured with a simplified hardware structure.

Additionally, because the A/D converter whose sampling frequency is several kHz and the CPU whose clock frequency is several MHz can be employed in combination with the VCO 6 without need for using the A/D converter of extremely high accuracy as the A/D converter 1, the electronic electricity meter can advantageously and profitably be implemented in a monolithic integrated circuit without incurring any appreciable increase in the chip area.

Further, because the oscillation frequency is controlled by the VCO 6, the control accuracy can be enhanced when compared with the case where the sampling frequency of the A/D converter 1 is directly controlled by making use of the clock signal of the CPU, whereby the electric energy of the harmonic component(s) can be measured with high accuracy as well.

Besides, since the oversampling frequency can finely be regulated by the VCO 6 even when the delta sigma type AD converter is employed as the A/D converter 1, the electric-power-related quantities inclusive of the electric energy can be measured with high accuracy, and thus the electronic electricity meter can advantageously be implemented in the form of a monolithic integrated circuit.

Furthermore, from the standpoint of the operation speed the sampling frequency of the A/D converter 1 should preferably be set to be equal to a positive integer multiple of the power source frequency and 2 N thereof with FFT being carried out in the Fourier transform means 3.

Moreover, by virtue of the arrangements described above, the electric-power-related quantities such as effective power, effective electric energy, reactive power, reactive electric energy, apparent power, distortion power, effective current value, effective voltage value, phase difference between current and voltage or these values of harmonic components of concerned degrees which could not be computed or measured with the conventional technique known heretofore can be computed or measured.

By way of example, the reactive power required for computing the power supply efficiency can be measured with high accuracy together with those of the harmonic components, whereby the power factor serving as the index for the effective electric power utilization can be acquired with high accuracy.

In addition, the sampling frequency correcting means 5 can be realized by the arithmetic function of the CPU.

Further, thearrangementdescribedabovecanberealizedsimply by combining the VCO 6 with the sampling frequency correcting means 5. Thus, the hardware configuration of the electronic electricity meter can be simplified.

Furthermore, since the sampling frequency correcting means 5 is locked with reference to the zero-cross point (rising or falling) on the basis of the data outputted from the A/D conversion data packing means 2, the CPU can be get rid of the burden for performing complicated arithmetic operations, which further contributes to simplification of the structure of the electronic electricity meter.

Further, because the follow-up of the error quantity Error appearing in the expression (21) mentioned previously is realized through the feedback control, there canbe realized the electronic electricity meter which ensures excellent follow-up or tracking capability and hence excellent real-time-based performance.

Embodiment 2 In the control of the voltage-controlled oscillator 6, the absolute phase of the fundamental wave of FFT may be locked at a same position.

Figure 5 is a block diagram showing a circuit arrangement of a major portion of the electronic electricity meter according to a second embodiment of the present invention, in which the absolute phase of FFT is locked at a same position.

Incidentally, in Fig. 5, constituents same as or equivalent to those described previously (see Fig. 2) are denoted by like reference numerals each affixed with and detailed description thereof will be omitted in the description which follows.

In the electronic electricity meter now concerned, theFouriertransformmeans 3Aconstitutesasamplingfrequency control means through cooperation with a sampling frequency correcting means 5AandaVCO (voltage-controlled oscillator) 6A and is comprised of a means for detecting the absolute phase of the power source voltage and a means for detecting a lag or a lead of the absolute phase as a voltage phase difference.

Further, the sampling frequency correcting means is designed to generate the VCO control voltage to be applied to the VCO 6A on the basis of the data derived from the output of the Fourier transform means 3A.

Figure 6 is a view for illustrating a correcting processing for causing the sampling frequency of the A/D converter 1A to follow up a positive integer multiple L of the power source frequency in the electronic electricitymeter according to the second embodiment of the invention and shows a vector spacedefinedbyareal axis (abscissa) andanimaginary axis (ordinate) similarly to Figs. 1 and 3 mentioned previously.

As described previously, the absolute phase of the fundamental wave of the voltage assumes the same value every cycle so long as the sampling frequency of the A/D converter 1A is locked to a multiple L of the power source frequency where L represents a positive integer or natural number.

However, the absolute phase rotates in the lagging direction as the power source frequency becomes low while it rotates in the leading direction as the power source frequency becomes high.

Firstly, let's consider the case where a preceding value Vlcmp_pre and a current value Vl_cmp of the complex number value derived through FFT performed for the fundamental wave are represented by the undermentioned expression (22).

Vcmp _pre V pre _re+j.V pre _im V cmp V re+j Vl_im On the above presumption, the phase difference Phase V1 Error can be determined in accordance with the following expression (23): Phase _V1 _Error V arctanVpre _nim-V1 _re-V1_pre_re-V1_ imn (23) VI_ prere.V1 re+V1_ preirn-V_ im) where the range of the phase difference Phase_VlError is degrees.

Further, assuming that the amplitude of the voltage is substantially constant at every sampling, the denominator of the expression (23) can be regarded to be a constant value.

On the other hand, the numerator of the expression (23) bears one-to-one correspondence to thephase difference and exhibits a monotone decreasing. Accordingly, assuming that the error quantity Error is represented by the numerator of the expression the error quantity Error can be expressed by the undermentioned expression (24).

Error VI _pre _im. V _re-Vl_ pre reV im (24) In that case, when the phase difference is greater than ±90 degrees inclusive, the sign (polarity) is inverted.

Since the VCO control voltage Vcntrl is locked at the frequency equal to a positive integer multiple of the sampling frequency or a quotient of division thereof by a natural number, it is necessary to limit the VCO control voltage Vcntrl to a value in the range within which the sampling frequency multiplied by a factor in a range of 1/2 to 2 becomes effective. Besides, limitation should also be imposed to the feedback quantity Error).

With the structural arrangement of the electronic electricity meter according to the second embodiment of the present invention, the effect mentioned below can be obtained in addition to those of the electronic electricity meter according to the first embodiment (locking upon rising or falling at the zero-cross point) described previously.

Namely, since the phase of the voltage (or current) undergone the Fourier transform (preferably FFT) through the Fourier transform means 3A is locked, the electronic electricity meter according to the instant embodiment is excellent in respect to the high accuracy performed upon occurrence of supervision of white noise and harmonic components when compared with the case where the voltage wave is locked at the zero-cross point as in the case of the electronicelectricitymeteraccordingtothe firstembodiment of the invention described previously.

Embodiment 3 In conjunction with the second embodiment of the invention, no description has been made concerning the reference value of the absolute phase. At this juncture, it should be mentioned that the absolute phase of the fundamental wave determined through FFT can be locked at 0 (zero) degree, degrees, 180 degrees or 270 degrees.

Now, a method of controlling the VCO (voltage-controlled oscillator) according to a third embodiment of the invention will be described. For carrying out the VCO control method now concerned, the position for fixing the coordinate of the fundamental wave of the voltage (absolute phase of the fundamental wave of FFT) is selected atanyoneof 0 degree, 90 degrees, 180 degrees and270 degrees.

Figure 7 is a view for illustrating a VCO control operation in the electronic electricity meter according to the third embodiment of the invention and shows a vector space defined by a real axis (abscissa) and an imaginary axis (ordinate) similarlyto Figs. 1, 3 and 6mentionedpreviously.

At first, let's consider the case where the fundamental wave of the voltage is locked at 0 degree.

Since the phase lags as the power source frequency becomes low, the real value V1_im appearing in the expression mentioned previously assumes negative polarity (minus sign) whereas the real value Vlim assumes positive polarity (plus sign) as the power source frequency becomes high.

The real value V1 im bears one-to-one correspondence to the phase difference within the range of degrees and bears a monotone increasing relation to the phase difference. Thus, the error quantity Error can be expressed by the undermentioned expression Error -V1 im At this juncture, it is also necessary to limit the VCO control voltage Vcntrl and the feedback quantity as in the case of the electronic electricity meter described previously in conjunction with the second embodiment of the invention.

With the method described above, the operational overhead involved in computing the error quantity Error can be reduced when compared with the second embodiment described previously. Besides, storage of the coordinates in the preceding cycle (immediately preceding cycle) is rendered unnecessary.

When the phase difference exceeds ±90 degrees, the error quantity Error does no more bear a monotonous increase relation but the polarity (sign) remains unchanged. Thus, by setting the error quantity Error in accordance with the undermentioned expression the error quantity Error can be made to decrease monotonously relative to the phase difference in the range of ±180 degrees.

ifV1 _reO0 Error -V1 _im else if VI ima 0 Error Vrmsl -VI im) else Error Vrmsl -V1 im) Ordinarily, the effective value Vrmsl of the fundamental wave of the voltage scarcely undergo change.

Accordingly, this effective value can be handled as a constant as well. By adopting this method, the feedback range of degrees can be made to be of ±180 degrees.

The error quantity can be defined similarly for the lock at 90 degrees, 180 degrees or 270 degrees.

Namely, the error quantity Error for the lock at degrees can be expressed by the undermentioned expression (27) similarly to the above-mentioned expression Error V1 re (27) Further, the error quantity Error when the lock is effected at 180 degrees can be expressed by the undermentioned expression (28): Error VI im (28) Furthermore, the error quantity Error for the lock at 270 degrees can be expressed by the undermentioned expression (29): Error -VI re (29) Besides, the error quantity Error for 90 degrees can be expressed by the undermentioned expression similarly to the expression (26) mentioned previously: if V _im 0 Error V _re else if Vl_re0 Error 2* Vrmsl V1 re else Error Vrmsl VI re Further, the error quantity Error in the case of 180 degrees can be expressed by the undermentioned expression (31): if Vl_re 0 Error V _im else if V1 im 0 Error 2* Vrmsl V1 im else Error -2 Vrmnsl VI im Furthermore, the error quantity Error in the case of 270 degrees can be expressed by the undermentioned expression (32): if VI _i 0 Error -V1 re else if Vl re 0 Error Vrmsl VI re) else Error Vrmsl V1 re) Asisapparentfromtheabove, bylockingtheabsolute phase of the fundamental wave of FFT at 0 (zero) degree, degrees, 180 degrees or 270 degrees, the feedback range of degrees can be extendedto ±180 degrees totherebybroaden the range for adjustment, providing an advantageous effect in addition to those mentioned hereinbefore in conjunction with the second embodiment.

Embodiment 4 In the description of the electronic electricity meters according to the first to third embodiments of the invention, no consideration is paid to the number of bits handled by a digital-to-analog or D/A conversion unit. In this conjunction, it shouldbementionedthattheD/Aconverter of a small bit number can be employed in carrying the invention as well.

Figure 8 is a block diagram showing a circuit arrangement of a major portion of the electronic electricity meter accordingtoa fourth embodimentofthepresent invention.

Incidentally, constituents same as or equivalent to those described previously (see Figs. 2 and 5) are denoted by like reference numerals each affixed with and detailed description thereof will be omitted in the description which follows.

It should further be added that in Fig. 8, only a peripheral portion of the sampling frequency correcting means and the VCO (Voltage-Controlled Oscillator) 6B which are to be combined with a D/A converter of a small bit number are shown.

Referring to the figure, the sampling frequency correcting means 5B includes a D/A converter of a small bit number.

An attenuator 51 and an adder 52 are inserted between the sampling frequency correcting means 5B and the VCO 6B.

The adder 52 serves for setting the VCO control voltage offset in correspondence to the D/A converter of the small bit number.

The adder 52 is deigned to output to the VCO 6B the VCO control voltage added with the offset voltage VOFF so that the VCO control voltage has a normal or prescribed frequency 60 Hz) of the power source. With this arrangement, the frequency which can be controlled per bit of the D/A converter can be regulated finely.

In the case where the range of the clock frequency outputted from the VCO 6B is large, the number of bits required for the D/A converter increases in order to control finely the sampling frequency. Under the circumstances, control is performed such that the clock frequency outputted from the VCO 6B becomes closer to or assumes the prescribed frequency 60 Hz) of the power source when the output of the D/A converter is Preferably, the adder 52 is so designed as to add the offset voltage VOFF so that the clock frequency of the VCO 6B is equal to the prescribed frequency of the power source.

Figure 9 is a view for illustrating the control operation of the electronic electricity meter according to the fourth embodiment of the invention and shows relations among the output voltage of the D/A converter (sampling frequency correcting means 5B), the output voltage of the attenuator 51, the VCO control voltage and the offset voltage

VOFF.

Referring to Fig. 9, by controlling the output voltage of the D/A converter with reference to the offset voltage VOFF, the bit number required for the D/A converter can be suppressed or reduced, which means in turn that the electric-power-related quantities can be measured with high accuracybyusinganinexpensiveD/Aconverterofasmall scale.

Next, description will be directed to an arrangement whichallowstheVCOcontrolvoltagetobeadjusted. Figure is a block diagram showing in concrete a circuit arrangement including the attenuator 51 and the adder 52 shown in Fig. 8, wherein the adder 52 is so arranged as to be capable of adjusting the offset voltage VOFF.

Referring to Fig. 10,the attenuator 51 is comprised of a resistor R1 and a variable resistor R2 for dividing the output voltage of the D/A converter and is so designed as to adjust the range for controlling the VCO control voltage by adjusting the divided voltage.

Further, the adder 52 is comprised of a resistor R3 and a variable resistor R4 for dividing the offset voltage VOFF and a voltage adder circuit 52B for adding together the divided voltage of the D/A output voltage (output voltage of theattenuator51) andthedividedvoltageoftheoffsetvoltage VOFF to thereby output the VCO control voltage.

The adjustable parts of the variable resistors R2 and R 4 areprovidedoutsideoftheelectronicelectricitymeter so that they can arbitrarily be manipulated externally of the electronic electricity meter.

ByvirtueofthecircuitarrangementshowninFig. not only the range of the VCO control voltage but also the offset voltage actually added is rendered adjustable.

With the term "range" of the VCO control voltage, itiscontemplatedtomeanthevoltagerangeswhichcorresponds to the length of the arrow indicating the VCO control voltage in Fig. 9 in the case where a frequency in the range of Hzto 66Hz, bywayof example, isallocatedto theD/Aconverter of 8 bits.

Further, the offset voltage VOFF corresponds to a deviation from0V (zerovolt) shown inFig. 9 (refer to arrow).

In the electronic electricity meter now under consideration, this deviation can be adjusted as occasion requires.

More specifically, by adjusting the variable resistor R2 incorporated in the attenuator 51, the range of the VCO control voltage can be changed, whereas the offset voltage actually inputted to the voltage adder circuit 52B can be changed by adjusting the variable resistor R4 which constitutes a part of the adder 52.

In the electronic electricity meter according to the instant embodiment of the invention, the control range oftheD/Aconverter (oroffsetmagnitudeoftheoffsetvoltage) is rendered to be variable. By virtue of this feature, the electric-power-related quantity (or quantities) can be measured with high accuracy in conformance with the power line to which theelectronicelectricitymeterisapplied. Besides, the D/A converter can be implemented inexpensively on a small scale even in the case where the prescribed frequency changes.

Embodiment In the electronic electricity meter described above in conjunction with the second to fourth embodiments of the invention, an electric energy pulse output means 7 for outputting a pulse signal indicative of an electric energy may additionally be provided in a stage succeeding to the electric-power-related quantity arithmetic means 4C.

Figure 11 is a block diagram showing a circuit arrangement of a major portion of the electronic electricity meteraccordingtoa fifthembodimentofthepresentinvention.

Referring to Fig. 11, the electric energy pulse output means 7 is designed for sampling the electric power computed at a fixed clock timing to thereby perform a pulse output processing.

In general, in the electric energy meter, it is required that the electric energy pulse be outputted at a shorter interval than one period of the power source.

Inthecaseoftheconventional electricenergymeter, the electric energy is computed or arithmetically determined through direct multiplication of the current I and the voltage V. Accordingly, the requirement imposed on the output of the electric energy pulse can be satisfied without difficulty.

However, in the electronic electricity meter according to the present invention, because the current and the voltage are computedonaperperiodbasisthroughFourier transform (FFT), whereon the electric energy is determined on the basis of the result of the arithmetic operation mentioned just above.

Consequently, the electric energy pulse is generated on aper period, which cannot satisfy the above-mentioned requirement.

It is assumed, by way of example only, that the electric-power-related quantity arithmetic means 4C shown in Fig. 11 is activated upon every completion of the FFT arithmetic processing for thereby computing the desired electric power quantity such as the effective power, reactive power, apparent power or the like.

In that case, by holding the length or duration of the one period of the power source during every period in a counter or the like and by multiplying the counter value (one period of the power source) and the electric power value and summing the products, the electric energy (time-quadrature value of the electric power) can certainly be obtained.

However, with the arithmetic processing mentioned above, the electric energy pulse output means 7 can output no more than one electric energy pulse in every one period of the power source.

With the teaching of the present invention incarnated in the fifth embodiment, it is contemplated to improve the real-time-based performance of the electronic electricity meter in which the Fourier transform is adopted by solving the problem mentioned above. In the following, operation of the electronic electricity meter according to the fifth embodiment of the invention will be described.

Figure l2isatimingchart forillustratingelectric power computing operation and electric energy pulse generation/output operation performed by the CPU in the electronicelectricitymeteraccordingtothe fifth embodiment of the invention. In Fig. 12, t, t+l, denote correspondingly the data contents at every execution timing of processing operations.

At this juncture, it should be mentioned that the A/D conversion data packing means 2C has a function for detecting a packing time taken for packing the data at the point L, while the electric-power-related quantity arithmetic means 4C serves for a function for outputting the electric power computed in response to each of sampling commands.

At first, thetime (timecorrespondingtooneperiod) taken for the A/D conversion data packing means 2C to pack the data at the point L in the timing shown at the topmost row in Fig. 12 is recorded.

In succession, the CPU performs Fourier transform of the packed data with the aid of the Fourier transform means 3C to thereby compute the electric power (first electric power value) with the electric-power-related quantity arithmetic means 4C (see second row in Fig. 12). In that case, the data contents at the electric power computation timing undergo a lag corresponding to one period as compared with the contents at the timing for data packingprocessings (topmost row) (see t-1, t, Subsequently, the time taken for packing the data at the point L (recorded time, see the topmost row) and the electric power computed on the basis of the above-mentioned packeddata powercomputedaftertheFouriertransform, see the second row) are transferred to the electric energy pulse outputmeans 7 fromthe electric-power-related quantity arithmetic means 4C (see the third row in Fig. 12).

In that case, the duration of the time (cell width shown at the third row) taken for the transfer to the electric energy pulse output means 7 changes in dependence on the time taken for recording. The data contents at this electric power output timings undergo a lag corresponding to two periods as compared with the contents in the timing for the data packing processings (see t-2, t-l, The CPU performs sampling of the electric power transferred to the electric energy pulse output means 7 periodically at a predetermined interval (at a fixed clock) shorterthanoneperiodofthepowersource frequencytothereby add together the sampled electric power values (values shown at the third row, second electric power) to thereby arithmetically determine the electric energy (see the fourth row in Fig. 12).

More specifically, in the case of the example illustrated in Fig. 12, the sampling is performed about six timesperperiodofthepowersourcefrequency, andtheelectric energyofasamevalue is cumulativelyaddeduponeverysampling of the power output corresponding to the same data packing.

Finally, the electric energy pulse output means 7 outputs the electric energy pulse every time the cumulatively added value of the above-mentioned electric energies has attained a desired electric energy (see the fifth row in Fig. 12) With the arrangement described above, the electric energy pulses can be outputted with high accuracy while satisfying the requirement for the real-time-based performance.

Incidentally, it is to be noted that since the accuracy can be increased as the sampling period is selected shorter, the predetermined period (fixed clock) mentioned previously should preferably be set as short as possible when compared with the electric energy pulse output period.

Embodiment 6 In the electronic electricity meter according to the fifth embodiment of the invention, any reference has not especially been made to. However, the phase difference betweenthevoltageandthecurrentmaypreferablybe corrected on a per harmonic basis.

Figure 13 is a block diagram showing a circuit arrangement of a major portion of the electronic electricity meteraccordingto a sixthembodiment ofthepresent invention, which is so implemented that the phase difference between the voltage and the current can be corrected on a per harmonic basis. Incidentally, constituents same as or equivalent to those described previously (see Fig. 11) are denoted by like reference numerals each affixed with and detailed description thereof will be omitted in the description which follows.

Referring to Fig. 13, a phase correcting means 8 is inserted between the Fourier transform means 3D and the electric-power-related quantity arithmetic means 4D for correcting the phase difference between the voltage and the current on a per harmonic basis by arithmetically rotating the absolute phase of the voltage or alternatively that of the current.

In general, the current sensor (instrument current transformer CT or the like) and the voltage sensor (instrument potential transformer PT or the like) exert influence to the phases of the voltage and the current on the power supply line.

Further, the phases of the voltage and the current are also affected by the analog circuitry disposed on the input side of the A/D converter 1D.

Such being the circumstances, in order to measure accurately the electric-power-related quantity (or quantities) on the power supply line, the phase distortions brought about by the analog circuitry mentioned above have to be corrected.

Bywayofexample, inthecasewherethephasebetween the voltage and the current of the n-degree harmonic component is such as illustrated in Fig. 1, the voltage is rotated by (0 0 or the current is rotated by (q 6 when the phase difference on the power supply line is On the other hand, unless the phase difference on the power supply line is the voltage or the current may be rotated so that the desired phase difference can be realized.

At this juncture, let's assume that the new current Incmpnew obtained through the current rotating arithmetic processing is represented by the following expression (33): hI7 cnp_ new 1n7 new -e j-In newi imn (33) In that case, the new current In_cmp_new can be determined or computed in accordance with the undermentioned expression (34): In new re =cos(A-0) -sin(-0) In _re (34) In new -im) sin(-0) cos( In _im) For computing the electric-power-related quantity or quantities, the value of the new current In_cmp_new is used.

By performing this correction for each of the harmonic components, the electric-power-related quantity or quantities can be determined with high accuracy.

In the foregoing description, it has been presumed that the voltage or the current is selected as the electric-power-related value and the absolute phase of the selected one is arithmetically rotated to thereby correct the phase difference on a per harmonic basis. However, it goes without saying that the effective power and the reactive power computed by the electric-power-related quantity arithmetic means 4D may be arithmetically rotated, substantially to the same advantageous effect.

INDUSTRIAL APPLICABILITY As is apparent from the foregoing description, it is possible according to the teachings of the present invention to compute or arithmetically determine the effective power, theeffectiveelectric energy, thereactivepower, thereactive electric energy, the apparent power, the distortion power, the effective current value, the effective voltage value (effective value of voltage) and the phase difference between the current and the voltage as well as the electric-power-related quantity (or quantities) suchas those mentioned above of harmonics of various degrees. Thus, the present invention can effectively and profitably be applied 45 to the electronic electricity meter and the electricpower-related quantity arithmetic circuit designed not only for home use in general but also for industrial utilities for which electric energy control is required on a time-zone basis. Further, because the reactive electric energy can be measured as well, the electronic electricity meter according to the present invention can profitably find application to the electronic electricity meter and the electric-power-related quantity arithmetic circuit not only for the consumers where power factor management is necessary but also for those who use the electric machines such as the inverter and others whose operation is accompanied with generated of harmonics.

A reference herein to a prior art document is not an admission that the document forms part of the common general knowledge in the art in Australia.

For the purposes of this specification it is to be clearly understood that the word "comprising" means "including but not limited to", and that the word "comprises" has a corresponding meaning.

Claims

1. An electronic electricity meter having a microprocessor (CPU) which comprises an A/D converter for converting measurement signals indicative of a current and a voltage of a power line into digital values to be fetched and electric energy arithmetic means for thereby arithmetically determining an electric energy of said power line on the basis of said digital values, wherein said electronic electricity meter comprises: power source frequency detecting means for detecting a power source frequency of said power line; sampling frequency control means provided separately from a clock circuit of said microprocessor (CPU) for thereby controlling a sampling frequency of said A/D converter such that said sampling frequency is set to be equal to a positive integer multiple of said power source frequency; and said electronic energy arithmetic means comprises a harmonic arithmetic unit for arithmetically determining electric energy of a harmonic component.

2. The electronic electricity meter as claimed in claim 1, wherein said sampling frequency control means controls said sampling frequency so that said sampling frequency is set to be equal to 2 N of the power source frequency, and that said electric energy arithmetic means comprises fast Fourier transform means and is so arranged as to arithmetically determine the electric energy of the harmonic component through fast Fourier transform. 47

3. The electronic electricity meter as claimed in claim 1, wherein said A/D converter is implemented in the form of a delta-sigma-type A/D converter for oversampling.

4. The electronic meter as claimed in claim 1, wherein said sampling frequency control means comprises: zero-cross point detecting means for detecting a zero-cross point rising or alternatively upon falling of said power source frequency; and power source phase difference detecting means for detecting a lag or lead of said power source frequency as a phase difference of the power source on the basis of an A/D conversion value of said power source frequency sampled at a time point after lapse of one period from a preceding zero-cross point upon rising or alternatively falling of said power source frequency, wherein said sampling frequency control means is so arranged that the sampling frequency of said A/D converter is controlled on the basis of said power source phase difference.

The electronic electricity meter as claimed in claim 1, wherein said sampling frequency control means comprises: absolute phase detecting means for detecting absolute phase of the voltage of said power line; and voltage phase difference detecting means for detecting as a voltage phase difference a lag or lead of said absolute phase at a time point after lapse of one period from the preceding absolute phase of said voltage, wherein said sampling frequency control means is so arranged that the sampling frequency of said A/D converter 48 is controlled on the basis of said voltage phase difference.

6. The electronic electricity meter as claimed in claim 5 wherein, wherein said voltage phase difference detecting means is so arranged that an absolute phase of 0 degree, degrees, 180 degrees or 270 degrees is set as a reference.

7. The electronic electricity meter as claimed in claim 1, wherein said sampling frequency control means comprises: sampling frequency correcting means for correcting the sampling frequency of said A/D converter; and that said sampling frequency correcting means comprises: control quantity arithmetic means for arithmetically determining a control quantity for said sampling frequency; D/A conversion means for converting said control quantity for said sampling frequency into an analog voltage to be outputted through D/A conversion; and offset means for offsetting the output voltage of said D/A conversion means, wherein said offset means is so arranged as to offset said output voltage of said D/A conversion means by a predetermined voltage toward a prescribed frequency of said power line.

8. The electronic electricity meter as claimed in claim 7, wherein said offset means comprises offset voltage regulating means for adjusting said predetermined voltage from outside. 49

9. The electronic electricity meter as claimed in claim 1, wherein said electric energy arithmetic means comprises: A/D conversion data packing means for packing measurement signals of the voltage and the current over one period of said power source frequency; an electric power arithmetic unit for arithmetically determining a first power value through Fourier transform of data packed by said A/D conversion data packing means; packing time detecting means for detecting a packing time taken for packing said voltage or current of one period; electric power output means for holding said first power value for every packing time and outputting said first power value as a second power value every time a sampling command is issued at a predetermined arithmetic operation period; and electric energy pulse output means for integrating said second power value to thereby output an electric energy pulse every time a quadrature value of said second power value attains a predetermined value.

The electronic electricity meter as claimed in claim 1, wherein said electric energy arithmetic means comprises: A/D conversion data packing means for packing measurement signals of the voltage and the current over one period of said power source frequency; an electric power arithmetic unit for arithmetically determining electric-power-related values and phase differences thereof for harmonic components, respectively, on a per degree basis through Fourier transform of data packed by said A/D conversion data packing means; and 50 phase correcting means for correcting said current or voltage of one period through rotational operation so that the phase differences of said electric-power-related values represent true phase differences of electric-power- related values on a measured system side.

11. An electric-power-related quantity arithmetic circuit, comprising: an A/D converter for converting measurement signals indicative of a current and a voltage of a power line into digital values to be fetched; A/D conversion data packing means for packing digital values for one period; Fourier transform means for performing Fourier transform of data packed by said A/D conversion data packing means; electric-power-related quantity arithmetic means for arithmetically determining electric-power-related quantities on the basis of result of transforation performed by said Fourier transform means; sampling frequency correcting means for arithmetically determining a correcting quantity for a sampling frequency of said A/D converter on the basis of a frequency of said current or said voltage; and a voltage-controlled oscillator for outputting a sampling frequency which changes in dependence on said correcting quantity to said A/D converter.

12. An electronic electricity meter substantially as herein described with reference to the accompanying figures. 51

13. An electric power related quantity arithmetic circuit substantially as herein described with reference to the accompanying figures. Dated this 2 nd day of April 2004 Mitsubishi Denki Kabushiki Kaisha By their Patent Attorneys GRIFFITH HACK